A novel APD-based speech bandwidth optical receiver

In a previous posting I wrote about a novel application of a JFET (Read about that in the article "Gate current in a JFET - The development of a very sensitive, speech-frequency optical receiver" - link) in which the flow of gate current was integral to the operation of a photodiode-based optical detector.  In testing this circuit, which included tests using an indoor "photon range" and out in the field, it was observed that the sensitivity of this circuit was, at "audio" frequencies, on the order of 8-20 dB better in terms of signal/noise ratio than any of the more conventional "TIA" (TransImpedance Amplifier - read about that circuit here - link) circuits that had been tried.

In the analysis of this circuit it was determined that several factors contribute to the ultimate sensitivity, including:

• The intrinsic noise of the JFET.  This can be minimized by hand-selection of the device itself for the lowest-possible noise as well as selecting a device that can operate at a higher drain current to reduce noise - or even the use of several JFETs in parallel.
• The contribution of noise by other circuitry.  In the design this was minimized through the use of a cascode circuit topology as well as the use of a low noise, high impedance current source to supply the bulk of the drain current and the complete avoidance of other components being connected to the photodiode-JFET circuit junction.
• The reducation of parasitic circuit elements, such as capacitance (including the Miller effect) that reduces the amplitude of the signals from the photodiode, particularly as the frequency increases, effectively reducing the signal-noise ratio.
• The noise contribution of the photodiode.

Of these factors, the majority of the noise would appear to be due to the JFET itself, particularly above the low audio frequencies frequencies (e.g. below 100Hz or so) where ¹/_F noise would dominate. One of the possible approaches to get better noise performance is to cool the circuitry, but this is fraught with difficulties related to condensation which would require that the device itself be sealed in an atmosphere (e.g. dry nitrogen) in a manner similar to that used to cool CCD imagers for astronomy.

Figure 1:
The outside view of the completed APD-based optical receiver.  Because
of its extreme sensitivity it must be well shielded to minimize the pick-up
of stray fields such as those from AC mains and radio transmitters/phones.
Click on the image for a larger version.

What else may be done to improve the performance?

Perhaps counter-intuitively, the use of a smaller photodiode can help a bit and provide at least as much signal output as a larger one, provided that the optics can focus the given amount of light from the distant source of light efficiently onto its active area:  A smaller device will have lower self-capacitance shunting a smaller amount of the AC currents being produced in response to the impinging, modulated light in addition to having a lower intrinsic noise contribution.  In the case of an optical receiver the active area of the device is less important than in some other applications as lenses and mirrors may be used to concentrate the light from the distant source onto the photoactive area.

When reducing the size of the device one must assure that the optics themselves will resolve the distant spot of light to an area that is not larger than the active area of the device as well as taking into account additional constraints with respect to the accuracy and stability of the aiming and pointing mechanisms.  For example, using reasonable-quality molded Fresnel lenses of common focal lengths (e.g. an f/D ratio of approximately unity) one can expect only to resolve a spot with a "blur circle" of approximately around 0.2mm at best while high-quality glass optics should be able to reduce this by an order of magnitude or better assuming a suitably-distant source, a corresponding small subtended angle and proper paraxial alignment and focus.  If the resolved spot of light is much larger than the active area of the device - perhaps due to the device being too small for the optics' ability to resolve or due to the quality of and/or misalignment of the lens(es) - there may be an additional loss of available optical energy and signal-noise ratio as some of the light from the distant source is being "wasted" when it spills beyond the active area of the photodetector.

For more information on "spot sizes" using inexpensive, molded plastic Fresnel lenses see the article "Fresnel Lens Comparison:  A Comparison of inexpensive, molded plastic lenses and their relative 'accuracy' and ability to produce collimated beams" - link.

Aside from the reduction of the size of the photodiode or cooling, where else may one eke out greater performance from this circuit topology?

The Avalanche Photodiode:

The Avalanche Photodiode (APD) is a type of photodiode that contains an internal mechanism for amplification.  Simply put a single photon has a given probability of mobilizing a single electron when it impinges the active area of a standard PIN photodiode.  In an APD, what might have been a single electron being loosed as in a normal PIN diode, that same single-electron event can cause the mobilization of many electrons via an "Avalanche" effect, providing amplification of the optical signal and hence the name.  The result of this intrinsic amplification is that the output signal from this diode from a given photon flux can be much higher than that of a standard PIN photodiode.

Because the signal from the Avalanche photodiode itself is amplified internally it is more likely to be able to overcome the effects of the capacitance on frequency response as well as the noise intrinsic to the JFET amplifier, support circuitry and components, providing the potential of producing a greater signal/noise ratio for a given signal. Typically an Avalanche photodiode is incorporated into a TIA (TransImpedance Amplifier) with good effect, but what about its use in the previously-described "Version 3" photodiode receiver circuit that utilizes JFET gate current?

The basic design:

From the previous article (link) one can see the basic topology of the "Version 3" circuit using a "normal" PIN photodiode depicted in Figure 2, below.

Figure 2:
A diagram of the "Version 3" optical detector that utilizes JFET gate current.  In this circuit Q1 and Q2 comprise a cascode
circuit with Q3 providing the majority of Q1's drain current while U1b is configured as a differentiator to compensate
for the low-pass effects of the intrinsic capacitance of D1, the photodiode and Q1.  Resistors R1 and R2 along with
C1 provide a filtered reverse bias for D1 which not only decreases its capacitance, but it also biases Q1 to
its operating state where it is drawing maximum drain current.  In this circuit the connection between the Photodiode (D1)
and the gate of the JFET is made in air and not on a circuit board to minimize capacitance, stray signal pickup and
most importantly a source of leakage currents and related noise.  The fundamental circuit around Q1 and its art was described in a 2008 article in an SPIE journal (Vol. 6878) written, in part, by the author.
Click on the image for a larger version.

In this design PIN photodiode D1, a BPW34, is reverse-biased via R1 and R2.  One of the main benefits of doing this is that the capacitance of D1 decreases from approximately 70pF at zero volts to around 20pF at the operational voltage, reducing the degree to which high frequency signal are attenuated by this capacitance.  A somewhat less tangible benefit of this is that in addition to photovoltaic currents produced by the impinging light, the bias also allows photoconductive currents to flow from the bias source, through the photodiode and into the gate of the JFET.  As noted in the original article it is the presence of the gate-source junction of the JFET (Q1) and its conduction that limits the gate-source differential to around 0.4-0.6 volts, permitting D1's reverse bias to become established without the need of any additional noise-generating or lossy components.  In this configuration the drain current of the JFET is still proportional to the gate-source voltage (but with an offset of drain current greater than the "zero bias" drain current) and like a bipolar transistor's base voltage and current, the relationship between gate voltage and gate current is logarithmic.

A question now comes to mind:  What about replacing D1 with an avalanche photodiode?

Testing with an Avalanche Photodiode:

Like its more-sensitive distant cousin, the Photomultiplier tube, the avalanche photodiode requires a rather high bias voltage in order to function at maximum gain.  Rather than requiring a kilovolt or so as is needed for a photomultiplier, typical photodiodes may operate with up to "just" a few hundred volts.  Like the photomultiplier, the current required for "dark" operation is minuscule - a few hundred microamps in these "dark" conditions is more than enough.

In perusing the various component catalogs I noted that Mouser Electronics carried some avalanche photodiodes - but as expected, there was a price - literally:  Around US$150 at the time for just one APD.  In a compromise between size, availability and cost I chose the AD1100-8-TO52-S1 by First Sensor  (previously known as "Pacific Silicon Sensor") - a device with a round, 1mm² (1.128mm diameter) active area - a reasonable compromise between cost, size, and the practical limit of the Fresnel-based optics.  This device, which came with its own test sheet, indicated a maximum gain ("M" factor) of approximately 1000 occurring at 134 volts at a temperature of 25C.

In most ways using an APD is just like using a reverse-biased PIN photodiode - except that the reverse bias voltage will be much higher.  Perusing the literature and manufacturer's specifications one will note that many designs depict a temperature-compensated bias voltage supply, but further investigation reveals that this is necessary only if the device is being used at/near maximum gain (and maximum voltage) and/or if it is necessary to precisely maintain the gain over a wide temperature range.  For our application we don't really care if the gain changes with temperature, so an arbitrarily adjustable high voltage supply is fine - and actually preferred.

In my initial research I noted that the internal action of any APD suffers an inevitable, but expected, effect:  As the gain goes up with increasing bias voltage, the intrinsic noise of the device itself increases at a faster rate than the gain.  What this means was that there is going to be a point above which a further increase of device gain will cause the signal to noise ratio to decrease even though the actual signal level continues to increase with bias voltage.  With this in mind, the question is "At what voltage might this happen, and would this 'crossover' point occur at a point where we can expect the overall 'gain+noise' to offer a net advantage over a PIN photodiode?"

Building a prototype receiver similar to that depicted in Figure 2 I substituted an APD for D1 using a string of sixteen 9 volt batteries and a 1 megohm potentiometer with a 100k resistor in series with the wiper (and some bypass capacitors to ground on the "hot" side of the diode) in lieu of R1 to set the bias voltage.  Placing this prototype in my "Photon Range" - a windowless room in my house where there is an LED mounted to the ceiling that may be modulated - I compared the sensitivity of this prototype to both my "standard" TIA receiver (the VK7MJ design) and an operational exemplar of my "Version 3" design.

Varying the voltage from 10 volts to around 140 volts I noted that at a bias voltage comparable to the reverse bias applied in Figure 2 (approx. 8 volts) the apparent sensitivity was roughly on par with that of the Version 3 unit using a normal PIN photodiode after the signal levels were corrected to compensate for the smaller area of the APD as compared with the BPW34 (e.g. 1mm² of the APD versus 7mm² of the BPW34 - the larger size gathering proportionally more light in this lens-less system).  At around 130-135 volts, the output of the APD-based prototype was very high, but the weak, optical signals from the test LED were lost in the noise.  In the area of 35-45 volts I observed that the overall signal levels, while significantly higher than they were at 8-10 volts, were a fraction of what they were at 130 volts but the signal/noise ratio was roughly 6-10dB higher than it was at the lowest voltage when the differences in active area of the APD versus the photodiodes in the test receivers were taken into account.  As expected, even though the signals were much "stronger" at the higher voltage, the signal noise ratio at that voltage was very poor and would have submerged a much weaker signal completely.

Comments:

• The test receivers used BPW34 PIN photodiodes with an active area of 7mm² while the APD has an active area of just 1mm².  Because there were no optics in front of the photodiodes in this test there was 7 times as many of the LED's photons hitting the larger device, resulting in an approximate 8.5 dB difference in signal/noise - assuming all other parameters being equal.  It is when using the device in this "lens-less" configuration that this factor must be accommodated.
• While it is theoretically possible to use a photomultiplier tube (PMT) in lieu of an APD, there are several practical concerns.  Even though an "S-1" type of photocathode has a peak in the red-NIR area, its low quantum efficiency makes it a rather poor performer overall.  The "931A" PMT - widely available surplus - has a more typical blue/violet peak response (type "S-4") in which the longer red wavelengths suffer greatly in terms of quantum efficiency.  Field testing of these devices by British amateur radio operators has shown that they offered no obvious advantage over the "Version 3" PIN photodiode design for "red" wavelengths.  As of the time of this writing the use of PMTs with more exotic photocathodes (such as multialkalai and GaAs) that are better suited for "red" wavelengths (but much more difficult to find surplus!) have not been field-evaluated.

A practical design:  The high voltage APD bias supply:

First, a few weasel words:

Even though the currents are very low, there is some risk of injury with the voltages involved (e.g. several hundred volts) and it is up to you to educate yourself about high voltage safety!

If you wish to construct these circuits, be aware of possible hazards and always assume that any capacitors are charged, even after power is removed.

You have been warned!

Because it is not convenient to carry around a lot of 9 volt batteries connected in series, a simple high voltage converter was designed to provide the  microamp-level current required for the APD bias supply and it is depicted below in Figure 3.

Figure 3:
High voltage supply for the APD receiver.  U101a is an oscillator that drives Q101 to produce a high-voltage, low-current bias for the APD.  The output is regulated via U101b and associated components to the voltage set by potentiometer R111.  R109 is used to set the highest voltage that may be obtained when R111 is adjusted for "maximum".  R109 is shown with the two "ends" grounded only because it was convenient to wire it this way when the prototype was built.
Click on the image for a larger version.

 This design is a simple "boost" type switching converter using a high voltage transistor and an inductor to produce the needed bias.  In this circuit U101A forms an oscillator that drives the high voltage transistor Q101, and when Q101 switches off, the magnetic field of L101 collapses, producing a high voltage spike that is rectified by D101 and filtered and stored by C102, R106 and C103.  To regulate this high voltage a sample is divided-down by R108 and R109 and compared with a 5-volt reference from U102 that is made variable with R111:  If the output voltage is too high, U101b turns on Q102 to pinch off the drive for Q101.  Because I used an "ordinary" op amp with an output that could not go all of the way to the negative supply rail, LED101 was put in series with the transistor's base to provide a drop of around 2 volts to assure that Q101 could be shut completely off.

Figure 4:
Inside the high voltage (bias) supply for the APD receiver.  Potentiometer
R111 and the indicator, LED101, are mounted in the front of the
case.  Both the high voltage generator and the receiver itself are powered
from a single 9 volt battery.  The typical combined current consumption
for the both sets of circuits is less than 35 milliamps.
Click on the image for a larger version.

LED101 also provides two other features:  It functions as a "power on" indicator, and since it is in series with Q101's base drive it is modulated at approximately 6.5 kHz (determined by experiment to be the frequency at which Q101 and L101 produced the highest voltage with the best efficiency) and can be used as an optical signal source to verify that the receiver is working.  Worth noting is that R112 is placed across the "hot" end and the wiper of R111 to "stretch" the high voltage end of the linear potentiometer's adjustment range a bit to compensate somewhat for the fact that near the maximum voltage, the gain goes up exponentially with the bias voltage, making fine adjustments at this setting easier.

The APD (optical) receiver:

The optical receiver section is depicted in Figure 5, below:

Figure 5:
The optical receiver which works in a manner very similar to that depicted in Figure 3.  In this implementation
the high voltage bias is applied to the cathode of D201, the APD, which has its anode connected to the gate of the JFET,
Q201.  Q201 and Q203 comprise a self-biasing, AC-coupled cascode amplifier while Q202 provides the a high-
impedance source for the bulk of Q201's drain current.  The components in the sections marked "HV Filter"
and "LV Filter" are used to keep the residual switching frequency energy from being conducted into these circuits.
As with other circuits of this type, the connection from the photodiode to the JFET's gate is made in air and not via a
circuit board trace - this, to minimize capacitance, leakage currents and noise.
Click on the image for a larger version.

Not surprisingly this circuit looks very similar to the "Version 3" optical receiver of Figure 2.  Notable features include an R/C filter consisting of R201, R202, C201 and C202 to remove traces of the 6.5 kHz power supply ripple from the high voltage supply while L201, C211, R215 and C212 do the same for the 9 volt supply that the receiver circuitry shares with the high voltage generator.  The two sections - high voltage supply and optical receiver sections - are separate, connected by a 3 foot (1 meter) umbilical cable, both to provide isolation of the extremely sensitive optical receiver from the electrostatic and electromagnetic fields of the high voltage converter and also to remotely locate the controls on the high voltage supply away from the lens assembly on which the receiver portion is mounted so that adjustments can be made without disturbing it.

Figure 6:
Inside the receiver portion of the APD receiver.  This section is physically
separated from the high voltage converter to prevent the switching energy
from getting into these extremely sensitive circuits.  In the center is
a small sub-board with the APD and JFET that is mounted on short pieces
of 18AWG wire to allow its position to be adjusted in all three dimensions
to provide both paraxial alignment and focus.
Click on the image for a larger version.

The APD itself is mounted on a small sub-board along with Q201 (the JFET) and the other capacitors noted in the box in Figure 5.  Most of Q201's drain current is provided by Q202's circuit, a current source, that operates at high impedance while Q203 is the rest of a cascode amplifier circuit that is designed to be self-biasing at DC and to provide gain mainly to AC signals.

The output of the cascode amplifier is passed to U201b, a unity gain follower amplifier.  This signal then passes to the circuit of U201a, a differentiator circuit that is designed to provide a 6dB/octave boost to higher frequencies to compensate for the similar R/C low-pass roll-off intrinsic to the APD and JFET itself:  Without this circuit, higher frequency audio components of speech would be rolled off, reducing intelligibility.  By design the frequency range of the differentiator and its surrounding circuitry is intentionally limited so that low frequencies (below several hundred Hz) are strongly rolled off to prevent AC mains related hum from urban lighting from turning into a roar as are very high frequencies - above 5-7 kHz - which would otherwise become an ear-fatiguing "hiss" were the differentiation allowed to continue to frequencies much higher than this.

An interesting property of the photodiode circuit is that the "knee" related to this 6dB/octave roll-off occurs varies somewhat with the bias voltage and thus amount of device capacitance and, to a certain degree, its gain.  Because of this the frequency response of the APD/JFET circuit and the differentiator don't match under all operating conditions but experience has shown that it is better to have a bit of extra "treble boost" than not when it comes to making out words when the distant voice is immersed in a sea of noise.

A sample of the output from U201b, before differentiation, is also passed to J20, the "Flat" output.  The audio taken from this point, lacking differentiation, will sound a bit muffled under normal low-light conditions as it is not subject to either the high or low pass effects of the U201a differentiator which means that it will pass both subsonic and ultrasonic components as detected by the APD amplifier itself.  On the low end, the sensitivity is limited by ¹/_F noise which becomes increasingly dominant below a few 10s of Hz while on the high end it is again the capacitance associated with the APD and JFET circuits.  In testing it was observed that at this "Flat" output it was possible to detect signals from an LED modulated up to several MHz, albeit with significantly reduced sensitivity.  The main purpose of this output is to provide a signal point suitable for both subsonic digital communications as well as ultrasonic for experimentation with low/medium rate data, FM carriers and SSB signals.

In this circuit the amount of drain current in the JFET will vary depending on the individual properties of the JFET itself, the bias voltage, and the amount of impinging light.  Under "dark" conditions the "standing" JFET current was set to approximately 7-10 milliamps by the current source and the drain-source voltage varied from around 0.21 volts when the APD bias was just 12 volt to around 0.155 volts when the APD was operating at its maximum rating of 135 volts.  The specified JFET, the BF862, is typically capable of handling more drain current than this - and to do so would likely reduce its noise contribution slightly - but it was set at this level (with R205) to moderate battery current consumption.

Circuit testing:

Although it may have risked component damage, the APD circuit was "torture tested" to check ruggedness.  In a completely dark room a xenon photo flash was set off just inches/centimeters away from the photodiode with the bias set at 135 volts.  While the receiver was deafened for a second or two - the time it took for the various circuits to recover (e.g. power supply, re-equalization of various capacitors, etc.) - repeated tests like this did not do any detectable damage to the receiver sensitivity or its noise properties indicating that the APD and JFET were more than rugged enough to handle any conceivable event that might happen in the field, aside from directly focusing the sun on the photodiode!

This circuit has also been successfully used in broad daylight.  While the receiver worked, the background thermal noise from the sunlit landscape was the limiting factor for sensitivity and the recovered audio had quite apparent nonlinearity (distortion) with an altered frequency response (e.g. "tinny") because the ambient light and resulting photodiode conductivity effectively shunted the high voltage bias and device capacitance.  In short, in such high ambient light conditions this circuit has no advantage over other optical receiver topologies such as the original "Version 3" or even a more conventional TIA (TransImpedance Amplifier) but its ability to be useful under such conditions is indicative of its versatility.

The results of in-field testing:

This receiver was first field-tested on a 95+ mile (154km) optical path during the September 2012 segment of the ARRL "10 GHz and up" contest:  For detail on this communication, read the blog entry "Throwing One's Voice 95 Miles on a Lightbeam" - link
 

Figure 7:
My end of the 95+ mile optical path during the session where the APD-
based optical receivers were first field-tested.  As seen in the picture
the optical path passes over urban lighting which tends to slightly raise
the noise floor due to both Rayleigh and lens-related scattering
effects.
Click on the image for a larger version.

During this test the optical (voice) link was first established using the "Version 3" PIN Photodiode receiver depicted in Figure 2.

With the reasonably clear air and the moderately long path we noted that we could reduce the LED current to a tiny fraction of the maximum before significant signal/noise degradation was noted.  At this lower LED current each station at opposite ends of the path switched from the PIN photodiode to the APD receivers and after tweaking our pointing and reducing the LED current even more we observed what turned out to be between 6 and 10 dB improvement in the signal-noise ratio - about what was observed on the indoor "Photon Range" with the initial prototype circuit.  It is likely that the actual improvement in sensitivity was greater than this, but because our respective optical paths passed directly over populated areas (see Figure 7) our ultimate noise floor was degraded by light pollution which included a thermal "hiss" from the urban lighting and a low-level, harmonic-rich 120 Hz hum.

As was determined in the lab, the best signal-noise ratio in the field occurred with the APD biased in the 35-45 volt range where the "M" (amplification) factor was in the area of 3-10 (approximately 10-20dB gain).  At this rather modest bias voltage the "Gain+Noise" from the APD itself was sufficient to overcome much of the intrinsic noise of the JFET.  At higher voltages the gain continued to increase but the signal-noise ratio decreased at a faster rate until the APD's own avalanche noise drowned out the desired signal.

* * *

For more information about (speech bandwidth) free space optical communication, check out these links from my "Modulated Light" web site (link):

• Using Laser Pointers for voice communications -  This page describes in more detail the methods by which one may successfully use inexpensive laser pointers to cast voice through many miles of the ether!
• A Highly-Sensitive Optical Receiver Optimized for Speech Bandwidth - This is one of the most sensitive speech-range optical receivers yet devised that uses standard photodiodes and is much more sensitive than the simple receiver depicted in figure 1, above.
• Receiver for low-bandwidth optical (through the air) communications using an Avalanche Photo Diode (APD) - This optical receiver is an enhancement of the one above, achieving another 6-10dB of ultimate sensitivity using an APD.  This receiver is about as sensitive as you can get without resorting to an exotic red-sensitive photomultiplier tube!
• A "Cheap" Optical Transceiver lens assembly - If you really want to improve your receive sensitivity, the best way to do this is with a large lens.  This article describes how one could use inexpensive foam-core poster board to make an assembly that will focus the distant light on the detector diode of an optical receiver.
• A "Mini" full-featured Pulse Width Modulator for high-power LEDs and laser diodes -This describes a simple, computer-based PWM modulator that will not only transmit audio, but generate test tones as well.

Be sure to check out the "ModulatedLight.org" web site's other pages as well!

[End] 

This page stolen from "ka7oei.blogspot.com". 
 

Gate current in a JFET: The development of a very sensitive, speech-frequency optical receiver.

Back in 2007-2008 I was working on equipment for "new" ham band - for me at least - the one that is now labeled as "...above 275 GHz" in the FCC rules.  As you might expect the most accessible portion of this infinity of electromagnetic spectrum is that containing visible light, and that is where I was directing my interest.

At this time "high power" LEDs were starting to appear on the market at reasonable prices, and by "high power" I mean LEDs that were capable of dissipating up to 5 watts, each.  What this meant was that from a single emitting die of rather small dimensions one could pump into it enough current and, with the good efficiency of the device, obtain a quantity of light that was suitable for long-distance optical communications.

To be sure, I was building on the fine pioneering work of others, including that of two Australians, Dr. Mike Groth (VK7MJ) and Chris Long (now VK3AML) who had determined that it was the noncoherent light produced by LEDs that offered the greatest probability of practical, very long-distance atmospheric optical communications.  (As a primer as to why this is the case, read the article Optical Communications Using Coherent and Noncoherent Light - link).

Optical receiver needed:

In the midst of producing the various pieces of equipment required for experiments in optical communications (e.g. optical transmitters, modulators, receivers, support equipment, etc.)  I was investigating the different circuit topologies of practical optical detectors.  My goal was not to achieve extremely high data speeds, but rather to use audio-frequency signalling (speech, tones) to start with and, perhaps, work up from there.

One of most common such detectors is the phototransistor - but I quickly dismissed that owing to its very small photoactive area and the fact that the various pieces of literature relating to weak-signal optical detection noted that they are inferior in comparison to practically any other device owing to their intrinsic noise level.  (CdS cells - article here -  were not seriously considered because they are too slow to respond - even for audio frequencies.)

One option was the venerable Photomultiplier Tube (article here) and while this was technically possible and, in theory, the best choice, it was ruled out because of its fragility (electrically and mechanically), its large size, the limited response at the wavelengths of interest (more below on that) and the need for a high voltage supply (around 1000 volts).

While these technical difficulties are surmountable I could not overlook the fact that available literature on these devices - and advice from the Australians, who'd actually used them - pointed out that there were but a few photomultipier tube types that have good sensitivity in the "red" end of the optical spectrum where there is also good atmospheric transparency - and even fewer of these rare types, in known-usable condition, available for a reasonable price on the surplus market!

Figure 1: 
The transimpedance amplifier in its simplest form. 
This circuit converts the photodiode currents
into a proportional output voltage.

The Transimpedance Amplifier:

This left me with the photodiode (article here) and the most commonly-seen circuit using this device is the "TIA" - TransImpedance Amplifier (article here).  As can be seen from Figure 1 this is very simple, consisting of just an operational amplifier with a feedback loop with the photodiode connected directly to the noninverting input.  In this circuit the photodiode currents are converted directly to voltage (hence the name) with the gain set by the feedback resistor with the added capacitor being used to assure stability, compensating for photodiode and op-amp capacitance.

This particular circuit has the advantage that it is very predictable and the frequency response can be determined by the combination of the bandwidth of the op amp and the intrinsic capacitance of the phototransistor.  To a degree, one can even increase the frequency response for a given set of devices by reducing the feedback, but this comes at the expense of gain and ultimate sensitivity.

In other words:  With photodiodes you can have high sensitivity, or you can have wide bandwidth - but not both!

Figure 2:
 A practical, daylight-tolerant TIA optical receiver circuit.  This has good sensitivity in both darkness and light and does not suffer from "saturation" in high ambient light conditions because of a built-in "servo" that self-adjusts the phototiode's virtual ground to offset photon-induced bias currents.  Because of this "servo" action this receiver does not have DC response like the circuit of Figure 1 with the low-end frequency being limited by the values of R104 and C106.
While the LM833 is a reasonable performer, there are other (more expensive!) op amps that have lower noise.
Click on the image for a larger version.

While very simple (there are even single-chip solutions such as the "OPT101" that include the photodiode, amplifier, and even feedback resistor in a clear package) there are some very definite, practical limitations to the ultimate sensitivity of this sort of circuit if the goal is to detect extremely weak, low-frequency currents.  When you get to very low frequencies, "1/f" noise (a.k.a. "flicker noise") becomes dominant from a number of sources and there are various other types of noise sources (thermal, shot, etc.) that can be produced by the various components.

As it turns out, this circuit - with practical op amps - has very definite limitations when it comes to trying to divine the weakest signals at low-ish frequencies (audio, sub-audio):  For an article on why this is so - and some of the means of mitigation - see the January, 2001 Electronic Design article, "What's All This Transimpedance Amplifier Stuff, Anyway?" - link by Robert Pease.

Figure 3: 
The VK7MJ optical receiver using TIA and cascode techniques - used as the "reference" optical detector.
The optional "daylight" circuit provides AC coupling to prevent saturation of the circuit under high ambient
light conditions at the expense of low-light performance.
Click on the image for a larger version.

One can build transimpedance amplifiers using discrete components that outperform most of the integrated-circuit based designs and for a reference circuit I constructed and used one devised by Dr. Groth, VK7MJ and depicted in Figure 3.  In this circuit one may see the feedback path via R3/R4 with compensating capacitor Cf.  In this particular circuit Q1, the input FET, is rather heavily biased to increase its "bulk current" (a term used in the referenced Robert Pease article) with Q2 acting as a cascode circuit - link (e.g. current-based) amplifier with subsequent follower stages.  Additionally, the photodiode itself (D1) is reverse-biased, reducing its capacitance significantly and thereby improving high frequency response.  By hand-selecting the quietest JFETs one can obtain excellent performance with this circuit and since it is discrete, there is room for adjusting values as necessary to accommodate component variations and for experimentation.

This particular circuit is quite good across the audio range from a few 10's of Hz to several kHz, but above this range it is largely the capacitance of the photodiode (at least for devices that have square millimeter-range surfaces areas) that quashes the high frequency response.  Even though the photodiode's capacitance - and that of stray wiring and the JFET itself - may be only in the 10's of picofarads, at hundreds of k-ohms (or megohms) even small amounts of capacitance quickly become dominant - another good reason to implement the aforementioned cascode circuit and its tendency to minimize the "Miller Effect" - link to help optimize frequency response.

The K3PGP circuit and variations:

Figure 4: 
The K3PGP Optical receiver.
Click on the image for a larger version.

Building the above circuit as a "reference" I began testing on a "Photon Range" - a darkened room in my basement with a red LED affixed to the ceiling - where I characterized the various receiver topologies.  In this environment a small and adjustable amount of current (10's of microamps, typically) would be fed to the LED, modulated at an audio frequency, and the receiver under test would be placed on the floor below with its output connected to a computer in an adjacent room running an audio analysis program such as "Spectran" or "Spectrum Lab" to measure the signal-noise ratio at different frequencies.  Before and after each session I would measure the performance of my "standard" optical receiver - the VK7MJ circuit - and use it as a basis of comparison.

The receiver named after K3PGP (see his web site - link) was the next receiver to be tested.  This receiver is much more sensitive than the VK7MJ receiver - at least at very low audio frequencies (<200 Hz) and as may be seen in Figure 4 it is devoid of a feedback mechanism and the connection between the photodiode and JFET is made directly, with no external biasing components of any kind.

While a seemingly simple circuit, there is more going on here than one might first realize:  Without any feedback or any other components between the FET and photodiode the opportunity to introduce noise from such components or reduce the signal from the photodiode in any way is minimized.  In fact, when constructing this circuit there is the strong admonition that the photodiode-gate connection to the JFET be done in mid-air (and that one clean both components with alcohol to remove residue!) as leakage paths on circuit board material can cause significant signal degradation!

Effectively, the K3PGP circuit acts as a charge integrator with the energy slowly (in relative terms) bleeding off due to the leakage of the photodiode, its photoconductivity, and the gate-source leakage currents of the FET itself.  While extremely sensitive at low frequencies - specifically those below 200 Hz - above this, the sensitivity and output suffers due to the rather long R/C constant associated with the high gate-photodiode leakage resistance and capacitance and, to a lesser degree, the Miller effect.  This circuit also functions only in total and near-total darkness conditions:  More light than that, the voltage across the photodiode reaches equalibrium while turning the FET "on", effectively quashing the signal.

Inspired by the above circuit I made the modification indirectly depicted in Figure 5, below:

Figure 5:
 The version "2.02" optical receiver, used as a test bed for various circuit configurations - see text.
For the "K3PGP" configuration the photodiode would be reversed from what is shown
in the drawing above and the anode grounded with nothing else connected at point "C".
Click on the image for a larger version.

This circuit was devised as a "test bed" and although not shown in the diagram, it was configured by connecting the cathode of the photodiode to the gate and grounding the anode and having no other photodiode-gate connections present - just as in the K3PGP receiver.

In this circuit one has a FET input and a cascode circuit - just like that of the VK7MJ circuit - to reduce the Miller effect, but this particular cascode circuit has a modification:  Q3 forms a current source, in parallel with the cascode, that supplies the bulk of the drain current for the JFET - several milliamps.  Because the amount of current provided by the current source - which has a high operating impedance and is largely "invisible" - is fixed (but adjustable by varying R4 to suit specific characteristics of Q1) and it is left up to the cascode to supply the remaining drain current - which varies depending on the gate voltage.  In this particular circuit, due to the "cascode action" the voltage at the drain of Q1 and emitter of Q2 varies very little while the cascode - which is allowed to bias itself at DC, but is bypassed at AC with C3 - produces the recovered modulation at the collector of Q2, greatly amplified.  From the collector of Q2, noninverting amplifier U1a amplifies the signal further and presents a low-impedance output.

In other words, it is mostly the K3PGP circuit, but with a cascode amplifier and higher FET drain current:  By reducing Miller capacitance with the cascode the frequency response was to be improved somewhat and by increasing the drain current the noise contribution of the FET itself should be reduced as noted in the Pease article mentioned above.

In testing it was observed that this particular circuit was, in fact, several dB more sensitive than the original K3PGP circuit and also that the frequency response was slightly better - but not as much as one might first think, mostly owing to the fact that it is mostly the photodiode capacitance that is limiting the response rather than the Miller effect - but every little bit helps!

I then rewired the circuit using the "Standard Config" noted in Figure 5 which, if you draw in the lines, converts it into a TIA circuit like that of the VK7MJ design with both adjustable reverse bias of the photodiode and adjustable feedback.  In this configuration the performance at very low frequencies was reduced, likely due to the noise contribution of the feedback resistor, increased leakage currents from the photodiode at reverse bias and also signal attenuation caused by the feedback submerging the lowest-level, low-frequency signals into the noise.  At "speech" frequencies it was slightly better than that of the VK7MJ receiver - probably due to the higher JFET current or, perhaps, random component variances - and the frequency response was also comparable to that of the VK7MJ circuit, the parameters varying according to the amount of applied feedback and compensation.

Improving the receiver:

My goal was a circuit that offered the sensitivity of the K3PGP circuit, but usable speech response - the latter not being available from the K3PGP circuit due to the R/C rolloff.  A quick check revealed that this was the typical 6dB/octave rolloff so I reconfigured the circuit, again, as a K3PGP-like circuit and followed it with an op-amp differentiator circuit with a breakpoint calculated to compensate for the measured "knee" frequency (e.g. that at which the 6dB/octave rolloff of the K3PGP circuit) began - the result being that I now had a fairly flat frequency response.  Not unexpectedly, while the signal-noise ratio was quite good at the very low frequencies, it decreased fairly quickly as it went up as that energy was simply submerged in the circuit noise.

In staring at the circuit, with the grounded anode of the photodiode, I wondered about reverse-biasing the photodiode to reduce the capacitance - but if I did this, how would I keep the voltage at the gate from rising without needing to add another (noise generating, signal-robbing) component to clamp it to ground?  Knowing that the gate-source junction of a JFET was much like that of a bipolar transistor in that there would be an intrinsic diode present, I knew also that the gate-source voltage would limit itself to 0.4-0.6 volts, but how would the FET behave?

Using JFET Gate current for "good":

In doing a bit of research on the GoogleWeb when I derived this circuit I could not come up with any sort of useful answer to the "gate current" question, so I simply did it, constructing a "gate current amplifier":  The photodiode was reverse-biased with the minute leakage, dramatically reducing its capacitance, and photoconducting currents being sinked by the gate-source junction.  As expected, the drain current increased noticeably, but the circuit worked extremely well, with both frequency response and apparent gain increasing dramatically!

Putting this "new" circuit back on the photon range I observed that although its low frequency (<200 Hz) sensitivity was slightly worse than that of the K3PGP circuit (see comment below), the higher speech-range frequencies (300-2500 Hz) were, on average, 10-12dB better than the VK7MJ circuit and approximately 20 dB better than the best, low-noise op-amp based TIA circuit that I'd built to date!

In analyzing the circuit, there are several things happening:

• Reverse bias of the photodiode:  This reduces the capacitance - typically by a factor of 3-6, depending on the specific device and voltage applied.
• The photodiode will produce current in the presence of light.
• Being reverse-biased, the photodiode will also operate in a photo-conductive mode, passing current from the bias supply in response to light.
• With the gate-source junction conducting, the reverse bias across the photodiode is maintained since the gate-source voltage will never exceed 0.4-0.6 volts.
• As described above, the amplifier is connected in "cascode" configuration to minimize Miller effects.
• There are NO other components or signal paths connected to the photodiode-gate junction that can contribute noise or attenuate the signals.
• In parallel with the cascode circuit is a current source which provides a high-impedance current source to increase the JFET's bulk currents, further reducing its noise.

 About the gate-source conductivity of the JFET, two things surprised me:

• The "diode action" of the gate-source clamping seems not to be a significant contributor of noise - at least at "dark" currents of the photodiode.
• There is little or no documentation about using a JFET this way, anywhere else!

It is likely that the main reason that this doesn't perform quite as well at the K3PGP circuit at low (<200 Hz) frequencies is because of the intrinsic leakage current noise endemic to the reverse biasing of the photodiode, particularly in a "1/F" manner:  At higher frequencies where this sort of noise falls away it performed far better. 

Note:

In "photon range" testing it was difficult to tell at which frequencies the K3PGP receiver performed better.  My K3PGP exemplar receiver was certainly better at, say, 20 Hz, but even at 100 Hz or 60 Hz it was a difficult call to make.  At such frequencies and under such conditions careful selection of the "quietest" photodiode and FET can make a significant difference and with most of these circuits, reducing their temperature - while somehow managing to avoid condensation - can help even more!

Plotting Gate current versus Drain and Gate voltages:

Later, I constructed a test fixture to analyze the gate-source voltage and gate-source current response of a 2N5457 JFET and plot this against the drain current - see Figure 6 below.

Figure 6:
 Gate-source voltage and Gate current plotted against drain current for a typical, real-life JFET - not a simulation!  Note the logarithmic scale of the gate current and also that the drain current continues to increase linearly with gate-source voltage, even after the gate-source junction is conducting.
Click on the image for a larger version.

As can be seen, as the gate-source voltage increases, the drain increases linearly - even after the gate-source diode junction starts to conduct:  In fact, there does not appear to be inflection of the drain current curve when this happens!  Following the other line representing gate current we can see that once our gate-source "diode" starts to conduct, the gate current follows the classic logarithmic curve that one associates with diodes - which should not come as a surprise.

Equation 1:
The relationship between drain current and
gate-source voltage.
Vgs= Gate-source voltage
Vp=FET Pinch-off voltage
Idss=Zero gate voltage drain current

According to typical JFET models, in the saturation region the FET operates such that the drain current is generally independent of the drain voltage as can be seen in Equation 1 and the graphs in Figure 6 indicate that this seems to be true even when the gate-source junction is conducting.

So, now we know what is happening.  At first glance, one might presume that with this diode in conduction that the logarithmic response would make the circuit unsuitable for general audio recovery - but this is not so:  At very low light levels the detector has lower than 1% harmonic distortion.

Figure 7:
Test circuit used to derive the curves in Figure 6.
For measuring the voltage at "Vgate Monitor" it will be
required that the negative lead of the voltmeter be referenced
to a regulated, negative (with respect to ground)
voltage source.  Q1 is the device being tested and
Q2 is just another JFET which need not be the
same type as J1.

In case you are interested, Figure 7 shows the circuit that was used to derive the curves in Figure 6, above.  10.0 volts was used for V+ and the drop across source-follower Q2 was easily characterized so that the drop across R1 - and thus the gate current in Q1 - could be determined.  The drain current was determined by measuring the voltage across R2.  Different values of R1 were used to achieve the measurement range depicted in Figure 6 which accounts for the very slight bend in the "Gate Current" curve.

Putting this into practice:

The circuit depicted in Figure 8 was developed for speech-bandwidth optical communications use.

As can be seen, this looks very similar to the circuit of Figure 4 with the exception that the reverse-biased photodiode is connected to the JFET and that there is the added circuit, U1b, that forms a bandwidth-limited differentiator - the component values chosen to approximately correlate with the low-frequency "knee" of the BPW34 photodiode and also to cease its frequency boost above 5-8 kHz.  (The "Flat" audio output, uncompensated by the differentiator for the 6dB/octave rolloff, is provided for both very low frequency - below 200 Hz - and high frequency - above 5 kHz - signals to be applied to a computer for analysis.)

The circuit in Figure 8 - and minor variations of it - have been replicated many times over the years using different components.  The important considerations are that both Q2 and Q3 be low-noise, high-beta transistors such as the MPSA18 (or 2N5089) and that the JFET used for Q1 be capable of rather high drain current.  In the original design, the 2N5457 was specified as this device is better-characterized that many other, similar FETs and is capable of quite low-noise operation:  The more common MPF102, with its extremely wide variation of parameters, might be suitable if an appropriate device is "cherry picked" from amongst several based both on high zero gate-source voltage drain current and tested "noisiness".  A more modern JFET is the BF862 - available in surface-mount only (as are most JFETs these days!) - that is even better for this application than the 2N5457 and capable of much higher drain ("bulk") current to the point where utilizing its full potential might compromise 9-volt battery life!

Figure 8:  
Version "3" of the optical receiver.  This receiver must always be operated on its own, completely isolated power supply to avoid feedback.  V+ is 8-15 volts and is typically a 9-volt battery.  D4 and TH1 prevent damage should the applied polarity of the power source be accidentally reversed.  After Q1's drain current has been measured and adjusted, jumper "J1" is closed.
A version of this circuit by the author of this page also appeared in an article published in the SPIE proceedings (#6878) which was presented at the 2008 "Photonics West" conference by another one of the paper's co-authors, Chris Long.
Click on the image for a larger version.

In a circuit such as Figure 8, above, the drain-source voltage will be much lower than one might initially expect - on the order of 0.2-1.0 volts for a JFET such as a 2N5457 and between 0.1 and 0.5 volts for the BF862 - but this is normal operation.  While the setting for Q3 current, adjusted via R5, (in Figure 8) at 120 ohms is suitable for most 2N5457 devices, the current may need to be reduced (e.g. R5 increased in value to 180 or 220 ohms) for some "lower 0 Vgs" current devices such as the MPF102.  In general, the higher the drain current, the lower noise contribution from the FET - but if you exceed the "magic" value and attempt to force too much current, the circuit will suddenly stop working:  Overall it is better to have a bit lower drain current than optimal and have a little bit more noise than to have too much drain current!  (Don't forget that the properties of the current source and the JFET itself will also change with temperature - but they generally seem to track.)

Interestingly, the circuit depicted in Figure 8 also works in daylight, albeit with some caveats.

When very high levels of light are present, the photoconductivity will shunt the reverse bias to the gate-source junction, and the frequency rolloff "knee" associated with the photodiode capacitance will shift upwards due to photoconductive shunting causing the audio to become "tinny".  The audio will also become somewhat distorted owing to the different light-to-audio transfer curve that occurs under such conditions, in which case the frequency response of the audio on the "flat" output is more suitable than otherwise.  In such situations one does not really need the high sensitivity of this type of receiver, anyway, and a typical TIA circuit with AC coupling such as that depicted in Figure 2 or Figure 3 could be used or one could apply optical attenuation in front of the detector to reduce the light level.

Practical use:

Figure 8:
An as-built "Version 3" optical receiver, constructing using
prototyping techniques and enclosed in a shielded, light-tight
enclosure using pieces of printed circuit board material.  For this
unit "feedthrough" capacitors are used for power and audio
connections to prevent the incursion of RF energy on
the connecting leads.
Click on the image for a larger version.

Entire web pages could be written (and have been - see the Modulated Light web site - link) about through-the-air, free-space optical communications over long distances (well over 100 miles, 160km) using both LEDs and low-power lasers, but even the most sensitive receiver - no matter the underlying technology - requires supporting optics (lenses!) in order to function properly:  It is through such lenses that 10's of dB of noiseless signal gain may be achieved, not to mention directionality and the implied rejection of off-axis light sources.

The circuits described on this page are likely to be suitable only for speech frequencies and low-rate data but this is, in part, due to the medium involved (the atmosphere) and method of transmission.  At the extreme distances that have been achieved with the above equipment (>173 miles, 278km) the signals are weak enough that only low-rate signalling techniques would likely be feasible under typical conditions at safe, practical optical power levels.

Additional web pages on related topics:

• Modulatedlight.org - This web site has a wide variety of information related to amateur, free-space optical, through-the-air communications.
• Optical Receivers for Low-Bandwidth, Through-the-Air Communications - This is a reference article that gives additional detail about the design of the circuits discussed in this article.
• Using Laser Pointers for Free-Space Optical Communications - This describes how one might use low-power laser pointers for low-rate optical communications, the practicalities of various circuits, the methods and the realities.
• The Modulated Light DX page - This page has several articles describing the practical aspects of free-space, through-the-air optical communications including various atmospheric effects.
• A description of this circuit appeared in the SPIE Conference Proceedings, Volume 6878, “Atmospheric Propagation of Electromagnetic Waves II” in the article "Dollars versus Decibels:  Long-Range atmospheric optical communications on a tight budget".  This article was presented at the January, 2008 Photonics West conference and a copy of this article may be read here.

The above web pages also contain links to other, related pages on similar subjects.


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This page stolen from "ka7oei.blogspot.com".