Thursday, November 25, 2021

Fixing the CAT Systems DL-1000 and AD-1000 repeater audio delay boards

Figure 1:
The older DL-1000 (top) and the newer
AD-1000, both after modification.
Click on the image for a larger version.

A few weeks ago I was helping one of the local ham clubs go through their repeaters, the main goal being to equalize audio levels between the input and output to make them as "transparent" as possible - pretty much a matter of adjusting the gain and deviation appropriately using test equipment.  Another task was to determine the causes of noises in the audio paths and other anomalies which were apparent to a degree at all of the sites.

All of the repeater sites in question use CAT-1000 repeater controllers equipped with audio delay boards to help suppress the "squelch noise" and to ameliorate the audio delay resulting from the slow response of a subaudible tone decoder.  Between the sites, I ran across the older DL-1000 and the newer AD-1000 - but all of these boards had "strange" issues.

The DL-1000:

This board uses the MX609 CVSD codec chip which turns audio into a single-bit serial stream at 64 kbps using a 4-bit encoding algorithm, which is then fed into a CY7C187-15 64k x 1 bit RAM.  To adjust the amount of delay in a binary-weighted fashion, a set of DIP switches are used to select how much of this RAM is used by enabling/disabling the higher-order address bits, feeding the RAM's contents back into the chip to be decoded back to audio.

The problem:

It was noticed that the audio from the repeater had a bit of an odd background noise - almost a squeal, much like an amplifier stage that is on the verge of oscillation.  For the most part, this odd audio property went unnoticed, but if an "A/B" comparison was done between the audio input and output - or if one inputted a full-quieting, unmodulated carrier, this strange distortion could be heard.

Figure 2:
The location of C5 on the DL-1000.  A 0.56 uF capacitor was
used to replace the original 0.1 (I had more of those than
I had 0.47's)
and either one would probably have been fome
As noted below, I added another to the bottom of the board.
Click on the image for a larger version.

This issue was most apparent when a 1 kHz tone was modulated on a test carrier and strange mixing products could be heard in the form of a definite "warble" or "rumble" in the background, superimposed on the tone. Wielding an oscilloscope, it was apparent that there was a low-frequency "hitchhiker" on the sine wave coming out of the delay board that wasn't present on the input - probably the frequency of the low-level "squeal" mixing with the 1 kHz tone.  Because of the late hour - and because we were standing in a cold building atop a mountain ridge - we didn't really have time to do a full diagnosis, so we simply pulled the board, bypassing the delay audio pins with a jumper.

On the workbench, using a signal tracer, I observed the strange "almost oscillation" on pin 10 of the MX609 - the audio input - but not on pin 7 of U7B, the op-amp driver, implying something amiss with the coupling capacitor - a 0.1uF plastic unit, C5.  Because these capacitors almost never fail - particularly with low-level audio circuits - I suspected something fishy and checked the MX609's data sheet and noted that it said "The source impedance should be less than 100 ohms.  Output channel noise levels will improve with an even lower impedance."  What struck me was that with a coupling capacitor of just 0.1uF, this 100 ohm recommendation would be violated at frequencies below 16 kHz - hardly adequate for voice frequencies!

Figure 3:
The added 2.2uF tantalum capacitor on the bottom of
the board across C5.  The positive side goes toward
the MX609, which is on the right.
Click on the image for a larger version.

Initially, I bridged C5 with a 0.1uF plastic unit and the audible squealing almost completely disappeared.  I then bridged C5 it with a 0.47uF capacitor which squashed the squealing sound and moved the 100 ohm point to around 4 kHz, so I replaced C5 with a 0.56uF capacitor - mainly because I had more of those than small 0.47uF units.

Not entirely satisfied, I bridged C5 with a 10uF electrolytic capacitor, moving the 100 ohm impedance point down to around 160 Hz - a frequency that is below the nominal frequency response of the audio channel - and it caused a minor, but obvious quieting of the remaining noise, particularly at very low audio frequencies (e.g. the "hiss" sounded distinctly "smoother".)   Because I had plenty of them on-hand, I settled on a 2.2 uF tantalum capacitor (100 ohms at 723 Hz) - the positive side toward U2 and tacked to the bottom of side of the board - which gave a result audibly indistinguishable from 10 uF.  In this location, a good-quality electrolytic of 6.3 volts or higher would probably work as well, but for small-signal applications like this a tantalum is an excellent choice, particularly in harsh temperature environments.

At this point I'll note that any added capacitance should NOT be done with ceramic units.  Typical ceramic capacitors in the 0.1uF range or higher are of the "Z5U" type or similar and their capacitance changes wildly with temperature meaning that extremes may cause the added capacitance to effectively "go away" and the squealing noise may return under those conditions.  Incidentally, these types of ceramic capacitors can also be microphonic, but unless you have strapped your repeater controller to an engine, that's probably not important.

Were I to do this to another board I would simply tack a small tantalum capacitor - anything from 1 to 10 uF, rated for 6 volts or more - on the bottom side of the board, across the still-installed, original C5 (as depicted in Figure 3) with the positive side of the capacitor toward U2, the MX609.


One of the repeater sites also had a "DL-1000A" delay board - apparently a later revision of the DL-1000.  A very slight amount of the "almost oscillation" was noted on the audio output of this delay board, too, but between its low level and having limited time on site, we didn't investigate further. 
This board appears to be similar to the DL-1000 in that it has many of the same chips - including the CY7187 RAM, but it doesn't have a socketed MX609 on the top of the board, and likely a surface-mount codec on the bottom.  It is unknown if this is a revision of the original DL-1000 or closer to the DL-1000C which has a TP4057 - a codec functionally similar to the MX609.

The question arises as to why this modification might be necessary?   Clearly, the designers of this board didn't pay close enough attention to the data sheet of the MX609 codec otherwise they would have probably fitted C5 with a larger value - 0.47 or 1 uF would have probably been "good enough".  I suspect that there are enough variations of the MX609 - and that the level of this instability - is low enough that it would largely go unnoticed by most, but to my critical ears it was quite apparent when an A/B comparison was done when the repeater was passing a full-quieting, unmodulated carrier and made very apparent when a 1 kHz tone was applied.

* * * * * * * * * * * * * * *

The AD-1000:

This is a newer variant of the delay board that includes audio gating and it uses a PT2399, a chip commonly used for audio echo/delay effects in guitars pedals and other musical instrument accessories as it has an integrated audio delay chip that includes 44 kbits of internal RAM.

The problems:

This delay board had two problems:  An obvious audio "squeal", very similar to that on the older DL-1000, but extremely audible, but there was a less obvious problem - something that sounded like "wow" and flutter of an old record on a broken turntable in that the pitch of the audio through the repeater would warble randomly.  This problem wasn't immediately obvious on speech, but this pitch variation pretty much corrupted any DTMF signalling that one attempted to pass through the system making the remote control of links and other repeater functions difficult.

RF Susceptibility:

Figure 4:
The top of the modified AD-1000 board where the
added 1k resistor is shown between C11/R13 and
pin 2 of the connector, the board trace being severed.
Near the upper-right is R14, replaced with a 10 ohm resistor,
but simply jumpering this resistor with a blob of solder
would likely have been fine.
Click on the image for a larger version.
This board, too, was pulled from the site and put on the bench.  There, the squealing problem did not occur - but this was not unexpected:  The repeater site is in the near field of a fairly powerful FM broadcast and high-power public safety transmitters and it was noticed that the squealing changed based on wire dressing and by moving one's hand near the circuit board.  This, of course, wasn't easy to recreate on the bench, so I decided to take a look at the board itself to see if there were obvious opportunities to improve the situation.

Tracing the audio input, it passes through C1, a decoupling capacitor, and then R2, a 10k resistor - and this type of series resistance generally provides pretty good resistance to RF ingress, mainly because a 10k resistor like this has several k-ohms of impedance - even at VHF+ RF frequencies, which is far higher than any piece of ferrite material would provide!

The audio output was another story:  R13, another 10k resistor, is across the output to discharge any DC that might be there, but the audio then goes through C11, directly to pin 1 of U2, the output of an op-amp.  While this may be common practice under "normal" textbook circumstances, sending the audio out from an op-amp into a "hostile" environment must be done with care:  The coupling capacitor will simply pass any stray RF - such as that from a transmitter - into the op amp's circuitry, where it can cause havoc by interfering/biasing various junctions and upsetting circuit balance.  Additionally, having just a capacitor on the output of an op amp can be a hazard if there also happens to be an external RF decoupling capacitor - or simply a lot of stray capacitance (such as a long audio cable) as this can lead to amplifier instability - all issues that anyone who has ever designed with an op amp should know!

Figure 5:
The added 1000pF cap on the audio gating lead.
A surface-mount capacitor is shown, soldered to the
ground plane on the bottom of the board, but a small disk-
ceramic of between 470 and 1000 pF would likely be fine.
Click on the image for a larger version.
An easy "fix" for this, shown in Figure 4, is simply to insert some resistance on the output lead, so I cut the board trace between the junction of C11/R13 and connector P1 and placed a 1k resistor between these two points:  This will not only add about 1k of impedance at RF, but it will decouple the output of op amp U2 from any destabilizing capacitive loading that might be present elsewhere in the circuit.  Because C11, the audio output coupling capacitor is just 0.1uF, the expected load impedance in the repeater controller is going to be quite high, so the extra 1k series resistance should be transparent.

Although not expected to be a problem, a 1000pF chip cap was also installed between the COS (audio gate) pin (pin 5) and ground - just in case RF was propagating into the audio path via this control line - this modification being depicted in Figure 5.

Of course, it will take another site visit to reinstall the board to determine if it is still being affected by the RF field and take any further action.

And no, the irony of a repeater's audio circuitry being adversely affected by RF is not lost on me!

 The "wow" issue:

On the bench I recreated the "wow" problem by feeding a tone into the board, causing the pitch to "bend" briefly as the level was changed, indicating that the clock oscillator for the delay was unstable as the sample frequency was changing between the time the audio entered and exited the RAM in the delay chip.  Consulting the data sheet for the PT2399 I noted that its operating voltage was nominally 5 volts, with a minimum of 4.5 volts - but the chip was being supplied with about 3.4 volts - and this changed slightly as the audio level also changed.  Doing a bit of reverse-engineering, I noted that U4, a 78L05, provided 5 volts to the majority of the circuit, but the power for U3, the PT2399, was supplied via R14 - a 100 ohm resistor:  With a nominal current consumption of around 15 milliamps, this explained the 1.6 volt drop.

The output at resistor R14 is bypassed with C14, a 3.3 uF tantalum capacitor, likely to provide a "clean" 5 volt supply to decouple U14's supply from the rest of the circuit - but 100 ohms is clearly too much for 15 mA.  While testing, I bridged (shorted) R14 and the audio frequency shifting stopped with no obvious increase in background noise, so simply removing and shorting across R14 is likely to be an effective field repair, but because I had some on hand, I replaced R14 with a 10 ohm resistor as depicted in Figure 4 and the resulting voltage drop is only a bit more than 100 millivolts, but retaining a modicum of power supply decoupling and maintaining stability of the delay line.

Figure 6:
Schematic of the AD-1000, drawn by inspection and with the aid of the PT2399 data sheet.
Click on the image for a larger version.

Figure 6, above, is a schematic drawn by inspection of an AD-1000 board with parts values supplied by the manual for the AD-1000.  As for a circuit description, the implementation of the PT2399 delay chip is straight from the data sheet, adding a dual op-amp (U2) for both input and output audio buffering and  U1, a 4053 MUX, along with Q1 and components, were added to implement an audio gate triggered by the COS line.

As can be seen, all active circuits - the op-amp, the mux chip and delay line - are powered via R14 and suffer the aforementioned voltage drop, explaining why the the supply voltage to U3 varied with audio content, causing instability in audio frequencies and difficulty in decoding DTMF tones passed through this board - and why, if you have one of these boards, you should make the recommended change to R14!


What about the "wow" issue?  I'm really surprised that the value of R14 was chosen so badly.  Giving the designers the benefit of the doubt, I'll ignore the possibility of inattention and chalk this mistake, instead, to accidentally using a 100 ohm resistor instead of a 10 ohms resistor - something that might have happened at the board assembly house rather than being part of the original design. 

After a bit of digging around online I found the manual for the AD-1000 (found here) which includes a parts list (but not a schematic) that shows a value of 100 ohms for R14, so no, the original designers got it wrong from the beginning!

While the RF susceptibility issue will have to wait until another trip to the site to determine if more mitigation (e.g. addition of ferrite beads on the leads, additional bypass capacitance, etc.) is required, the other major problems - the audio instability on the DL-1000 and the "wow" issue on the AD-1000 have been solved.

* * * * * * * * * * * * * * *

Comments about delay boards in general:

  • Audio delay boards using the PT2399 are common on EvilBay, so it would be trivial to retrofit an existing CAT controller with one of these inexpensive "audio effects" boards to add/replace a delay board - the only changes being a means of mechanically mounting the new board and, possibly, the need to regulate the controller's 12 volt supply down to whatever voltage the "new" board might require..  The AD-1000 has, unlike its predecessor, an audio mute pin which, if needed at all, could be accommodated by simple external circuitry.
  • In bench testing, the PT2399 delay board is very quiet compared the MX609 delay board - the former having a rated signal-noise ratio of around 90 dB (I could easily believe 70+ dB) while the latter, being based on a lossy, single-bit codec, has a signal-noise ratio of around 45 dB - about the same as you'd get with a PCM audio signal path where 8 bit A/D and D/A converters were being used.

A signal/noise ratio of around 45 dB is on par with a "full quieting" signal on a typical narrowband communications radio link so the lower S/N ratio of the MX609 as compared with the PT2399 would likely go unnoticed.  Were I to implement a repeater system with these delay boards I would preferentially locate the MX609-based delay boards in locations where the noise contribution would be minimized (e.g. the input of the local repeater) while placing the quieter PT2399-based board in signal paths - such as a linked system - where one might end up with multiple, cascaded delay lines on link radios as the audio propagates through the system.  Practically speaking, it's likely that only the person with a combination of a critical ear and OCD is likely to even notice the difference!

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Friday, October 29, 2021

Quieting a Samlex 150 watt Sine Wave inverter

A few weeks ago I was on vacation in remote Eastern Utah - in Canyonlands National Park, to be precise and because we had some "down time" in the evenings, after hiking, after sunset, I was able to set up a portable HF station.  Using the homebrew end-fed halfwave antenna (EFHW) of Mike, K7DOU - one end of the rope tied around a rock laying on a shelf of slick rock some 40 feet above ground level and the other end tied to a bamboo pole attached to my Jeep - I connected my FT-100 through a manual tuner as the VSWR of the EFHW wasn't necessarily very low on some of the higher bands.

Figure 1:
150 Watt Samlex sine wave inverter, sitting on the workbench.
Click on the image for a larger version.

For whatever reason, I had brought along my old lap top and sound-card interface so I could work some digital modes, specifically FT-8 - a mode that I was familiar with, but had personally never worked.  The battery in my laptop had discharged, so I needed an alternate source of power and I connected my 150 watt Samlex Sine Wave inverter (a PST-15S-12A) to the battery to power the computer's power supply.

The (expected!) result of this was a tremendous "hash" all across the HF spectrum - an obvious result of the various high-power converters contained within the inverter.  On some bands the interference wasn't too bad, but on others the result was unusable.  While the battery charged, I operated on the band (20 meters, IIRC) that wasn't as badly affected.

I left the inverter running and the laptop battery charging during the cooking and eating of dinner, and with a reasonable amount of power banked I could turn off the inverter and get a zero noise floor while operating.

Why so noisy?

Modern AC inverters first convert the DC input power to something around the peak voltage found on the AC output - typically around 155 volts for 120 volt mains.  This conversion is done using a switch-mode inverter with a transformer, typically operating in the 20-60 kHz range and this output is rather rich in harmonics.

For the less-expensive "modified sine wave" inverters, the DC output is chopped, typically using an "H" bridge switch using FETs (Field Effect Transistors) with the duty cycle being varied to provide the equivalent of a 120 volt sine wave - and this switching can also add a bit of extra RFI, most notably in the form of a "buzz" - but this action produces less energy at radio frequencies than the initial voltage conversion.

The "Sine Wave" inverters perform the same step of producing the high DC voltage, but will chop the output into much smaller bits.  The method that this is done can vary, but it's sometimes done by using a "buck" type switching converter to transform the higher voltage into a varying - usually lower - voltage to simulate a sine wave on the output.  This second conversion adds yet another source of RF interference atop what is likely already the significant source that already present in the high voltage converter.

Comment:  The power converter (wall wart) that I was using to charge my laptop is particularly quiet, so I did verify that the vast majority of noise was, in fact, from the AC inverter.

Figure 2:
Various mains filtering components:  All of these are bifilar,
common-mode chokes, except for that in the upper-left with is
a combination filter and IEC power connector.
Click on the image for a larger version.

Quieting the inverter:

Fortunately, the internal space of this inverter wasn't terribly cramped so there was just enough room to add the necessary components to suppress the RF "hash" that was being conveyed on both the DC and AC lines.  While the methods of doing this sort of RF quieting have been discussed in previous blog posts (see the references at the end of this article) I'll review them in detail here.

It's worth noting (several times!) that simply winding the power cord (DC and/or AC) around a ferrite device (e.g. a clamp-on or even a large toroid) would likely NOT be enough to solve this problem.  While doing so may knock down RFI by, perhaps, 6-10 dB - maybe 20 dB if one is really lucky - this sort of noise egress must be attenuated by 10s of dB to effectively quash it.  In other words, knocking down the "grunge" by 1-2 S-units is nice enough, but there will still be a lot of hash left over to bury the weakest signals! 

Internally, this inverter did pass through some rather large ferrite cylinders the DC input and (separately) AC output connections, but this very small amount of inductance would have practically no effect at all at HF - likely having been added to make a dent in the noise at VHF so that it would pass muster when subjected to EMC compliance tests.

Filtering the AC output:

I presumed (but didn't actually measure) that the majority of the noise being radiated would be from the AC output as it is "closest" to the circuits most likely to generate a lot of noise, so I concentrated most of my effort there.

The most helpful component in filtering the mains voltage output is the bifilar choke - several varieties of these being displayed in Figure 2.  This component consists of two windings in parallel on the same ferrite core - typically both leads of the mains voltage.  For the low-frequency AC currents, the halves of the choke carry equal and opposite current so there is no DC component to magnetize the core and reduce its efficacy due to saturation, but because RF energy is likely not flowing in a differential manner as is the AC mains voltage, the inductance of the two parallel windings come into effect - the magnitude of this typically being in the 10s of microHenries to milliHenries range.

Where does one get these things?  They can be found at surplus outlets if you look around, but perhaps the easiest source is from defunct PC power supplies:  These devices, found in supplies made by reputable manufacturers, are typically the first things through which the AC mains voltage pass (after any fusing) before going to the rest of the circuitry.

Figure 3:
Schematic of the output filter.  While it's likely that just one bifilar inductor would have sufficed, I decided that since there was room to do so, a second one would be added for even more filtering of the "grunge" that can emanate from such a noisy circuit.
Click on the image for a larger version.

This much inductance has significant impedance to RF energy - but inductance alone will have only limited efficacy and intrinsic capacitance of the windings will also reduce the amount of attenuation that would otherwise happen - as would have winding the mains cord/cable on a ferrite toroidal core as noted previously - so capacitors are also required to be placed strategically to help shunt away some of the residue.

Figure 4:
The AC output filter in the process of being installed.  L1 and
C1-C4 are mounted to the outlet itself while the connection
to L2 is made using the orange leads.
Click on the image for a larger version.

The diagram in Figure 3 shows the as-installed filter.  As can be seen, two separate bifilar filters (both of them being the sort as seen as the second from the lower-right in Figure 2) were used to maximize attenuation.  In this circuit, C3 and C4 are used to force any RF on the two wires to be common-mode to maximize the efficacy of the bifilar chokes' attenuation and any residual RF - which will be at rather low level and high impedance - will then be shunted to the metal case of the inverter by capacitors C1 and C2.

Figure 4 shows the installation of the filtering components in the inverter.  C1 and C2 are the disk-shaped blue capacitors seen in the upper-left, mounted directly to the inverter's single AC outlet and capacitor C3 is just in "front" of the two round disks, also mounted directly to the socket.  The first inductor, L1, can be seen in the shadows, connected to the outlet with very short, flexible leads to the plug.

Earlier, I had removed this outlet from the body of the inverter and mounted C1, C2, C3 and L1 to it and with a bit of "tetris" action, was able to reinstall the outlet back in place with the components attached.  From that point I installed C4 (to the "other" side of L1) and the (orange) connecting wires from C4 to L2, which is shown floating in space.

You might ask why there isn't another capacitor (like C4) across the "inverter" side of L2 - or other capacitors to ground other than C1/C2:  There is already a degree of filtering on the AC output of the inverter, so there is little point in adding another capacitor like C4.  As for other capacitors to "ground" like C1/C2 elsewhere in the circuitry:  These were deemed unnecessary - and doing so, particularly at the "inverter" side of L4 would simply put relatively strong RF currents onto the ground lead (e.g. inverter's case) - and our cause won't be helped in making RF currents appear we don't need them to be.  

Figure 5:
Noise filter on the DC input.  It looks suspiciously like the filter on the AC output - because it's the same type, although the current-carrying capacity of L1 is much higher and the values of the capacitors are orders of magnitude larger.
Click on the image for a larger version.

Filtering the DC input:

While I would presume that most of the noise would be emitted via the AC output port, filtering the DC port must be considered as well.  With the inverter's rating being 150 watts, the maximum current on the AC output would be around 1.25 amps and rather light-gauge wire could be used in the inductors - but because this same power level represents 12.5 amps at 12 volts (likely more if the battery voltage is on the low side) the filtering inductance must be made using much larger wire.

Rummaging around in my box of toroids, I found a ferrite device that was about 1" (2.54cm) in outside diameter and wound as many turns of 14 AWG flexible wire onto it as would fit (about 6 bifilar turns) and measured it to have about 30 uH of inductance per winding.  This may not seem like much, but at 1 MHz, this represents about 180 ohms of reactance.   

In referring to Figure 5, above, you'll notice that it is pretty much identical to that of the output filter - except that there is only one section of filtering.  The capacitor values are different, too:  C1 and C2 are 0.1uF units that shunt residual RF getting through L1 to ground (the case) while C3 is a low-ESR electrolytic connected across the DC leads to help force any residual AC noise on the DC lead to common-mode.  Compared to the 180 ohms of reactance of the DC bifilar choke (at 1 MHz) a good-quality, monolithic ceramic capacitor like the 0.1uF units are likely to have well under an ohm of impedance and very little of the RF hash will remain after they do their job to bypass it to the chassis ground.

Figure 6:
The DC input filter.  The capacitors (not visible) are mounted
to the bottom side of the terminal strip, which serves as the
RF "grounding" point to the case.  L1 is just visible.
Click on the image for a larger version.

Because of the limited amount of room, only one inductor was used - although it would likely be possible to have crammed another in the limited space should the above filter have proved to be inadequate (it wasn't).

As can be seen in Figure 6, a small terminal strip is visible and to it is mounted C1-C3 (not visible as they are obscured by the strip itself).  The mounting point for this strip is the ground lug near the DC input cable and the center lug is the common point for C1 and C2.

An important point to mention is the fact that this inverter - like many - have their DC and AC lines isolated from the case - and that's also important here:  Because the DC has no connection to the inverter's metal case, ALL of the DC current passes through L1 of Figure 5 - but with both halves carrying the same current, the core is not magnetized:  Magnetizing the core would likely cause it to saturate and the result would be its effective inductance plummeting - possibly reducing its efficacy as an RF filter.  It is for this reason that a bifilar choke was used on the DC input as well.

As with the AC output, the "inverter" side of L1 of Figure 5 also lacks a common-mode capacitor, but this is well represented on the input of the inverter itself with its own, built-in capacitor.

Figure 7:
The final arrangement of the added filtering components.  Liberal use of RTV (silicone adhesive) was used to stabilize the components as it works well, and can be removed should repairs/modifications be required.  On the left, a generous blob of RTV has been used to keep the terminal strip's lugs at the DC input from touching the inverter's bottom cover.
Click on the image for a larger version.

Additional comments:

Figure 7 shows the final arrangement of the added components.  In the upper-left corner can be seen the components of the DC input filter with come clear RTV (silicone adhesive) added to the top of the terminal strip to insulate it and keep any metal parts of it from touching the bottom cover when it was reinstalled.

On the right side is the AC output filter.  In the foreground can be seen L2, now with the "hot" terminals covered by heat-shrink tubing.  This choke was first attached "temporarily" to the inverter's end plate using instant (cyanoacrylate) glue - and then several large blobs of RTV were later added to permanently hold it in place.  Just above it can be seen the orange wires that connect L2 to L1 and these components were also stabilized with rather large blobs of RTV to keep them from "flapping in the breeze".  It's worth noticing that the original ferrite cylinder is still on the AC output connection (on the black and white wires) where it connects to L4 - mainly because there was still room for it, and its efficacy, such as it is, is likely only enhanced by the addition of the new filtering components. 

Did it work?

You might ask the question:  Did this filtering work?

The answer is yes.  Placing a portable shortwave radio next to either the DC or AC power leads from the inverter, one can't detect that it is running at all.  If the radio is placed right atop the inverter, some hash can be detected, but this is likely from direct radiation of magnetic fields from the inductors/transformers within, but detectable amounts do not appear to be emanating from DC and AC wires themselves - and that's the important part as they would otherwise be acting as antennas.

Perhaps the most important part of this modification is the fact that any bypass capacitors are placed on the "quiet" (not the inverter) side of the filtering inductances and that these bypass capacitors are connected, with short leads, to a large, common-point ground - namely the case of the inverter.  If any of the "ground" leads had been more than an inch or two long, it's likely that the impedance of it would have reduced the efficacy of the filtering - but the case, being a solid chunk of extruded aluminum, forms a nice, low-impedance tie point - effectively a single-point ground, preventing an RF current differential between the DC input and AC output leads.

* * *

Links to other articles about power supply noise reduction found at


This page stolen from


Tuesday, September 28, 2021

Pink bits of rubber causing a blinking light...

 A bit more than a week ago I volunteered for an aid station along the route of the Wasatch 100 mile endurance run - which, as the name implies, is a 100 mile race, starting and ending some distance apart in Northern Utah.  This year, I was asked to be near-ish the start of the race, about 20.9 miles (30.4 km) from the start at a location in the mountains, above the Salt Lake Valley - a place that required the use of a high-clearance and somewhat rugged vehicle - such as my 2017 Jeep Rubicon.

Figure 1:
The blinking "Sway Bar" light - not something that you
want to see when you have shifted out of four-wheel drive!
Click on the image for a larger version.

Loaded with several hundred pounds of "stuff" I went up there, bouncing over the rough roads and despite enduring several bouts of rain, hail, lightning and thunder, managed to do what needed to be done in support of the race and runners and headed down.

Because of the rather rough road, I decided to push the button marked "Sway Bar" that disconnects the front left and right front tires from each other, allowing more independent vertical travel of each wheel, making the ride smoother and somewhat improving handing over the rougher parts.  Everything went fine until - on the return trip, near the bottom of the unimproved portion of the mountain road, I pushed the button again and...  the light kept blinking, on for a second and off for a second - and a couple minutes later, it started blinking twice as fast, letting me know that it wasn't "happy".

"What's the problem with that?"

Pretty much all modern road vehicles have a sway bar - or something analogous to it - that couple the vertical travel of the wheels on the same axle together to reduce body roll, which improves handling as one makes a turn - particularly around corners.  At low speeds, such roll isn't too consequential, but at high speeds excess roll can result in... well... "problems" - which is why I was a bit apprehensive as I re-entered the city streets.

Knowing that this type of vehicle is known for "issues" with the sway bar disconnect, I did the normal things:  Pushed the button on and off while rocking the vehicle back and forth (while parked, of course!), stopped and restarted the engine - and even pulled the fuse for the sway bar and put it back in - all things suggested online, but nothing seemed to work.

Stopping at a parking lot and crawling under the front of the vehicle while someone else rocked it back and forth did verify one thing:  Despite the indicator on the dashboard telling me that the sway bar wasn't fully engaged, I could see that it was, in fact, locked together as it should be as evidenced by the fact that the two halves of the bar seemed to move together with the vehicle's motion - so at least I wasn't going to have to drive gingerly back on the freeway.

Fixing the problem:

Figure 2:
Sway bar and disconnect mechanism, removed from the
vehicle with the lead screw/motor in the upper-right.
Click on the image for a larger version.
As mentioned before, this is a common problem with this type of vehicle and online, you will find lots of stories and suggestions as to what might be done.  Quite a few people just ignore it, others have it fixed under warranty - but those that have vehicles out of warranty seem to mostly retrofit it with a manual disconnect, if they care about the sway bar at all.

The reasons for the issue seem to be various:  Being an electromechanical part that is outside the vehicle, it's subject to the harsh environment of the road.  Particularly in the case of some die-hard Jeepers (of which I'm not particularly, although I've made very good use of its rough and off-road capabilities) reports online indicate that it is particularly prone to degradation/contamination if one frequently fords rivers and spends lots of time in the mud:  Moisture and dirt can ingress the mechanism and cause all sorts of things to go wrong.

Fortunately, one can also find online a few web pages and videos about this mechanism, so it wasn't with too much trepidation that, a week after the event - when I was going to change the oil, filters and rotate the tires anyway - I put the front of the vehicle on jack stands and removed the sway bar assembly entirely.  This task wasn't too hard, as it consisted of:

  • Remove the air dam.  My vehicle had easily removable plastic pins that partially popped apart with the persuasion of two screwdrivers - and there are only eight of these pins.
  • Disconnect the wire.  There's a catch that when pressed, allows a latch to swing over the connector, at which point one can rock it loose:  I disconnected the wire loom from the bracket on the sway bar disconnect body and draped it over the steering bar.
  • Disconnect the sway bar at each of the wheels.  This was easy - just a bolt on either side.
  • Undo the two clamps that hold the sway bar to the frame.  No problem here - just two bolts on each side.
  • Maneuver the sway bar assembly out from under the vehicle.  The entire sway bar assembly weighs probably about 45 pounds (22kg) so it's somewhat awkward, but it isn't too bad to handle.

Figure 3:
Inside the portion where the lead screw motor
goes:  Very clean - no contamination!
Click on the image for a larger version.
Before you get to this point I'd recommend that anyone doing this take a few pictures of the unit and also watch one or two YouTube videos as you'll want to be sure where everything goes, and under which bolt the small bracket that holds the wiring harness goes.

With the sway bar removed from the vehicle, I first  removed the end with the motor and connector and was pleased to find that it was perfectly clean - no sign at all of moisture or dirt. Next, I removed the other half of the housing, containing the gears and found that this, too, was free of obvious signs of moisture or dirt:  The only thing that I noticed was that the original, yellow grease was black in the immediate vicinity of the gears and the outside ring - but this was likely to due to the very slight wear of the metal pieces themselves.

The way that this mechanism works is that the motor drives a spring-loaded lead screw, pushing an "outside" gear (e.g. one with teeth on the inside) by way of a fork, away from two identical gears on the ends each of the sway bar shafts which decouples them - and when this happens, they can move separately from each other.  The use of a strong spring prevents stalling of the motor, but it requires that there be a bit of vehicle motion to allow the outside gear, under compression of the spring, to slip off to decouple the two shafts as they try to move relative to each other.

Figure 4:
The fork with the outside gear-cam thingie.  To disengage
the sway bar, the outer gear is pushed out further than
shown, disconnecting it from the end of the sway bar
seen in the picture above and allowing the two halves of
the rod to move independently.
Click on the image for a larger version.
When one "reconnects" the sway bar for normal driving, the motor retracts the lead screw and another (weaker) spring pushes the fork that causes tension on the outside gear so that it will move back, covering both of the gears on the ends of the  sway bar.  Again, some vehicle movement - particularly rocking of the vehicle - is required to allow the two gears to align so that the outer gear can slip over the splines and lock them into place.

In order to detect when the sway bar shafts are coupled properly, there's a rod that touches the fork that moves the outer gear and this goes to a switch to detect the position of the fork - and in this way, it can determine if the sway bar is coupled or uncoupled.  With everything disassembled, I plugged the motor unit back in and pushed the sway bar button and the lead screw dutifully moved back and forth - and pushing on the bar used to sense the position of the fork seemed to satisfy the computer and when pushed in, it happily showed that the sway bar was properly engaged.



What was wrong?

I was fortunate in that there seemed to be nothing obviously wrong mechanically or electrically (e.g. no corrosion or dirt) - so why was I having problems?

I manually moved the fork back and forth, noticing that it seemed to "stick" occasionally.  Removing the fork and moving just the outer gear by itself, I could feel this sticking, indicating that it wasn't the fork that was hanging up.  Using a magnifier, I looked at the teeth of the gears and noticed some small blobs in the grease - but poking them with a small screwdriver caused them to yield.

Figure 5:
Embedded in the grease are blobs of pink rubber
from the seal, seen in the background.
Click on the image for a larger version.

Digging a few of these out, I rubbed them with a paper towel and discovered that they were of the same pink rubber that comprised the seals:  Apparently, when the unit was manufactured, either the seal was pushed in too far, or there was a bit of extra "flash" on the molded portion of the seals - and as things moved back and forth, quite a few of these small pieces of rubber were liberated, finding their way into the works, jamming the mechanism.

Using paper towels, small screwdrivers and cotton swabs, I carefully cleaned all of the gears (the two sets on the sway bar ends and the "outside" ring gear) of the rubber.  A bit of inspection seemed to indicate that wherever these rubber bits had been coming from had already worn away and more were not likely to follow any time soon.

Figure 6:
More pink blobs - this time on the gear on the other sway bar.
Hopefully whatever "flash" from the seal had produced them
has since worn down and no more will be produced!
Click on the image for a larger version.

Putting an appropriate of synthetic grease to replace that removed, I reassembled the unit and put it back on the car, pushed the button.  Upon reassembly, I applied a light layer of grease on all of the moving surfaces involved with the shifting fork - some of which may have been sparsely lubricated upon installation.  I also put a few drops of light, synthetic (PTFE) oil on the leadscrew and the shaft that operated the sensing switch as both seemed to be totally devoid of any lubrication.

Although there was no sign of corrosion, I applied an appropriate amount of silicone dielectric grease to the electrical connector and its seal - just to be safe.

Did it work?

With the engine off, but in "4-Low", I could hear the lead screw motor move back and forth, and upon rocking the car gently I could hear the fork snap back and forth as it sought its proper position.  Meanwhile, on the dashboard, the "Sway Bar" light properly indicated the state of the mechanism:  Problem solved!

All of this took about two hours to complete, but now that I know my way around it, I could probably do it in about half the time.

Random comments:

I'd never really tried it before, but I was unsure if the motor would operate if the engine was not running:  It does - pressing the "Sway Bar" button alternately winds the lead screw in and out - but it's not really obvious as to its position if the cam doesn't lock into place and the light turns on solid or goes out.  Of course, this thing doesn't operate unless one has shifted to four wheel drive, low range.

This page stolen from


Wednesday, June 30, 2021

A "portable", high power, high-sensitivity remote repeater covering deep river gorges in Utah

From the late 1950s until about 2021, there was a (mostly) annual event held in southeastern Utah that was unique to the local geography:  The Friendship Cruise.

The origins are approximately thus:  In the late 1950s, an airboat owner - probably from the town of Green River, Utah - decided to go down the Green River, through the confluence of the Green and Colorado rivers, and back up to the town of Moab.  Somehow, that ballooned into a flotilla in later years - with as many as 700 boats - in the 60s and 70s.  By the mid 90s, interest in this unique event seemed to have waned and by about 2012, it seems to have petered out.

Communications is important:

Figure 1:
A high-Q 80 meter magnetic loop
on one of the rescue boats
Click on the image for a larger version
From the beginning it was realized that there was a need for the boats and support crews to be able to communicate with each other - but the initial attempts using CB and/or public safety VHF radios were unsuccessful, reaching only a few miles up and down the river - not too surprising considering that most of the course runs through winding, deep (1200 foot deep, 365 meter) gorges.  In later years, cell phones - and even satellite phones - were tried, but due to the remoteness and narrowness of the gorges they were of extremely limited use.

At some point, probably in the mid 1960s, amateur radio operators got involved, successfully closing the communications link using the 80 meter amateur band.  This tactic worked owing to the nature of 80 meters:  During the daytime, coverage is via skywave over a radius of about 200 miles (300km) and this high angle of radiation allowed coverage into and out of the deep canyons.  Furthermore, the same antennas that were small enough to be usable on boats, vehicles and temporary stations on this band were well-suited for radiation of RF energy at these steep angles.

For (literally!) decades, this system worked well, providing coverage not only anywhere on the river, but also to the nearby population centers (e.g. Salt Lake City) where other amateur radio operators could monitor and relay traffic as necessary and summon assistance via land line (telephone) if needed.  Because the boats were typically on the river only during the day, this seemed to be a good fit for the extant propagation.

While it worked well, it was subject to the vagaries of solar activity:  An unfortunately-timed solar flare would wipe out communications for hours at a time, and powering and installing a 100 watt class HF transceiver and antenna was rather awkward.  Occasionally, there was need to communicate after dark, and this was made difficult by the fact that 80 meters will go "long" after sunset - often requiring stations much farther away (e.g. in California or Nebraska) to relay to stations just a few 10s of miles away on the river!  Finally, it was a bit fatiguing to the radio and boat operators to have to listen to HF static all day long!

Enter VHF communications:

Figure 2:
General coverage map of the course
showing coverage of various sites.
Click on the image for a larger version
While VHF communications had been tried early on - and had been available in the intervening years - the biggest problem was that these signals could not make their way along the river for more than a few miles between twists and bends in the deep river gorges.  While useful for short-range communications, it simply wasn't suitable for direct boat-to-boat communications along the vast majority of the river's course.

By the time that the 1990s had come along, there was renewed interest in seeing if we could make use of VHF, on the boats, on the river.  The twist was that instead of direct communications between boats, we would try to relay signals from far above, on the plateaus farther away, and a few experiments were tried.  It 1996, I was on a boat on the river and took notes on what sites covered and where, trying nearby mountaintop repeaters and temporary stations set up at places near-ish the river courses themselves - the resulting map being presented in Figure 2.

Using the color-coded legend across the top and the markings on the map itself, one can see what sites covered where.  Included in this was the coverage from the 147.14 repeater near-ish Green River, Utah, the 146.76 repeater near Moab, and several other temporary sites atop the plateaus surrounding the river.  As can be seen, coverage was spotty and inconsistent over much of the route - with the exception of a site referred to as "Canyonlands Overlook" (abbreviated "Cyn Ovlk") which commanded a good view of the Colorado River side of the river course.  Clearly missing was reasonable coverage in the depths of the gorges along the lower parts of the Green River side - which started, more or less, where the coverage of the "Spring Canyon" (abbreviated "Spring Cyn") stopped.

Figure 3:
The two TacTecs used for 2 meter reception,
the voting controller (blue box) and the FT-470 used
as the UHF link radio.
Click on the image for a larger version.
As it happened, there were amateur radio operators camping at a site called Panorama Point when I was on the lower Green River and because we were using the Utah ARES simplex frequency, they just happened to hear the simplex activity on the river.  At that moment, I happened to be in areas that were not well-covered by any of the other sites and while their signals weren't extremely strong, it made me wonder what could be accomplished should I wield both gain antennas on the receiver and high power and gain antennas on the transmitter of a 2 meter repeater.

The birth of a repeater:

During the next year I put together a system that I'd hoped would make the most of the situation.  Because of the remoteness of the site, accessible via a high-clearance Jeep road and that we had to bring everything to live for a few days, it had to be relatively lightweight and compact - and I also wanted to avoid the use of any duplexers (large cavity filters) that would add bulk and - more importantly - losses to the system.  Taking advantage of a weekend to visit Panorama Point the next spring we determined that we could split the transmit and receive portions by about 0.56 miles (0.9km) apart, placing the receive antennas behind some local geographical features, using local topography to improve isolation.  The back-of-the-envelope calculations indicated that this amount of separation - and the rejection off the backs and sides of the beam antennas - would likely be sufficient to keep the receiver out of the transmitter.  The receive site - surrounded by three sides by vertical cliffs - also provided a commanding view of the terrain as can be seen in Figure 5, below.

Figure 4:
GaAsFET preamplifier mounted right at the
receive antenna to minimize losses.
Click on the image for a larger version.

In addition to site separation and gain antennas, I decided to go overboard, adding mast-mounted GaAsFET preamplifiers, right at each antenna (Figure 4) and implementing a voting receiver scheme - something made much easier with the acquisition of two, identical RCA TacTec "high band" VHF transceivers.  These receivers were modified - clipping the power lead to the transmitter and adding a 3.5mm stereo plug to each radio to bring out both discriminator audio and the detector voltage from the squelch circuit.

A relatively simple PIC-based repeater controller was constructed, using a simple comparator to determine which receiver had the "best" signal, based on the detector voltage from the squelch circuit, and also using another set of comparators and onboard potentiometers to set the COS (squelch) setting for the receivers.  As it turned out, the front-panel squelch control adjusted the gain in front of the squelch detectors in the radios themselves, allowing each receiver to be "calibrated" from that control, allowing easy fine-tuning in the field.

To link the receiver site to the transmitter site, a single UHF channel was used and I modified my old Yaesu FT-470 handie-talkie to this task.  The mysterious rubber plug on the side of this radio was replaced with a 3.5mm jack, providing a direct connection to the modulation line of the UHF VCO while using the top panel 2.5mm external microphone jack for transmitter keying.  As it turns out, not only did this transmitter provide linking to the nearby transmitter site, but its UHF beam was pointed across the way, to another 2 meter repeater at Canyonland's Overlook that provided coverage on the Colorado River - providing what amounted to a linked repeater system.  A later addition was a CdS photocell on a grommet and a piece of "Velcro" strap allowed the detection receiver activity by "looking" at the front-panel LED to prevent the link transmitter from "doubling" (transmitting at the same time) and clobbering an ongoing transmission from the other repeater site.

Figure 5:
The remote RX site, surrounded on
3 sides by sheer cliffs.  The mast
has two 2 meter and one UHF link
beam antenna.  The solar panels are
just visible along the far right edge.
Click on the image for a larger version.
One of my goals was to minimally process the audio, causing as little "coloration" as possible to maintain quality, and to this end I took the receivers' discriminator audio from the voter and put it directly into the modulator of the UHF link radio, completely avoiding the need for de-emphasis and pre-emphasis.  This worked pretty well - but I noticed during the first year that it was used that when weak signals were present on the input, the noise and hiss from weak signals would sometimes cause "squelch clamping" on the receivers being used by us and others owing to the fact that such noise was being passed along the link without alteration:  For the next year I added a 3.5 kHz low-pass filter in the transmit audio line to remedy this.

The receive site itself was solar-powered, using lead-acid batteries to provide the energy when insufficient sun was available (e.g. heavy clouds, night).  In later years, the PIC controller was modified to not only read the battery voltage, but to regulate the solar panels' charging of the battery bank using a "bang-bang" type charger (See note 1) but also to report the battery voltage when it did its legal identification.  In this way, we could keep an "eye" on things without having to walk out to the receive site.

The two 2 meter and the 70cm link antennas were mounted on a single mast, the VHF antennas pointed in different directions to take advantage of the slight difference in physical location and in the hopes of providing diversity for  the weak signals from the depth of the canyons - which were all reflections and refractions.  As it turns out, despite the close proximity of the antennas, this worked quite well:  At the site, one could monitor the speakers on the receivers and watch the voting controller's LED and see and hear that this simple, compact arrangement was, in fact, very effective in reducing the number of weak-signal drop-outs caused by the myriad multipath.

In testing on the work bench, the measured 12dB SINAD sensitivity of each of the receivers (plus GaAsFET preamps) was on the order of 0.9 microvolts - far and away better than a typical receiver.  Later, I did the math (and wrote about it - see the link at the bottom of this article) and determined that it was likely that the absolute sensitivity of this receiver was limited by the thermal noise of the Earth itself and that it could not, in fact, be made any more sensitive.  This notion would appear to be borne out by a careful listening to the repeater in the presence of weak signals:  Very weak signals - near the receive system's noise floor - sounded quite different than what one might hear on a typical FM receive system near it's noise floor.  Instead of a "popcorn" type noise, signals seemed to gradually disappear into an aural cloud of steam.

The transmitter site:

Figure 6:
The transmit site.  The tall (30 foot) mast and 2 meter transmit
antenna is visible in the background with the UHF link
antenna and the VHF "backup" TX antenna in the foreground.
Click on the image for a larger version.

With so much effort having gone into maximizing receiver performance, I decided to do the same on the transmit site in the years that this system was used.  For the first year, the transmitter was modest:  A Kenwood TM-733, on low power, driving a 50 watt RF amplifier into a vertical on a short mast.

The next year I decided to erect a taller mast and place atop it a 5 element beam, pointed in the general direction of "up river".  To boost my RF output power, I scavenged a pair of 110 watt RF amplifiers from some ancient Motorola Mocom 70 mobile radios (with some DC fans for cooling) and used two Wilkinson Power divider - one to split the input power and another to combine the outputs of the amplifier, yielding a bit over 200 watts of RF and about 1500 watts of ERP (Effective Radiated Power) - all without causing any measurable desensitization of the receiver system.  After a few days, one of these amplifiers failed, but the remaining 110 watt amplifier, now operating without the combiner, happily chugged along.

The next year I acquired a 300 watt Vocomm amplifier and was able to use it for the remainder of the times that the Friendship Cruise was held.  Requiring 50 watts of drive, I still had to use the 50 watt amplifier, driven by 5 watts from the TM-733 to attain the full RF output.  When keyed down, the entire transmitter system drew about 60 amps at 12 volts from the battery bank, requiring frequent topping-off by a generator and DC power supply that were brought along. (See note 2)

With that much transmit power, the antenna was held aloft by a 30 foot (9 meter) mast to keep it well clear of people - and to help clear the local terrain and its effects.  As can be seen in Figure 6, there was a second mast with the UHF link antenna and a "back-up" 2 meter antenna.  When we arrived at the site, the first order of business was usually to set up the receive site, but once back at camp, we used a radio in cross-band mode and the two antennas on the short mast to get it on the air, providing "reasonable" transmit coverage.  Because of the effort required to set up the tall mast, battery bank and power amplifier, we often waited until the next morning to complete the setup, bringing our radiated transmit power up to its full glory!

"Listening" on the link frequency, this transmitter not only relayed my own, nearby receive site, but also the "other" repeater at Canyonland's Overlook. 

How well did it work?

The Panorama Point repeater itself worked better than we could have hoped:  It was "reachable" nearly everywhere on either the Green or Colorado River - although some sections of the upper Green and Colorado had somewhat weaker signals, requiring a good antenna and 50 watt radio - comparable to a typical car mobile installation - for reliable coverage.  Unexpectedly, it also provided coverage into the town of Moab, as far north as Price, Utah and even down near Hite, Utah - both well outside its expected coverage range.  I'm confident that if I'd simply plopped down a "store bought" repeater with a single antenna and cavities, its performance - particularly on receive - would have been very much inferior as the signals from the depths of the gorges on the upper Green River were very weak and "multipathy". (See note 3)

With about 2.5kW of ERP one would expect that this repeater would have been an "alligator" (all mouth, no ears) but this was not the case:  When users were operating from the more extreme fringe areas - as in a deep river gorge, using a 50 watt mobile radio - the transmitter and receiver seemed to be more or less evenly matched, and despite running this much power, we did not experience any detectable "desense" where the strong transmit signal would overload the receiver.  At least part of this was attributed to the receivers themselves:  The RCA TacTec receivers used only modest amounts of RF gain in their front ends and a passive diode-ring mixer.  I have little doubt that if we had used more "modern" receivers we would have experienced overloading and would have had to place notch cavities, tuned for the transmit frequency, between the GaAsFET preamps and the receivers.

As a system, the Panorama Point and Canyonlands Overlook repeaters completely replaced the need for HF gear on the boats in the last decade or so that the Friendship Cruise was held, providing nearly seamless coverage from start to finish.

 * * *

Note 1:   A "bang-bang" solar regulator simply connects the solar panels directly to the battery when the voltage is too low - say, 13.2 volts - and disconnects them again when it rises above about 13.7 volts.  The PIC software implemented a timer so that after a disconnect from the panel when the voltage was high, it would not reconnect for at least 30 seconds, preventing rapid cycling.  With an open-circuit voltage of around 15 volts for the panels used, this was a simple, safe and reasonably efficient approach that could simply not cause radio-frequency interference in the way many modern "MPPT" solar chargers might.

Note 2:  In the later years, a pair of 40 amp switching power supplies were used at the transmitter site to charge the battery as quickly as possible.  Not unexpectedly, we could load the generator to only about 60% of its rated output, owing to the terrible power factor of these supplies caused by their simple capacitor inputs:  Power-factor corrected supplies were not cheap and readily available at that time.  Also in later years, a very low power (1 milliwatt) 2 meter transmitter was constructed, connected to the battery bank, that telemetered the battery voltage using MCW (Morse Code).  If the battery voltage got too low, this transmitter would activate a subaudible tone and a receiver that had been parked on this frequency, configured to detect that tone, would remain silent unless/until the voltage dropped below the threshold, alerting us to the need to start the generator.

Note 3:  "Multipath" is when a signal - likely due to obstructions - finds more than one way to the other end of the communications path via reflection and refraction - a condition that is the rule rather than the exception when trying to get signals in/out of the deep gorges along these rivers.  While these multiple signals can reinforce each other, they are equally likely to cancel each other out.  By having multiple receivers and antennas - even two antennas very close to each other - the probability is significantly higher that at least one of the receiver/antenna combinations will be able to hear such a signal.  Because of the nature of FM signals, one can generally infer its quality by analyze the amount of noise on it:  By comparing the amount of noise on the same signal, from two different receiver/antenna combinations - and always selected the "better" signal - the probability is increased that the received transmission will suffer less degradation.

* * *

Additional (related) articles:

This page stolen from



Saturday, May 22, 2021

Characterizing the RTL-SDR Blog (Version 3) for HF reception using the "direct" input.

An inexpensive option for SDR (Software Defined Radio) reception on the HF (low frequency) bands is a device sold by "RTL-SDR Blog" - the current iteration being Version 3.  Originally intended for digital VHF/UHF TV reception - and that of FM broadcast - the hardware is also capable of tuning much lower frequencies.

Figure 1:  An RTL-SDR Blog V3 USB receiver "dongle".
Unlike most other inexpensive RTL-SDR dongles, this has - via a single SMA port - the ability to operate in "direct" mode where RF below the VHF frequencies is passed straight to the A/D converter rather than via a down-converter, allowing reception from (theoretically) a few hundred kHz to around 30 MHz.

How does it do this?

The typical RTL-SDR Dongle actually consists of two tunable devices:

  • The Rafael R820T.  This is simply a frequency converter, capable of handling an input signal from somewhere below 60 MHz into the GHz range and converting it to what we'll call an "IF" (Intermediate Frequency) - which is actually somewhere in the 3.5-4.6 MHz range.  In addition to having a programmable oscillator and mixer for frequency conversion, his device has some built-in filtering that provides some protection to strong-ish off-frequency signals, and it has an AGC (Automatic Gain Control) that can adjust the level being output from it to prevent overload of the A/D converter as well as some front-end attenuation control to reduce the likelihood of overload on the input.
  • The Realtek RTL2832U.  The down-converted output of the R820 chip is passed to this device, which consists of two 8 bit A/D (Analog-to-Digital) converters that are clocked at 28.8 MHz, a USB interface, a (reported) microcontroller and a digital frequency "converter" that is also capable of being "tuned" to produce a quadrature "baseband" signal that is output onto the USB port - the rate being programmable from around 250 ksps to 2880 ksps.  This device does NOT have an AGC or gain/attenuation control.

VHF/UHF operation:

Briefly, reception on the VHF/UHF frequencies is done the following way:

  • A simple high pass filter/diplexer passes only the VHF/UHF signals (e.g. those above approximately 40 MHz) directly to the R820T.
  • The filtering of the R820T is programmed for the desired characteristics at the operating frequency.
  • The frequency converter in the R820T is offset from the input signal to provide an output (IF) signal in the 1-14 MHz range - more likely somewhere between 5 and 12 MHz.
  • The level of this output signal may be automatically controlled by the AGC system of the R820T to keep it within the optimal range of the A/D converter.  Similarly, input gain adjustment on the R820T can prevent it from being overloaded by strong signals.
  • The tuner within the RTL2832U is set to about the same frequency being output by the R820T (likely in the 5-12 MHz range) so that it is within the range of output ("baseband") sample rate, and output via the USB port.

HF operation:

  •  The signal to be received is applied via the RF antenna port and diverted from the R820T to a very simple low-pass filter/diplexer to an RF amplifier.  This diplexer's effect begins at approximately 22 MHz - See the discussion near table 1, below.
  • The output of from the RF amplifier is passed through additional filtering and applied to one of the two A/D inputs of the RTL2832U - typically the "Q" input.
  • The tuner within the RTL2832U is set near the frequency to be received so that it is within the range of the output ("baseband") sample rate, and output via the USB port.

There's a penalty to pay for simplicity:

For those familiar with receiver topology - digital or analog - several things the above description of the HF operation of the RTL-SDR dongle without proper measures (e.g. filtering, gain control) should give cause for concern - namely:

  • There is NO bandpass filtering at all.  Whatever is being intercepted by the antenna system will be input directly to the A/D converter via the preamplifier.
  • ANY signal applied to the antenna input - no matter the frequency - can contribute to overload.  Because the inputted RF goes into the A/D converter, a strong signal well away from where you are listening can cause overload.  For example, if you are listening to 14 MHz, a strong AM broadcast signal (e.g. mediumwave) could well be the culprit if you are experiencing overload.
  • There is no AGC in the HF ("direct") signal path.  Considering that the A/D converter is only 8 bits, this means that compared to even low-end shortwave receivers, the range of signal levels over which this device will operate is very narrow.  The lack of an "AGC" (automatic gain conttrol) means that there will likely not be enough gain when signals are very weak and overload of the A/D converter is likely if the signals are very strong.
  • The A/D converter's sample rate is 28.8 MHz.  What this means is that the Nyquist limit - the frequency above which the digitized output can no longer represent the input signal - is half this, or 14.4 MHz, uncomfortably close to the 20 meter amateur band, and entirely below the 17, 15, 12 and 10 meter bands - and this does not even consider the fact that the sample frequency is within the 10 meter band itself. 
What this means is that there are unsuppressed image responses all across the HF spectrum.  For any frequency you tune below 14.4 MHz, you can also hear any signal above 14.4 MHz, the frequency of which is calculated by subtracting its frequeny from the sample rate (e.g. 28.8 MHz).  For example, 15 meter signals will also appear, spectrally inverted (e.g. USB = LSB) 7.80-7.35 MHz.  Image frequencies appear in table 1 - along with MDS and clipping values - below.

In other words, if you are tuned to any HF frequency, you are actually "hearing" TWO frequencies at the same time unless you have specific filtering to prevent such image response.  See the right-hand column of Table 1 (below) for the frequency at which you will see an image.

The above put strict limits on the performance of the RTL-SDR dongle as an HF receiver and these realities must be considered when the configuring any system that might use them.

The usable dynamic range:

In theory, 8 bits of A/D sampling would indicate about 48dB of useful signal range, but the reality is more complicated than this.  Compared to the bandwidth of the narrow signals typically sought on HF, the overall sample rate of the A/D converter - and even the rate after the RTL2832U's converter has reduced the signal to the sample rate being sent to the USB port - are very much higher, effectively improving the bit depth via oversampling - and the fact that the HF spectrum is backgrounded with what amounts to white noise works to our benefit, helping to spread discrete, spectral artifacts that are the inevitable result of the imperfect signal aquisition.

All of this makes the actual, usable range a bit difficult to divine.  To this end, an RTL-SDR (V3) unit was put on the workbench, using the "HDSDR" program as a receiver, and its operation was analyzed by observing CW (unmodulated) signals on narrow (SSB) bandwidths.


Ideally, we would be able to determine the RF level at which the A/D convert clipped directly, but the HDSDR program does not provide a means to see the peak A/D level, requiring us to infer it by noting the level from a known-accurate signal generator at which the S-meter starts to decrease at the same rate that the signal level is reduced - and also by noting the disappearance from the waterfall many of spurious signal caused by overload.  Typically, this is about 3dB below the "maximum" S-meter reading.

For sensitivity, DL4YHF's "Spectrum Lab" program was used to measure the SINAD in a 500 Hz bandwidth set by the HDSDR program, the "minimum discernible signal" being equivalent, in this test, to 3dB S/N.

Table 1:  Measured signal levels for A/D clipping and MDS, along with corresponding image frequencies on HF.  Because the sample rate of the RTL-SDR is 28.8 MHz, ALL signals above half this frequency (e.g. 14.4 MHz) are, by definition, Nyquist images.
Frequency (MHz)
Clipping (dBm)
MDS (dBm)
Image  (MHz)


Table 1 tells us several things:

  1. As mentioned in the documentation found at the RTL-SDR Blog web site, signals below 2 MHz - and especially below 1 MHz - are rolled off by the "Bias Tee" blocking choke which has insufficient inductance at frequencies below the AM broadcast band.  To a degree, this effect can be mitigated by removal of the RF choke which will remove the capability to inject DC onto the cable.
  2. The RTL-SDR has a low-pass filter to diplex the HF and VHF and above frequencies to separate signal paths, and the effects of this filter are becoming evident above 15 meters (21 MHz).  This also means that by itself, the RTL-SDR becomes "deaf as a post" on the 12 and 10 meter bands.
  3. Note that at the very low frequencies, the sensitivity appears to be somewhat reduced.  The effects of high-pass roll-off - likely caused by coupling capacitors and the input bias inductor - appear to be evident through at least 7 MHz.
  4. There is clearly no image rejection at all, considering that the sensitivity through 21 MHz is comparable to that below 14 MHz.  For example, 15 MHz WWV will also appear at 13.8 MHz.
  5. Low-pass effects are evident by 24 MHz, most likely a result of the diplexer used to split the HF (direct) and VHF/UHF signal paths.  This limits sensitivity on the 12 and 10 meter amateur bands.
  6. With only 8 bits of quantization, additional noise will be generated due to the imprecise nature of the process.  This "noise" will show up as spurious signals and, less obviously, as a rise in the overall noise floor - depending on the nature of what is being digitized.  In short, the fewer the number of bits, the less likely it is that weak signals will coexist (and be audible) in the presence of strong signals.
  7. Across most of the HF spectrum, the RTL-SDR will overload signal of about -30dBm - which is approximately equal to an S-meter reading of "40 over S-9".  While this seems like a fairly strong signal, this power level represents the total amount of RF energy - no matter the frequency!
  8. Note that the power at which the A/D converter starts to overload is approximately -30dBm across much of the HF spectrum.  This represents the TOTAL amount of RF power required, at all frequencies combined, that will result in overload.

Points 7 and 8, above, should be considered very carefully in terms of its implications:

  • You cannot simply connect an RTL-SDR Dongle to even an "average" performing HF antenna and expect reasonable results as the total power from ALL signals reaching the receiver are likely to exceed the -30dBm signal level.
  • Particularly on the lower bands (80, 40, 30 meters) the signal levels of amateur and especially shortwave broadcast signals can, by themselves, exceed the -30dBm overload level - particularly in Europe and the eastern U.S. On some bands, the shortwave broadcast band adjacent to the amateur band (e.g. 41 and 40 meters) are too close to effectively filter out and it may be that an RTL-SDR is simply not usable on these bands when propagation favors reception on these frequencies. 
  • As seen in Table 1 (above) the RTL-SDR dongle becomes increasingly deaf on the higher HF bands (particularly 12 and 10 meters) making them unusable at these frequencies without additional amplification AND filtering.  What's worse is that the lower HF frequencies (e.g. below 10 MHz) are typically very noisy while the higher frequencies are quiet by comparison.  If you connect an antenna to the RTL-SDR dongle with no band-pass filtering and try to tune in, say, the 15 meter band, you will likely hear only noise (and signals) around 4.8 MHz, which will likely overwhelm any weak, 15 meter signals.
  • If you plan to use an RTL-SDR for HF reception and expect even mediocre performance, you should precede it with a band-pass filter for the frequency band of interest.  For the highest HF bands (15, 12 and 10 meters) the typical noise floor in a quiet location is around -120 dBm (in a 500 Hz bandwidth) - which is well below the noise floor of the RTL-SDR at these frequency meaning that a preamplifier (along with a bandpass filter) will be required for reasonable performance.
  • It's worth remembering that unlike an analog receive system, one cannot always use all 8 bits for digitization:  The signal input must be kept well below the "full scale" level to accommodate for random fluctuations that are ever-present on signals input from the antenna.  What this means is that the A/D must be under-driven overall and that fewer bits are actually being used most of the time.  In order to maintain suitable margin, it's typical to drive the A/D converter at between 1/4 and 1/2 full scale, meaning that for 8 bits of A/D conversion, 2-4 bits are typically being used.


  • DO NOT simply connect an RTL-SDR to your HF antenna and expect it to work as well as even a low-end shortwave receiver:  If it doesn't get overloaded by local AM broadcast (mediumwave) signals, it will get overloaded by strong shortwave broadcast signals when conditions are favorable on certain bands.  This isn't to say that you won't hear anything if you do so, but know that normal signal levels present on even an "average" antenna will be enough to overload the RTL-SDR dongle.
  • ALWAYS precede an RTL-SDR with a band-pass filter that is specific for the frequency range of interest. For example, if you are interested in 40 meter reception (7.0-7.3 MHz for ITU region 2) your filter should pass only frequencies in this range, and this filtering will prevent unwanted reception of signals at the image frequency around 21.8 MHz.   Unfortunately, the use of a band-pass filter precludes reception outside its design range, but this is necessary considering the limited capability of the RTL-SDR dongle in terms of handling both strong and weak signals at the same time and its unfettered response to unwanted images.
  • In some cases - even with a mediocre antenna - the signals in the desired frequency range may exceed the signal handling capability of the RTL-SDR and cause overload.  As noted above, a single signal stronger than -30dBm can do this, but so could a number of signals whose total power can exceed this.  Typically, this overload is manifest as intermittent distortion across the entire receiver as signals fade in and out.  In such cases, it might be beneficial to attenuate the signals reaching the RTL-SDR as the degradation caused by overload is more "destructive" across the entire receive frequency range than too-little signal.
  • Because of image response and roll-off, the HF port of the RTL-SDR is really not well-suited for 12 and 10 meter (24-30 MHz) reception.

Comment about HF upconverters:

Specifically to address some of these issues, there are upconverters available for the RTL-SDR that will upconvert the HF spectrum to VHF (typically in the 100-130 MHz range) to allow the use of that signal path, making use of the Raphael R820T converter.  This converter has the advantage of having a degree of band-pass filtering and the ability to use AGC (automatic gain control) on the signal path.

This method can be useful, but there are several caveats:

  • If frequency stability is of importance, the addition of the upconverter introduces two additional frequency stability issues:  Drift of the upconverter itself, and the fact that any existing drift in the dongle itself will be multiplied because of its operation at the higher frequency.  This can be an issue in environments where the temperature is not stable and/or when a frequency sensitive mode like SSB or (especially) digital modes are used.
  • If multiple bands and receivers are to be used, there may be the temptation to upconvert the output of the upconverter to VHF and distribute this signal to the receivers.  While this may work in many cases, it's worth noting that if the entire HF spectrum is converted, the total signal power level can be significant, potentially overloading the upconverter itself, any RF amplifiers that might be used at VHF, and/or the front end of the RTL-SDR dongles themselves.
    • While a "fix" might normally be to filter out just the HF band(s) of interest, this can become impractical if the 40 meter band (7.0-7.3 MHz) is upconverted to, say, 137 MHz where it can become difficult to make an effective band-pass filter at that frequency.
    • It is possible to filter for a specific HF band before the upconversion, but this means that each RTL-SDR would require its very own upconverter.  While effective, the cost of an upconverter for each band may be prohibitive.

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Successfully using an RTL-SDR on HF:

As mentioned earlier, one must precede the RTL-SDR with a band-pass filter to obtain reasonable performance on HF - or any other frequency range where very strong and very weak signals will be simultaneously present at the antenna input:  Remember that, especially in the "direct" mode, all signals applied to the RF input - even those MHz away from where the receiver is tuned - will count "against" you in terms of the total amount of RF power that may be applied to the A/D converter before it clips/overloads.

To see some RTL-SDR based HF receive systems in operation - and to be able to directly compare them with higher-performance receivers using 16 bit A/D converters, visit the Northern Utah WebSDR site at (link).  This will take you to a "landing page" where you can select several receivers - specifically:
  • WebSDR #3:  This server uses RTL-SDRs for both the 80 and 40 meter bands.
  • WebSDR #1:  This server uses 16 bit sound cards and "softrock" receivers for reception of the 80 and 40 meter bands, split into 5 segments.

Both of these systems use the same antenna for 80 and 40 meters and it is possible to directly compare signals between the two in side-by-side windows.  Generally, the 8 bit RTL-SDRs used on WebSDR #3 hold their own compared to those on WebSDR #1, and this is possible only because the WebSDRs are preceded with both band-pass filtering and AGC as described in the link below.

* * *

Additional resources:

  • An article on using band-pass filtering and AGC (Automatic Gain Control) to improve the performance of an RTL-SDR when used for amateur band service may be found here: "test" receivers are currently in operation at the Northern Utah WebSDR that demonstrate the efficacy of doing this.

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