Sunday, May 17, 2020

A quick look at the QB-300 RF amplifier

Available from many surplus sellers (e.g. via EvilBay) - and (usually) for a reasonable price - is the QB-300 RF amplifier.  Originally made by Q-Bit corporation, this same device has borne several different manufacturers markings over the 30+ years since it was introduced - but it is (pretty) much the same device.
Figure 1:
The BNC-connectorized version of the QB-300.
This appears to be the "original" version, actually made by Q-bit
Corporation.  The voltage specification is slightly ambiguous, being
shown as "+15/24 Vdc".
Click on the image for a larger version.

Having several of these on-hand I decided to take a quick look at its apparent performance - with the general specifications for this device being listed below for your convenience:
  • Frequency range:  1 MHz-300 MHz
  • Gain:  23dB (or 24.5 +/- 1 dB, depending on source)
  • Gain flatness:  1 dB
  • Noise figure:  3.8dB (frequency not specified)
  • Input/Output VSWR:  <=1.5:1
  • Power output (1dB compression):  +22dBm
  • 3rd order Intercept:  +37dBm
  • Current consumption:  155mA (voltage not specified)
Depending on which data sheet you consult, there are a few discrepancies - for example:
  • The data sheet from "API Technologies" shows the input/output return loss as 1 dB - clearly a typo.
  • The maximum voltage rating is all over the map:  Some versions of the data sheet show a maximum of 20 volts, others show 24 volts.  The units that I have clearly show the voltage rating as being "+15/24Vdc" and the equipment from which it was pulled provided 24.0 volts.
Knowing the provenance of this equipment, I would have no problem running my amplifier from 24 volts, but based on the ambiguity of the data sheets, I would operate a version that did not explicitly specify 24 volts ONLY from 15 to 18 volts.

A quick test:

Curious about a few aspects of these amplifiers I decided to test it with my DG8SAQ Vector network Analyzer, checking its gain versus frequency in the input matching (e.g. S11) - the results being displayed in Figure 2, below:

Figure 2:
A sweep of the amplifier from 100 kHz to 500 MHz showing the apparent gain and input matching over the frequency range.  Because the DG8SAQ and the interconnecting cables are increasingly imperfect with increasing frequency, expect increasing uncertainty in the S11 readings above 100 MHz or so.
The gain, S11 and VSWR values at specific frequencies can be seen in the upper-left corner.
Click on the image for a larger version.
Of particular interest was the usability of this amplifier above and below its "official" frequency range - and we can see that it's probably useful down to at least 250 kHz and above 450 MHz, albeit at reduced performance (e.g. lower gain, maximum output power, increased noise figure, increased input VSWR.)  Specifically, I measured, below the 1 MHz minimum frequency specification:
  • The gain at 285 kHz was still above 21dB - a bit more than a 2 dB drop from the peak gain.  The input VSWR was still below 1.5:1.
  • The gain at about 200 kHz was around 18.7dB
  • The gain at about 173 kHz was around 16.8 dB and the VSWR had increased to about 2.7:1. 
Above the 300 MHz maximum frequency specification:
  • The gain was above 22dB at 400 MHz
  • The gain was about 21 dB at 440 MHz
  • The gain was around 18dB at 500 MHz.
 The reader is reminded that it is likely that above and below the rated frequencies that the maximum output power - not to mention noise figure - is likely to degrade.

Gain versus operating voltage:

Figure 2 was captured with the unit operating at 18 volts and readings were taken at lower voltages, comparing the gain - but your mileage may vary:
  • Gain dropped by approximately 0.1 dB at 15.0 volts.
  • The gain was about 0.2 dB lower at 12.0 volts than at 18 volts.
  • The gain was about 1 dB lower at 8.0 volts than at 18 volts.
  • The gain was about 5 dB lower at 5.0 volts than at 18 volts.
  • The amplifier began to exhibit signs of low-frequency instability below 5 volts.

Although not directly measured, one should expect the maximum output power (P1dB) and the intercept point to drop below the specifications when operating it from lower than 15 volts:  The amplifier is likely to be perfectly usable in the 12-14 volt range, but it's likely marginal at 8 volts and below.

A peek under the hood:

Popping the top cover, we see this:

Figure 3:
A look inside the QB-300 amplifier:  The input and output is on the left and right sides, respectively.
Click on the image for a larger version.

 It is immediately apparent that this is not a run-of-the-mill consumer device:  Rather than a circuit board, the unit is built onto an alumina substrate with both soldering of components and spot welding of wires being used.  Two RF transistors are obvious:  The black, 3-lead device near the upper-left corner and the white ceramic device marked with "Q-21" just to its right.  The rest of the components are likely related to feedback/equalization as well as regulation of the operating and bias voltages for the RF devices.

Clearly, it's not hermetically sealed or conformally coated, so  weather protection is certainly warranted if this were to be operated outside.

Uses for this amplifier:

This amplifier was designed as a general-purpose gain block in the HF-VHF range, but it is likely useful into the low UHF range meaning that it should work from the 630 meter amateur band (on the low end) into the 222 MHz - and possibly the 70cm - amateur bands on the high end.

For general HF (amateur radio) amplification purposes, it should be an excellent performer - provided that one keeps in mind that it's gain may be a bit too high in certain applications in that a signal input level of a around -5dBm will push it into overload - and off-air signals of this strength might appear from:

  • Local AM broadcast stations.  Especially on a long wire antenna (longwire, rhombic, end-fed half-wave) these signals can, by themselves, overload the amplifier if you live anywhere near  a transmitter.  A simple high-pass filter can effectively reduce such signals and prevent overload.
  • High-power shortwave stations.  On a good antenna, signals on the 49, 41 and 31 meter band can be extremely strong in Europe and some parts of the U.S.
If you have strong signals that could overload the amplifier, beware using an attenuator on the input of the amplifier in your receive system.  As an example, if you wish to be able to hear the background noise at 10 meters to be able to hear the weakest possible signals you will need to make sure that your system noise figure is no more than about 15dB - but if you had a cable loss of 3 dB in "front" of your amplifier (between the antenna and the amplifier) and you used a 10dB attenuator in this signal path, you are already at 13dB - and the nominal 3.8 dB of noise figure of this amplifier will push that number to about 16.8dB meaning that your system noise will now likely be high enough that you can no longer hear atmospheric noise if you are fortunate enough to be in a very "RF quiet" location.

In short:  If you hear more noise when you connect your antenna system to your receiver system than when you connect it to a dummy load, you are OK - but if you can't hear the difference, your system will not be sensitive enough to hear the weakest signals.

For receive-only purposes it is often the case that with a low-noise amplifier, a good, quiet (in terms of noise) receive antenna will not need to have much gain from the antenna itself - and if the gain is low, you are less likely to intercept enough absolute signal power to overload the amplifier.  Here are just two of the many possible examples of antennas to consider:
  • Small receive loop.  This type of antenna - usually around 3 feet (1 meter) diameter for MF and HF use can offer local noise rejection as well as the ability to null signals from directions broadside the plane of the loop.  This type of antenna will have negative gain (e.g. less than 0 dBi) but its performance can be quite good with a decent, low-noise amplifier like the QB-300.  For an antenna like this, one would place the amplifier at the antenna to minimize cable losses.
  • Beverage on the ground.  Also known as the "BOG" antenna, this is simply a wire - as long as possible - laying on the dirt and working against a good (and electrically quiet) ground consisting of one or more ground rods and counterpoise wires and its feedline electrically decoupled (with a "current" balun) to prevent noise from the shack from being brought to the antenna.  This antenna - mostly useful in rural areas - is reported to work well overall despite the likely "negative" gain.  As with the receive loop, it's best to place the amplifier at the antenna feedpoint.
Amplifier and receiver protection:

It should go without saying that any amplifier (or receiver) connected to a large antenna should be preceded by adequate lightning protection to prevent damage to the amplifier from wind static/discharge and nearby lightning strikes as depicted in Figure 4, below.  Such filtering should be placed after any filtering that might precede the amplifier.

Decent protection can be had with four ordinary silicon diodes - two series pair connected anti-parallel (back-to-back) with a bleed resistor (4.7-100k) to shunt voltages above about 1.2 volts.  It's worth noting that the amplifier itself would already have overloaded before signals can a high enough level to cause the diode protection to conduct and cause distortion!
Figure 4:
Depiction of simple input protection circuitry.
On the left, the diodes ("D") are ordinary silicon diodes connected in series for approximately 1.2 volts of conduction.
On the right, a common full-wave rectifier module is used with its DC "output" shorted, providing an equivalent to the circuit on the left.  It is suggested that a low current (2-5 amp) rectifier be used.
The voltage rating of the diodes is not particularly important - a 50-100 volt rating being just fine.
Resistor "R" is not critical and can be anything from 4.7k to 100k and it is used to dissipate any accumulated DC in case the antenna itself does not have a DC ground.  An inductor can be used in addition to or instead of "R" - a value of 22-100uH (e.g. 8-10 turns on an FT50-75 toroid) being suitable for 630-10 meters.
Click on the image for a larger version.

Provided that one avoid excessive signal input level, it can also be used as the basis for a receive multi-coupler.  For example, following the amplifier with an 8-way RF splitter - which, itself, will have a loss of around 10dB - the overall gain will be in the range of 14dB while preserving the system's overall noise figure to allow reception of weak signals on the higher HF bands.

This page stolen from


Saturday, May 9, 2020

A "curiously sharp" 40 meter band-pass filter to reduce 41 meter SWBC overload.

At the Northern Utah WebSDR ( we recently added another server (WebSDR #4) that is connected to an existing east-pointing beam antenna on site. This antenna, it is hoped, will better-allow users to hear stations on the 40-10 meter bands in the eastern U.S. and, to a lesser extent, the DX locations to which it is pointed.
Figure 1:
East-pointing beam antenna at the Northern Utah WebSDR.
This antenna has 10-13 dBi gain and signals in the 39,41
and 31 shortwave broadcast bands can be extremely strong!
Click on the image for a larger version.

As one would expect, this antenna has gain - between 10 and 13 dBi, depending on the frequency - and this has implications when propagation between Utah and the Eastern U.S. is favorable:  Already-strong shortwave broadcast (SWBC) signals become even stronger.

Because the S-meter (signal meter) on the receivers have known calibration it is possible to make an indirect measurement of some of these signals' strength and, at times, individual signals have been observed in the -20 to -15 dBm range at the antenna - these levels being in the "60 over S-9" range.  At times - particularly during evening "gray line" propagation (where both the transmit and receive sites are entering/in twilight/evening) signals can peak significantly.  What's worse is that there may be several such signals, increasing the total RF power impinging on the receiver risking not only receiver overload, but also providing a ready source of multiple, modulated carriers to mix together and reappear within the receiver's passband among the desired signals.

These sorts of signals are far above those that might be expected due to amateur-only transmissions owing to the widely disparate signal levels.  For example, a very well-equipped "DX" stations may be able to run 1500 watts of RF into a (monster!) 15 dBi gain antenna and attain an EIRP (Effective Isotropic Radiated Power) in the area of 50kW, but this does not compare with an SWBC station which may be running 500kW peak (about 125kW carrier) into an antenna with (a conservative) 18dBi gain - an EIRP of about about 32 million watts - a signal level nearly 1000-fold (30dB) stronger than one that would be transmitted by law-abiding amateurs.

What's worse is that some of these SWBC bands are adjacent amateur bands - and the 40 meter amateur - which runs from 7.0-7.3 MHz is no exception as the 41 meter SWBC band is just above it, starting at 7.3 MHz.  With such close spacing, typical filtering in receivers have little hope in effectively rejecting these nearby, strong signals.

Addressing the problem:

There are two time-honored ways of dealing with strong signals impinging on receivers:
  • AGC (Automatic Gain Control):  This circuit "monitors" the signal level at the receiver and automatically reduces the gain when they exceed a certain amount.  In the past, this has been applied only to signals within the passband of the receiver's IF to keep the audio level constant regardless of the actual signal strength, but this is also applied to modern SDRs where the level of the entire passband of signals being input to the A/D converter is monitored and adjusted to prevent overload. 
  • RF front-end filtering:  With the advent of solid-state radios starting in the 60s and 70s the design of RF filtering used in amateur receivers began to be wideband, typically covering MHz, rather than a narrow peak.  This was done not only because it was easier to do so with these designs, but also because it allowed "general coverage" reception outside the amateur bands and it was significantly less expensive than mechanically-complicated, ganged tuning systems - but it had the down-side that signals some distance away frequency-wise could still cause the receiver to experience overload.  These days - particularly with modern, high-performance direct-sampling Software-Defined Radios (SDRs) - "narrow" filtering is once again being used, along with AGC, owing to the need - more than ever - to strictly control the total amount of RF energy reaching the A/D converter to prevent overload. The receiver used at the Northern Utah WebSDR is a type of SDR where the RF energy is converted directly to audio and then digitized.  This has the advantage of simplicity, but it lacks the "AGC" circuit meaning that it is possible for strong, off-frequency signals to cause overload of not only the RF circuits, but also the audio circuits and the A/D converter.

While it is possible to add an AGC circuit to this receiver system to prevent overload (this has been done with some of the other receivers on site - and is still an option) the first step that we are taking is to build a "sharper" filter.

The "Curiously Sharp" band-pass filter:

Passing signals on the 40 meter amateur band - which ends at 7.3 MHz in the Americas - and filtering out signals on the 41 meter shortwave broadcast band - which starts at 7.3 MHz - is a tricky proposition:  How does one suddenly go from passage of signals to blocking them within just a few 10s of kHz?

Figure 2:
The completed 40 meter band-pass filter
in a Hammond 1590D die-cast box.
Click on the image for the larger version.
The answer is:  You don't - but you do the best that you can!

The limiting factor in constructing a "brick wall" filter - one that has an abrupt transition - is physics and is intrinsic to real-world components:  Real-world inductors have ohmic resistance and capacitors have dielectric losses - to name but two factors - that limit the unloaded "Q" of the circuits.

What does this mean?  A truly "sharp" filter will ultimately be limited in its performance by these factors:  One must trade off insertion loss and/or filter performance in terms of how quickly our band-pass filter cuts off.

Fortunately, the first of these - insertion loss - is pretty easy to mitigate:  Have enough extra signal gain in the receive system to accommodate the insertion loss.  At 40 meters, we have "signal to burn" - partly because our receive antenna has so much gain, but there is also a "strong" (resistant to overload) RF amplifier located near the antenna to mitigate the effects of cable losses at the higher HF bands (10 meters).

Even if we didn't have both antenna and amplifier gain, we could afford to lose a lot of signal at 40 meters:  A system noise figure of about 30 dB (assuming a unity gain antenna) is sufficient to "hear" the noise on even a quiet band, so a significant loss can still be made up by placing an RF amplifier after the filter and still be able to resolve the 40 meter noise floor during quiet band conditions.

Figure 3:
 Schematic of the 40 meter bandpass filter.  This is a 7-element Elliptical (Cauer) filter centered at 7.15 MHz - the middle of the U.S. 40 meter amateur band.  It was originally designed with the aid of the "A.A.D.E. Filter Design" program, version 4.5 being available from the AE5X web site.
The nominal impedance of the filter portion is 800 ohms to permit higher values of inductance and lower values of capacitance in an effort to ease construction and to reduce component losses (e.g. reduce the L/C ratio).
Click on the image for a larger version.

Figure 3 shows the schematic of the filter - and a few explanations are warranted.

  • ALL of the capacitors must be either NP0 (a.k.a. C0G) ceramic or silver mica capacitors - preferably the latter.  I did not use any silver mica capacitors, but I used known-good ceramic capacitors from a trusted source (e.g. Mouser-Key) rather than from a random EvilBay seller.
  • L1, L4 and L7 were wound using solid 12 AWG copper wire.  The wire that I used happened to be tin-plated, but enameled copper wire will be just fine with only the two ends (and the tap point) being bared for soldering.  If bare copper wire is used it is suggested that it be very clean and sprayed with clear lacquer after construction is completed to prevent oxidation.
  • The other inductors were wound using 17 AWG wire, which was on hand, but 18 AWG would be fine.
  • All of the inductor/capacitor pairs have their own resonant frequency, noted on the diagram in parentheses.  The 7.15 MHz resonances (C1/L1, C4/L4, C7/L7) will be adjusted very close to the stated frequency but the other resonances (C2/L2, C3/L3, C5/L5, C6/L6) are made adjustable by small ceramic (or air) variable capacitors and must be CAREFULLY adjusted for the proper filter response.
  • As can be seen, the filter's in/out ports are terminated with 2dB resistive attenuators to help assure a consistent source/termination impedance to the filter and prevent the likely-imperfect devices to which it is connected from too-badly affecting the response.
  • L1 and L7 show taps that are chosen to be at the 50 ohm points.  The "S11" port of a known-calibrated VNA may be used to best-set the 50 ohm points of the taps during filter construction/adjustment.
Constructing the filter:

Figure 5, below, shows the as-built filter:

During construction I used my DG6SAQ Vector Network Analyzer - and a tool such as this is invaluable as it will give "live", dynamic readings to facilitate adjustments.  The more economical (approx. $50 U.S.) "NanoVNA" will work fine (along with the "NanoVNA Saver" program) - and although its update/sweep rate is quite a bit slower than that of the DG6SAQ, it's still usable.  No matter what sort of VNA you might use, be aware that the limited number of data points per scan can "hide" details such as narrow, deep notches - and this is especially true with the NanoVNA.

The "through loss" measurements (in dB) were the most important in this case as the insertion loss versus frequency plots over a range of about 6.5 to 7.8 MHz allowed the "dialing in" of the resonant circuits.  During construction two "bloody ended" coaxial cables were used - one end of each being connected to the VNA and the other end having its ground shield tacked to the ground plane and the center conductor attached to the point under test:  These test cables are visible in Figure 4, below.  This "plywood and foil" test bed allowed an easily reconfigurable circuit design and test bed for ideas - and, most importantly, it helped me determine if a particular design was even practical.
Figure 4:
Early prototype built on a piece of plywood covered with self-adhesive
copper foil.  Originally, L1, L4 and L7 were wound on toroids - but
a switch was made to the larger, air-wound inductors to reduce losses.
This early version used input/output transformers for transformation of
the 50 ohm in/out to the 800 ohm (nominal) impedance of the filter itself -
but this was changed to tapped inductors as that was simpler and
lower loss.  This simple "breadboard" allowed several ideas to be tried
before settling on the final version, giving plenty of room to work.
This picture shows the short pieces of coaxial cable that were
tacked to the foil ground:  These cables connect to the two
ports of the VNA used to analyze the response of the filter.
 Click on the image for a larger version.

The first to be constructed were the large resonators (L1/C1, L4/C4, L7/C7) which needed to be set to 7.15 MHz and for this, two resistors (1k-4.7k - the precise values are unimportant) were connected in series with the center point connected at the "top" end of the parallel L/C network and the "ends" being connected to the VNA's in and out ports.  With this arrangement one can see the "peak" where the L/C circuit resonates - the two resistors minimizing loading - and one compresses/stretches the large inductor using a small screwdriver to increase spacing between turns or a pair of needle-nose to compress them - or, if necessary, removes fractional turns - to "dial it in" at 7.15 MHz.

After these have been adjusted, the other L/C networks are then constructed - and this is where it gets to a bit tricky:  The variable capacitors allow the resulting "notch" to be moved around, but it may be necessary to add/remove turns from the inductor - or add small amounts of capacitance (10pF at a time) to get the circuit's adjustment within range of the variable capacitor.  In some cases, one may temporarily "shunt" (short out) one or more of the series L/C networks to better-visualize the notch that one is trying to adjust.  If you can't find the "notch", don't forget that it may be above/below the sweep range and you may temporarily need to set the start/stop frequencies wired to find it.

As often happens, one's first ideas don't work quite as expected:  You will note that in Figure 4, L1, L4 and L7 are shown as being toroidal inductors, but it became clear that these inductors were just too lossy, a factor that severely affected "Q" and performance.  I ended up using air-core inductors wound from much heavier wire as can be seen in Figure 5 to minimize loss.  Ideally, superconducting inductors would have been used, but for some reason such devices that operate at room temperature aren't available!

Adjusting such a filter requires patience as everything interacts.  An examination of Figure 5 will reveal that each section is connected with jumper wires, allowing isolation of the individual tuned circuits.  Eventually, one can get a "feel" for how the adjustments interact - but it may still be necessary to  disconnect the sections and check/tune them individually back to a starting point if one gets "lost" and the response/tuning gets worse and worse.

Also visible in Figure 5 are shields around the large tuning elements made from pieces of double-sided copper-clad PC board material.  While shielding between the sections isn't really necessary from a performance standpoint, placing the filter - which was constructed on the lid of the Hammond 1590D box - into the box itself causes the filter to detune slightly due to proximity to the enclosure's walls:  The shielding on the sides of the large coils - and the bars across the top - "simulated" the filter being within the die-cast box and almost eliminated the effect, but still allowed access to permit adjustment if the large coils.

Figure 5:
As-built 40 meter band-pass filter.
This filter was constructed on a solid copper ground plane of circuit-board material.  to hold components in place and to isolate junctions from the ground.  "Manhattan" (island) pads were used for junctions that needed to be isolated from the ground:  The "Me Pads" (from "QRPMe") were used.  Blobs of RTV are used to mechanically support some of the larger components.
Click on the image for a larger version.


To be clear:  This should NOT be your first band-pass filter as it is VERY tricky to adjust - and you MUST have available a scalar and/or vector network analyzer to properly adjust it!  If both of these do not apply to you it is suggested that you obtain help - or prepare to get this gear and pull your hair out during adjustment!

Did it work?

The answer is Yes.

Figure 6:
A sample passband of the filter during adjustment:  The ultimate adjustment resulted in a somewhat flatter response.
The "upper" notches (L2/C2 and L5/C5) can clearly be seen as can the upper "lower" notch (L6/C6).
The intrinsic insertion loss, including the two 2 dB pads, is around 15dB.  The ultimate rejection is around 65 dB, correlating to a filter rejection of around 50 dB, taking into account the through losses.
Click on the image for a larger version.
This filter offers over 20dB of (additional) attenuation below 6.9 MHz and above 7.4 MHz and between careful adjustment of the receive system gain (e.g. just enough signal to comfortably "hear" the noise floor during the quietest part of the day) and the attenuation of the 41 meter signals, overload on the 40 meter receivers on the Northern Utah WebSDR #4 no longer occurs.  If you wish, you can check it yourself, particularly during the evening "gray line" hours when sunset is sweeping across North America at


The use of a similar filter in ITU Regions 1 and 3:
In Regions 1 and 3 the 40 meter amateur band covers 7.0-7.2 MHz with strong SWBC signals starting at 7.2 MHz.  Narrowing this filter to 200 kHz would require a redesign and would further-push the limits of standard components, but broadly similar results should be possible.  Alternatively, the center frequency of this filter design could be moved down by 100 kHz to 7.05 MHz and offer similar rejection to signals above 7.2 MHz.
Options for even "sharper" filtering:
While the filter described is starting to push the limits in terms of the use of reasonably-obtainable components, there is another option:  A frequency-converting band-pass filter.  For this, a local oscillator and a pair of mixers would be used to convert the 7.0-7.3 MHz 40 meter passband down to a lower frequency where a "sharper" filter would be easier to construct.
For example, using an 8 MHz oscillator would convert the 40 meter band from 7.0-7.3 MHz to 0.7-1.0 MHz, inverting the frequency, meaning that the most critical part of our filtering - that "above" 7.3 MHz - would now be happening below 700 kHz.  Of course, this "converting filter" would have to have decent band-pass filtering of its own to prevent response to undesired signals and the mixer used for the down-conversion would have to be adequately "strong" to withstand the insanely strong 41 meter signals.

This page stolen from