At this time "high power" LEDs were starting to appear on the market at reasonable prices, and by "high power" I mean LEDs that were capable of dissipating up to 5 watts, each. What this meant was that from a single emitting die of rather small dimensions one could pump into it enough current and, with the good efficiency of the device, obtain a quantity of light that was suitable for long-distance optical communications.
To be sure, I was building on the fine pioneering work of others, including that of two Australians, Dr. Mike Groth (VK7MJ) and Chris Long (now VK3AML) who had determined that it was the noncoherent light produced by LEDs that offered the greatest probability of practical, very long-distance atmospheric optical communications. (As a primer as to why this is the case, read the article Optical Communications Using Coherent and Noncoherent Light - link).
Optical receiver needed:
In the midst of producing the various pieces of equipment required for experiments in optical communications (e.g. optical transmitters, modulators, receivers, support equipment, etc.) I was investigating the different circuit topologies of practical optical detectors. My goal was not to achieve extremely high data speeds, but rather to use audio-frequency signalling (speech, tones) to start with and, perhaps, work up from there.
One of most common such detectors is the phototransistor - but I quickly dismissed that owing to its very small photoactive area and the fact that the various pieces of literature relating to weak-signal optical detection noted that they are very noisy in comparison to practically any other device owing to their intrinsic noise level. (CdS cells - article here - were not seriously considered because they are too slow to respond - even for audio frequencies.)
One option was the venerable Photomultiplier Tube (article here) and while this was technically possible and, in theory, the best choice, it was ruled out because of its fragility (electrically and mechanically), its large size, the limited response at the wavelengths of interest (more below on that) and the need for a high voltage supply (around 1000 volts).
While these technical difficulties are surmountable I could not overlook the fact that available literature on these devices - and advice from the Australians, who'd actually used them - pointed out that there were but a few photomultipier tube types that have good sensitivity in the "red" end of the optical spectrum where there is also good atmospheric transparency - and even fewer of these rare types, in known-usable condition, available for a reasonable price on the surplus market!
|Figure 1: |
The transimpedance amplifier in its simplest form.
This circuit converts the photodiode currents
into a proportional output voltage.
The Transimpedance Amplifier:
This left me with the photodiode (article here) and the most commonly-seen circuit using this device is the "TIA" - TransImpedance Amplifier (article here). As can be seen from Figure 1 this is very simple, consisting of just an operational amplifier with a feedback loop with the photodiode connected directly to the noninverting input. In this circuit the photodiode currents are converted directly to voltage (hence the name) with the gain set by the feedback resistor with the added capacitor being used to assure stability, compensating for photodiode and op-amp capacitance.
This particular circuit has the advantage that it is very predictable and the frequency response can be determined by the combination of the bandwidth of the op amp and the intrinsic capacitance of the phototransistor. To a degree, one can even increase the frequency response for a given set of devices by reducing the feedback, but this comes at the expense of gain and ultimate sensitivity.
In other words: With photodiodes you can have high sensitivity, or you can have wide bandwidth - but not both!
While very simple (there are even single-chip solutions such as the "OPT101" that include the photodiode, amplifier, and even feedback resistor in a clear package) there are some very definite, practical limitations to the ultimate sensitivity of this sort of circuit if the goal is to detect extremely weak, low-frequency currents. When you get to very low frequencies, "1/f" noise (a.k.a. "flicker noise") becomes dominant from a number of sources and there are various other types of noise sources (thermal, shot, etc.) that can be produced by the various components.
As it turns out, this circuit - with practical op amps - has very definite limitations when it comes to trying to divine the weakest signals at low-ish frequencies (audio, sub-audio): For an article on why this is so - and some of the means of mitigation - see the January, 2001 Electronic Design article, "What's All This Transimpedance Amplifier Stuff, Anyway?" - link by Robert Pease.
cascode circuit - link (e.g. current-based) amplifier with subsequent follower stages. Additionally, the photodiode itself (D1) is reverse-biased, reducing its capacitance significantly and thereby improving high frequency response. By hand-selecting the quietest JFETs one can obtain excellent performance with this circuit and since it is discrete, there is room for adjusting values as necessary to accommodate component variations and for experimentation.
This particular circuit is quite good across the audio range from a few 10's of Hz to several kHz, but above this range it is largely the capacitance of the photodiode (at least for devices that have square millimeter-range surfaces areas) that quashes the high frequency response. Even though the photodiode's capacitance - and that of stray wiring and the JFET itself - may be only in the 10's of picofarads, at hundreds of k-ohms (or megohms) even small amounts of capacitance quickly become dominant - another good reason to implement the aforementioned cascode circuit and its tendency to minimize the "Miller Effect" - link to help optimize frequency response.
The K3PGP circuit and variations:
|Figure 4: |
The K3PGP Optical receiver.
Click on the image for a larger version.
Building the above circuit as a "reference" I began testing on a "Photon Range" - a darkened room in my basement with a red LED affixed to the ceiling - where I characterized the various receiver topologies. In this environment a small and adjustable amount of current (10's of microamps, typically) would be fed to the LED, modulated at an audio frequency, and the receiver under test would be placed on the floor below with its output connected to a computer in an adjacent room running an audio analysis program such as "Spectran" or "Spectrum Lab" to measure the signal-noise ratio at different frequencies. Before and after each session I would measure the performance of my "standard" optical receiver - the VK7MJ circuit - and use it as a basis of comparison.
The receiver named after K3PGP (see his web site - link) was the next receiver to be tested. This receiver is much more sensitive than the VK7MJ receiver - at least at very low audio frequencies (<200 Hz) and as may be seen in Figure 4 it is devoid of a feedback mechanism and the connection between the photodiode and JFET is made directly, with no external biasing components of any kind.
While a seemingly simple circuit, there is more going on here than one might first realize: Without any feedback or any other components between the FET and photodiode the opportunity to introduce noise from such components or reduce the signal from the photodiode in any way is minimized. In fact, when constructing this circuit there is the strong admonition that the photodiode-gate connection to the JFET be done in mid-air (and that one clean both components with alcohol to remove residue!) as leakage paths on circuit board material can cause significant signal degradation!
Effectively, the K3PGP circuit acts as a charge integrator with the energy slowly (in relative terms) bleeding off due to the leakage of the photodiode, its photoconductivity, and the gate-source leakage currents of the FET itself. While extremely sensitive at low frequencies - specifically those below 200 Hz - above this, the sensitivity and output suffers due to the rather long R/C constant associated with the high gate-photodiode leakage resistance and capacitance and, to a lesser degree, the Miller effect. This circuit also functions only in total and near-total darkness conditions: More light than that, the voltage across the photodiode reaches equalibrium while turning the FET "on", effectively quashing the signal.
Inspired by the above circuit I made the modification indirectly depicted in Figure 5, below:
This circuit was devised as a "test bed" and although not shown in the diagram, it was configured by connecting the cathode of the photodiode to the gate and grounding the anode and having no other photodiode-gate connections present - just as in the K3PGP receiver.
In this circuit one has a FET input and a cascode circuit - just like that of the VK7MJ circuit - to reduce the Miller effect, but this particular cascode circuit has a modification: Q3 forms a current source, in parallel with the cascode, that supplies the bulk of the drain current for the JFET - several milliamps. Because the amount of current provided by the current source - which has a high operating impedance and is largely "invisible" - is fixed (but adjustable by varying R4 to suit specific characteristics of Q1) and it is left up to the cascode to supply the remaining drain current - which varies depending on the gate voltage. In this particular circuit, due to the "cascode action" the voltage at the drain of Q1 and emitter of Q2 varies very little while the cascode - which is allowed to bias itself at DC, but is bypassed at AC with C3 - produces the recovered modulation at the collector of Q2, greatly amplified. From the collector of Q2, noninverting amplifier U1a amplifies the signal further and presents a low-impedance output.
In other words, it is mostly the K3PGP circuit, but with a cascode amplifier and higher FET drain current: By reducing Miller capacitance with the cascode the frequency response was to be improved somewhat and by increasing the drain current the noise contribution of the FET itself should be reduced as noted in the Pease article mentioned above.
In testing it was observed that this particular circuit was, in fact, several dB more sensitive than the original K3PGP circuit and also that the frequency response was slightly better - but not as much as one might first think, mostly owing to the fact that it is mostly the photodiode capacitance that is limiting the response rather than the Miller effect - but every little bit helps!
I then rewired the circuit using the "Standard Config" noted in Figure 5 which, if you draw in the lines, converts it into a TIA circuit like that of the VK7MJ design with both adjustable reverse bias of the photodiode and adjustable feedback. In this configuration the performance at very low frequencies was reduced, likely due to the noise contribution of the feedback resistor, increased leakage currents from the photodiode at reverse bias and also signal attenuation caused by the feedback submerging the lowest-level, low-frequency signals into the noise. At "speech" frequencies it was slightly better than that of the VK7MJ receiver - probably due to the higher JFET current or, perhaps, random component variances - and the frequency response was also comparable to that of the VK7MJ circuit, the parameters varying according to the amount of applied feedback and compensation.
Improving the receiver:
My goal was a circuit that offered the sensitivity of the K3PGP circuit, but usable speech response - the latter not being available from the K3PGP circuit due to the R/C rolloff. A quick check revealed that this was the typical 6dB/octave rolloff so I reconfigured the circuit, again, as a K3PGP-like circuit and followed it with an op-amp differentiator circuit with a breakpoint calculated to compensate for the measured "knee" frequency (e.g. that at which the 6dB/octave rolloff of the K3PGP circuit) began - the result being that I now had a fairly flat frequency response. Not unexpectedly, while the signal-noise ratio was quite good at the very low frequencies, it decreased fairly quickly as it went up as that energy was simply submerged in the circuit noise.
In staring at the circuit, with the grounded anode of the photodiode, I wondered about reverse-biasing the photodiode to reduce the capacitance - but if I did this, how would I keep the voltage at the gate from rising without needing to add another (noise generating, signal-robbing) component to clamp it to ground? Knowing that the gate-source junction of a JFET was much like that of a bipolar transistor in that there would be an intrinsic diode present, I knew also that the gate-source voltage would limit itself to 0.4-0.6 volts, but how would the FET behave?
Using JFET Gate current for "good":
In doing a bit of research on the GoogleWeb when I derived this circuit I could not come up with any sort of useful answer to the "gate current" question, so I simply did it: The photodiode was reverse-biased with the minute leakage, dramatically reducing its capacitance, and photoconducting currents being sinked by the gate-source junction. As expected, the drain current increased noticeably, but the circuit worked extremely well, with both frequency response and apparent gain increasing dramatically!
Putting this "new" circuit back on the photon range I observed that although its low frequency (<200 Hz) sensitivity was slightly worse than that of the K3PGP circuit (see comment below), the higher speech-range frequencies (300-2500 Hz) were, on average, 10-12dB better than the VK7MJ circuit and approximately 20 dB better than the best, low-noise op-amp based TIA circuit that I'd built to date!
In analyzing the circuit, there are several things happening:
- Reverse bias of the photodiode: This reduces the capacitance - typically by a factor of 3-6, depending on the specific device and voltage applied.
- The photodiode will produce current in the presence of light.
- Being reverse-biased, the photodiode will also operate in a photo-conductive mode, passing current from the bias supply in response to light.
- With the gate-source junction conducting, the reverse bias across the photodiode is maintained since the gate-source voltage will never exceed 0.4-0.6 volts.
- As described above, the amplifier is connected in "cascode" configuration to minimize Miller effects.
- There are NO other components or signal paths connected to the photodiode-gate junction that can contribute noise or attenuate the signals.
- In parallel with the cascode circuit is a current source which provides a high-impedance current source to increase the JFET's bulk currents, further reducing its noise.
- The "diode action" of the gate-source clamping seems not to be a significant contributor of noise - at least at "dark" currents of the photodiode.
- There is little or no documentation about using a JFET this way, anywhere else!
It is likely that the main reason that this doesn't perform quite as well at the K3PGP circuit at low (<200 Hz) frequencies is because of the intrinsic leakage current noise endemic to the reverse biasing of the photodiode, particularly in a "1/F" manner: At higher frequencies where this sort of noise falls away it performed far better.
In "photon range" testing it was difficult to tell at which frequencies the K3PGP receiver performed better. My K3PGP exemplar receiver was certainly better at, say, 20 Hz, but even at 100 Hz or 60 Hz it was a difficult call to make. At such frequencies and under such conditions careful selection of the "quietest" photodiode and FET can make a significant difference and with most of these circuits, reducing their temperature - while somehow managing to avoid condensation - can help even more!
Plotting Gate current versus Drain and Gate voltages:
Later, I constructed a test fixture to analyze the gate-source voltage and gate-source current response of a 2N5457 JFET and plot this against the drain current - see Figure 6 below.
The relationship between drain current and
Vgs= Gate-source voltage
Vp=FET Pinch-off voltage
Idss=Zero gate voltage drain current
According to typical JFET models, in the saturation region the FET operates such that the drain current is generally independent of the drain voltage as can be seen in Equation 1 and the graphs in Figure 6 indicate that this seems to be true even when the gate-source junction is conducting.
So, now we know what is happening. At first glance, one might presume that with this diode in conduction that the logarithmic response would make the circuit unsuitable for general audio recovery - but this is not so: At very low light levels the detector has lower than 1% harmonic distortion.
Putting this into practice:
The circuit depicted in Figure 8 was developed for speech-bandwidth optical communications use.
As can be seen, this looks very similar to the circuit of Figure 4 with the exception that the reverse-biased photodiode is connected to the JFET and that there is the added circuit, U1b, that forms a bandwidth-limited differentiator - the component values chosen to approximately correlate with the low-frequency "knee" of the BPW34 photodiode and also to cease its frequency boost above 5-8 kHz. (The "Flat" audio output, uncompensated by the differentiator for the 6dB/octave rolloff, is provided for both very low frequency - below 200 Hz - and high frequency - above 5 kHz - signals to be applied to a computer for analysis.)
The circuit in Figure 8 - and minor variations of it - have been replicated many times over the years using different components. The important considerations are that both Q2 and Q3 be low-noise, high-beta transistors such as the MPSA18 (or 2N5089) and that the JFET used for Q1 be capable of rather high drain current. In the original design, the 2N5457 was specified as this device is better-characterized that many other, similar FETs and is capable of quite low-noise operation: The more common MPF102, with its extremely wide variation of parameters, might be suitable if an appropriate device is "cherry picked" from amongst several based both on high zero gate-source voltage drain current and tested "noisiness". A more modern JFET is the BF862 - available in surface-mount only (as are most JFETs these days!) - that is even better for this application than the 2N5457 and capable of much higher drain ("bulk") current to the point where utilizing its full potential might compromise 9-volt battery life!
In a circuit such as Figure 8, above, the drain-source voltage will be much lower than one might initially expect - on the order of 0.2-1.0 volts for a JFET such as a 2N5457 and between 0.1 and 0.5 volts for the BF862 - but this is normal operation. While the setting for Q3 current, adjusted via R5, (in Figure 8) at 120 ohms is suitable for most 2N5457 devices, the current may need to be reduced (e.g. R5 increased in value to 180 or 220 ohms) for some "lower 0 Vgs" current devices such as the MPF102. In general, the higher the drain current, the lower noise contribution from the FET - but if you exceed the "magic" value and attempt to force too much current, the circuit will suddenly stop working: Overall it is better to have a bit lower drain current than optimal and have a little bit more noise than to have too much drain current! (Don't forget that the properties of the current source and the JFET itself will also change with temperature - but they generally seem to track.)
Interestingly, the circuit depicted in Figure 8 also works in daylight, albeit with some caveats.
When very high levels of light are present, the photoconductivity will shunt the reverse bias to the gate-source junction, and the frequency rolloff "knee" associated with the photodiode capacitance will shift upwards due to photoconductive shunting causing the audio to become "tinny". The audio will also become somewhat distorted owing to the different light-to-audio transfer curve that occurs under such conditions, in which case the frequency response of the audio on the "flat" output is more suitable than otherwise. In such situations one does not really need the high sensitivity of this type of receiver, anyway, and a typical TIA circuit with AC coupling such as that depicted in Figure 2 or Figure 3 could be used or one could apply optical attenuation in front of the detector to reduce the light level.
Entire web pages could be written (and have been - see the Modulated Light web site - link) about through-the-air, free-space optical communications over long distances (well over 100 miles, 160km) using both LEDs and low-power lasers, but even the most sensitive receiver - no matter the underlying technology - requires supporting optics (lenses!) in order to function properly: It is through such lenses that 10's of dB of noiseless signal gain may be achieved, not to mention directionality and the implied rejection of off-axis light sources.
The circuits described on this page are likely to be suitable only for speech frequencies and low-rate data but this is, in part, due to the medium involved (the atmosphere) and method of transmission. At the extreme distances that have been achieved with the above equipment (>173 miles, 278km) the signals are weak enough that only low-rate signalling techniques would likely be feasible under typical conditions at safe, practical optical power levels.
Additional web pages on related topics:
- Modulatedlight.org - This web site has a wide variety of information related to amateur, free-space optical, through-the-air communications.
- Optical Receivers for Low-Bandwidth, Through-the-Air Communications - This is a reference article that gives additional detail about the design of the circuits discussed in this article.
- Using Laser Pointers for Free-Space Optical Communications - This describes how one might use low-power laser pointers for low-rate optical communications, the practicalities of various circuits, the methods and the realities.
- The Modulated Light DX page - This page has several articles describing the practical aspects of free-space, through-the-air optical communications including various atmospheric effects.
This page stolen from "ka7oei.blogspot.com".