Thursday, January 16, 2020

Improving the stability and performance of the FiFi SDR receiver

At the Northern Utah WebSDR link, among one of the several receiver configurations is that where a "SoftRock" receiver is used with a sound card.  This combination works very well - far exceeding in overall performance, especially dynamic range, almost any available "wideband" receiver including the SDRPlay, Red Pitaya and KiwiSDR Note 1 with the caveat that only a bandwidth equal to the sample rate of a sound card - 192 kHz or so maximum - can be covered "per band".
Figure 1:
One of the three FiFiSDRs obtained for use at the Northern Utah WebSDR.
Click on the image for a larger version.

Problems with the USB Sound Cards:

Up to now, we have been using a combination of plug-in (PCI, PCIE) sound cards, using USB sound cards when the number of such receivers exceeded the number of plug-in slots on the computer.  Among the few affordable USB audio devices that can sample at 192 kHz are the Asus Xonar U5 and U7 (including the MK 2).

These devices work very well for the task - when they work:  After nearly 2 years, only two of the ten U5 and U7 devices that we had acquired over that period still work, the majority having failed when the USB interfaces would fail to negotiate at full USB 2.0 speed (if they negotiated at all!) after a few months of operation - often after a reboot.  Unfortunately, an Internet search revealed that this is not an uncommon problem and a plausible explanation as to the reason for these failures - or a fix - was not to be found.

An alternative:

Rather than spend more money on the unreliable Asus U5 and U7 USB devices - most of which we had gotten on EvilBay, originally in proper working order - we decided to switch to the "Fifi SDR" device from Box73.com in Germany.  Originally introduced around 2010, the current "Version 2" increased the bandwidth from the original 96 kHz to 192 kHz - and rather than just a sound card in a box, the FiFiSDR includes an entire synthesized "Softrock" receiver with decent performance and best of all, they cost about the same as a brand new 192 kHz capable USB sound card.  Several popular WebSDR systems - including KFS in Half Moon Bay, CA - use these devices and have reported good performance and reliability.  If they had been available when we were acquiring the equipment for the Northern Utah WebSDR, we would have started using them earlier.

Three FiFiSDRs were ordered - exactly enough for our needs - and upon arrival, I assembled them (a bit of soldering and mechanical assembly) and began to test them.  For whatever reason my Windows 7 machine at my workbench steadfastly refused to recognize the Fifi's sound card interface, but my Windows 10 laptop did and after a few missteps - mostly related to the program I was using to interface with it (HDSDR) having been previously configured for different SDR hardware - I got all three up and running.

Initial impression:

My initial impressions of the performance of the three FiFiSDR were generally good - ignoring the "elephant in the room" discussed below:  The receiver sensitivity, although varying by 2-3 dB between receivers, was within advertised specifications and once they had been powered up for several minutes the frequency was quite stable (within a few Hz).  Immediately, I noticed a few low-level CW spurious signals, but these were at or below the microvolt level and would likely be submerged in the noise floor - at least on the lower bands.

Because the sound codec was integrated within the receiver itself, the center-frequency (so-called "Zero Hz") noise was quite low because a possible pick-up point (e.g. a cable going from the output of the SoftRock receiver to a sound card) has been eliminated.  I did notice a fairly strong artifact at or near zero Hz - likely a DC offset with a bit of 1/F noise - but this is typically removed by a low-frequency high-pass filter in software and is not likely to be an issue.

I did notice two artifacts typical of "SoftRock"+sound card receivers:
  • Under no-signal conditions, the noise floor would rise by several dB at "high audio" frequencies as manifest by a slightly "lighter" waterfall at the extreme low and high ends of the 192 kHz passband.  This is quite typical of sound cards and has been observed on nearly every sound card that I have used.
  • Under conditions where the external (ionospheric) noise exceeded that of the receiver's noise floor, there was a bit of "droop" at the extreme low and high ends of the 192 kHz passband.  I've noticed this effect on nearly every softrock-type receiver and attribute it largely to signal drop-off in the audio chain at high audio frequencies.
I did see something that alarmed me:  Unstable spurious signals that drifted about (the "elephant" mentioned previously) but a "fix" for this problem is pretty easy and is described later.

Static sensitivity!

I'd seen a mention or two that the Fifi SDRs would occasionally "lock up" - but it didn't seem to be a common theme in the groups online - but once I got the them up and running, I saw two things that concerned me:
  • When I touched the metal case and had no antenna connected the receiver's noise floor went way up.  This is bad news - particularly if it is to be installed in an electrically-noisy environment - like anywhere near a computer.
    Figure 2:
    The sole "official" case-to-board grounding point is at the corner near
    the 3.5mm jack
    Copper foil was wrapped over both the top and bottom and soldered
    to the board's ground on both sides with a bit of excess
    folded over on the end to provide a connection to the end plate.
    (The solder connection was re-done, but I didn't get a picture.)
    Click on the image for a larger version.
  • If I had even the slightest amount of static electricity on my body the Fifi SDR would crash when I touched it and refuse to come back to life until I unplugged the USB and plugged it back in again.
Upon observing either of these, my suspicion as to the problem was verified with an ohmmeter - the metal case was, in no way, connected to the internal PCB.

Inspection revealed why:  Not only was the case very heavily anodized, the two boards fit somewhat loosely in the slots inside - and there was only a single common ground in one corner of the main board.  Even if the board was snug, it probably would not have made electrical connection through the case's oxide coating.

Remedy:

Clearly, I needed to find a simple way to bond the board to the case.  The receiver's  main board's use of a single-point ground seemed reasonable - particularly when one pairs a computer with a very sensitive receiver as one must carefully avoid on-board ground loops - so I resisted any temptation to "bond it everywhere" - at least not without careful testing.
Figure 3:
The point where the copper foil makes contact with the end plate.  Note that
the ends of the drawn-aluminum case have had the oxide layer removed to
bare the metal:  The same was done on the end plates to allow the copper
to make contact.  The bared ends of the case and corresponding parts of
the end plates were coated with a light layer of anti-oxidant.
Click on the image for a larger version.

First, I used a rotary tool with a wire wheel to remove the anodization (clear oxide coating) from around the edges of the end panels as well as the very ends of the drawn aluminum case so that when the screws were installed, they would have metal-on-metal contact.

The next step was to provide a connection from that ground in the corner to the case - and I did this by wrapping the edge of the board in that corner  (Figure 2) with copper foil and soldering it, leaving a bit of excess to wrap around the end of the board (Figure 3) - the idea being that it would be compressed by the now-bare aluminum end panel and make connection to the rest of the case.

Because bare aluminum quickly forms its own insulator when exposed to atmospheric oxygen, a thin layer of anti-oxidant compound (e.g. "NoAlOx" or "DeOxIt" - both used in electrical wiring) was applied to the bared aluminum on the ends of the case as well as where the copper would press against it:  This would prevent re-oxidation and the loss of connection over time and with exposure to air and moisture.

Result:

Upon reassembly, the end of the board with the copper connection was tightly pressed against the inside of the end panel and there was a low-resistance connection between the boards inside and the case.  Because of this effort, not only does the receiver noise floor not go up when I touch the case, but I can give the unit a pretty good "zap" with a static spark and not have it affect the operation of the device!

If I'd had the time to do so, I would have tested the efficacy of additional board-to-case ground points, making sure that these additions did not reduce the performance of the receiver.

Comment:
Although not instructed to do so by the assembly guide, installing the nut on the 3.5mm audio connector may the connect the board to the case (via the end-plates) at that point. As they are, the connector's body is not is not long enough to protrude very far through the end plate and only a few threads were presented.
Figure 4:
On the top, an 80 volt gas-discharge tube.  Holes were drilled on the top to
provide connection and mounting.  The tube is bent away
from the case to prevent it being shorted to the case.
When drilling, take care to avoid intercepting any traces on either
side of the board.
Click on the image for a larger version.
Because of the thickness of the end plates, the installation of the nut will prevent proper insertion of a cable into the connector - but since I wasn't planning to use that jack, I installed it anyway.

The "floating" antenna jack:

Presumably to prevent circulating currents between the antenna system and the "ground" of the computer (via the USB cable) the antenna jack is coupled to the receiver via a 1:1 transformer.  Having a "floating" antenna connector made me nervous:  If one were to connect a FiFiSDR to an ungrounded wire antenna, wind static could easily cause high voltage to appear on the antenna connector which might not only cause a shock, but if it arced to ground somewhere - possibly within the receiver - it could damage the receiver's RF amplifier and/or be conducted to the USB interface to the computer where it could cause the FiFiSDR and/or computer to crash or worse, cause damage.
Figure 5:
The 150k resistor on the bottom of the board to drain static.
Any value between 47k and 220k would suffice.
For another modification (described below) 0.1 and 0.001
capacitors were soldered across this resistor for RF bypassing.
Click on the image for a larger version.

While I am not advocating using any antenna without appropriate grounding, I do know that it does happen and out of principle, I added circuitry to mitigate the risk:  The addition of a 150k resistor to prevent the accumulation of charge and an 80 volt gas-discharge tube between the RF and system ground.

Even if one does not add a gas-discharge tube, I would certainly advocate the addition of the drain resistor!


* * * * * * * *

Spurious signals in the receiver - the "elephant":

Having gotten all three receivers operational, I noticed something else that was alarming:  In two of the three receivers I could see, near the upper and lower edges of the passband (192 kHz sampling rate, 75 kHz and farther, symmetrical about the center frequency) some "ragged" signals that drifted about:  The third receiver also showed these same spurious signals, but they were much weaker, closer to the center frequency indicating that the same problem was evident, but likely farther out of the +/- 96 kHz passband of the receiver's sound card and showing up via aliasing.  Even if though the spurious signal on this third receiver was weak, I decided that was likely to mix with existing signals and cause additional, undesired signals to be produced within the passband, and would likely be the case with the other two receivers.

The spurious signals on the two worst receivers (example in the upper half of Figure 6) were fairly strong, about "S-9" in strength, making their existence unacceptable.  A quick check of the FiFiSDR wiki and trouble ticket system revealed that there were at least two tickets (#324 and #332reporting this issue - but both were years old and were still open with no suggested resolution.

Figure 6:
A screen shot from HDSDR showing the spurs before the modification (top half) and after the modification (bottom half).
These spurs are symmetrical about the center frequency (red line in the middle) indicating that they are NOT at RF, but rather at a point beyond the RF mixer in the audio chain.  The amplitude of these spurious signals make this receiver nearly unusable due to their strength.
Because of the rather low apparent signal strength, it is most probable that the actual frequency of these oscillations is not in the 0-96 kHz range, but much higher and being made visible because of the finite attenuation of the codec's low-pass filtering and aliasing of the A/D converter's sampling rate.  
Click on the image for a larger version.
The symmetrical nature of this signal - and the fact that its nature was completely independent of the receive frequency - indicated that the origin of this signal was not at RF, but was within the audio chain or related to the FiFiSDR's power supply.

Careful observation showed something else:  As an applied signal within the passband of the receiver (e.g. +/- 90 kHz or so of the tuned frequency) increased in amplitude above approximately -70dBm, these spurious signals would start to  "noise up" and disappear - finally vanishing by the time the signal achieved -30dBm.  Because the signal was affected by signals in the audio chain, this observation took the onus off the likelihood of the power supply oscillating, pointing directly at the audio chain indicating that whatever was causing it was directly in the audio signal path.

Figure 7:
The addition of the 470k resistor to the "ADC1LP" pin of the codec.
The ground plane was scraped and the resistor soldered between it and
the capacitor(s) as shown, taking care to avoid shorting the "ADC1LP"
line to ground.  The corresponding (original) resistor on the "ADC1RP" line
is the upper-most resistor at the left marked "474", upside-down.
Click on the image for a larger version.
Wielding an oscilloscope, I started probing the audio chain - but I could see nothing obvious in terms of unusual signals - but I noticed that when I touched the probe to pin 39 ("ADC1LP") of the audio codec - an Analog Devices AD1974 - the frequency would shift slightly. 
Touching a voltmeter probe to this pin I observed that this spurious signal would disperse widely - as if frequency-modulated by the AC mains field on the workbench - but the same did not happen when I touched the voltmeter probe to pin 41 ("ADC1RP") indicating that the problem was only on the "left" channel of the codec.  Disconnecting the DC blocking capacitor from IC5, the audio amplifier, the amplitude of this oscillation remained the same, shifting frequency very slightly:  This implied that the problem was the AD1974 itself.

At this point I noticed something else:  The voltage on pin 41 ("ADC1RP") was a few 10s of millivolts lower than that on pin 39 ("ADC1LP") - something that should not occur as both were presumably biased from the same internal voltage reference.  I then observed that there was a 470k resistor between the traces connecting pin 41 and ground - but this resistor was missing on pin 39.

On a hunch I added a 470k surface-mount resistor to ground at the bypass capacitors connected to pin 39 and found that the spurious signal disappeared.  Apparently, the designers of the Version 2 of the FiFiSDR had observed a similar problem and added the 470k resistor to "ADC1RP" - but did not do so on "ADC1LP".

This "fix" worked on all three receivers.


Improving common-mode (longitudinal) isolation:

(Sorry, no pictures at this time.)

One thing I noticed about this receiver was that its antenna input was transformer-isolated from the case.  Ideally, this is a good thing as it can reduce any ground loops that may contain circulating AC or DC currents which, unless everything is well-bonded to a common ground, may cause problems (e.g. hum on audio devices, potential USB instability).  Unfortunately, no transformer is or can be perfectly balanced - and this could be demonstrated on the KiwiSDRs by simply touching the outer shell of the BNC connector and observing a slight noise increase when in an indoor RF-noisy environment - even after the case grounding issue discussed above was solved.

To quantify this imbalance I took some measurements, applying an amount of signal between the KiwiSDR case after bonding it to the circuit board as described above and the shell of the BNC connector, observing the amount of signal that was required to achieve an "S9" reading.  Doing this at both 5 MHz and 29 MHz, I obtained the following results in terms of common-mode (longitudinal) isolation:
  • 5 MHz:
    • Isolation = 29dB unterminated
    • Isolation = 50dB terminated at 50 ohms
  • 29 MHz:
    • Isolation = 24dB unterminated
    • Isolation = 35dB terminated at 50 ohms
Whether or not this is acceptable in your situation is something that you will have to decide - but I chose to make a minor modification:  The addition of a 0.1uF and 0.001uF capacitor in parallel (two capacitors being used to provide low impedance from low to high frequencies) with each other (this was in parallel with the 150k resistor depicted in Figure 5) to bridge the "RF Ground" and the system ground:  Unlike connecting the two grounds together with a jumper, this would still provide low frequency DC and AC isolation.  The result was that there was no longer a significant difference between the readings when the BNC connector was terminated or unterminated.

With the addition of the capacitors, the isolation improved to about 50dB on both frequencies - terminated or not.  I was hoping for even greater improvement than 50dB, but I suspect that because the case-to-board mounting occurs in only one place, in a corner away from the RF connector, circulating currents were flowing across the board.  It is possible that bonding the system ground to the case near the antenna connector would have improved this - but I did not have time to test this and make sure that it did cause significant degradation.

* * * * * * * * * * * * * * * * * * * * * *

Note 1:
Unlike some of the inexpensive "Wideband" receivers based on the RTL chips (e.g. RTL-SDRS - which have just 8 bit A/D converters) higher-end receivers that have greater simultaneous bandwidth (SDRPlay, Red Pitaya, KiwiSDRs) have greater bit depths - typically 12-14 bits, offering greater dynamic range.

Even a higher bit-depth wideband receiver can be at a disadvantage compared to the combination of a SoftRock and sound card:  Not only does the 16 bit depth of a sound card offer more dynamic range, but the lower operational bandwidth (192 kHz maximum for a sound card based receiver) means that there is less overall "energy per Hz per bit" impinging on the A/D converter than a MHz-bandwidth A/D converter.
* * * * * * * * * * * * * * * * * * * * * *

Comment:  An attempt was made to post the solution to the problem of the spurious signals to the FiFiSDR ticket system, but the post was rejected by the system.  I have not had the time to register with the site.

This page stolen from ka7oei.blogspot.com

[End]


Thursday, December 26, 2019

Using TV (F-connector) 75 ohm splitters and taps in 50 ohm systems on the amateur HF, VHF and UHF bands

I recently posted several articles about using commercially-available splitters link - and making one's own splitters - link - particularly for the HF frequencies and below (e.g. 30 MHz, down to a few 10s of kHz).  A comment was posted asking about how useful inexpensive 75 ohm "TV and satellite" type splitters might be for amateur radio use.
Figure 1:
The assortment of 75 ohm TV and satellite splitters and
taps tested in this article.
Click on the image for a larger version.

Implied by this question is the use of these devices in receive-only or small signal applications:  They cannot be used for transmit purposes as putting even 100 milliwatts through one of these devices is likely pushing its power-handling capability.

I've used these devices in 50 ohm circuits before - typically for VHF and UHF (2 meters, 70cm) where, along with some attenuators, combined the outputs of multiple signal generators to do "multi-tone" testing of receivers - but the question seemed to be a good one.  Rummaging around, I gathered a bunch of devices of various manufacturers and decided to test them for insertion loss and port-to-port isolation.

Note:
Please do not ask questions like "How well does a 'brand X' splitter work over the [fill in the blank] frequency range?"
There have been thousands of makes and models of these devices sold around the world over the past several decades and I simply am not able to find, locate, and measure more than the tiniest fraction of devices that have been sold.  The information given here is expected to be generally representative of the devices available from reputable manufacturers and distributors - but your mileage may vary.
Limitations of the measurements taken:

Because my VNA (DG6SAQ WVNA) was constructed for use with 50 ohm systems (the changing of  both internal hardware components and software would be required for "proper" analysis of a 75 ohm system) I was able only to analyze them in that context - but because the question was about using them in amateur radio service - which presumes a nominal 50 ohm system - I believe that the results are still useful within the limits noted in this article.

Because the emphasis of the question was interpreted as being for amateur-band frequencies likely to be encountered by the average user, the measurement range was limited to frequencies below 1 GHz - in some cases down to 100 kHz.  The nature of the equipment and methods (e.g. 50 ohm test equipment and cabling, the use of inter-series adapters, etc.) used to test the splitters and taps increasingly limits the usefulness and accuracy of these measurements at frequencies above that of the 70cm amateur band (above 450 MHz).

The variety of splitters and taps available:

There are literally thousands of brands and models of TV/Satellite splitters and taps available on this planet - some of them from recognizable names, but most not.  For those devices from sources that might be suspect (e.g. not "name" brands from reputable suppliers) you are on your own to determine the suitability of those devices for your purpose.

Although not intended as an endorsement per se, it has been observed that devices marketed by Holland Electronics appear to consistently meet their stated specifications and is one of the few brands that is likely available worldwide from a number or different sellers - including Amazon - and major suppliers of electronic components and TV/satellite supplies.

Over the years I have seen many dozens of brands and models of these devices - and the vast majority of them are what they are purported to be, but I have run across some devices that claimed to be splitters, but were simply a box with wires connecting the ports together.  In many cases, the casual user would not have noticed anything amiss, but using several of these faux devices in a larger system would certainly result in cumulative signal degradation (e.g. "ghosting" of analog signals, degrading of quality - but not necessarily signal strength - of digital signals).

General types of devices:

There seem to be three general types of these devices out there:
  • "TV" and/or "VHF/FM/UHF" and/or "CATV" - These devices are typically designed to operate over the range of off-air TV stations across the world and the frequencies typically found on receive-only cable TV (with no Internet), encompassing the frequency range of about 40 MHz through 700 MHz, more or less.  While useful for use on the amateur bands from 6 meters through 70cm, inclusive, their usability on HF or above this range is limited as noted in the testing, below.
  • "Satellite" splitters - These devices are typically designed to operate starting at about 900 MHz, often extending to 1500 or as high as 2500 MHz, depending on the vintage and intended use.  These devices are not usable on the 70cm amateur band frequencies and below.
  • "TV/CATV/Satellite"- These devices are of a bit more recent vintage and are designed to accommodate a very wide range of frequencies - often from about 5 MHz through and above 2000 MHz - a band that includes off-air, cable and "L-Band" satellite signals - plus the "reverse" channels (sometimes called the "T" channels) often used by "cable Internet" modems that reside below 45 MHz.  These are the most useful to amateur service and can often be used on HF through 70cm.
If you do not see a specific frequency range noted on the device itself, assume the worst-case, smallest frequency range that covers that usage category - unless you can test them yourself.

* * *

General findings

For the TL;DR types, here is a summary of the results of the measurements described in more detail farther down the page.


Using 75 ohm devices in 50 ohm systems:

The most obvious issue is that TV-type consumer devices are almost universally equipped with type "F" connectors which means that one must use either an adapter or use a cable with an attached "F" connector.
Figure 2:
Left to right:  Two BNC female to male F connecitrs with an
F-type 75 ohm terminator on the right.
Click on the image for a larger version.

For receive-only systems, it's not too uncommon to simply use a 75 ohm cable like RG-6 - which is quite low loss and very inexpensive - to connect a 50 ohm antenna to a 50 ohm receiver.  The effects of this apparent "mismatch" are typically minimal as most receivers are only "approximately" 50 ohms, anyway.  In theory, the use of 75 ohm cable on 50 ohm devices will result in a 1.5:1 mismatch and commensurate losses, but this sort of mismatch is commonly observed on many antenna systems that are ostensibly designed to operate at 50 ohms and is usually of minor consequence.

When using an inexpensive cable like RG-6, it's worth noting that most of these cables use copper-coated steel (CCS) center conductors which may have implications for DC resistance of power is being sent on this cable (for a preamplifier, converter, controls) as this type of cable will have far more total resistance than one with a solid copper center conductor.  Copper-coated steel center conductors may also have implications in terms of skin effect at low frequencies (low HF and below) - but this is beyond the scope of this article - see, instead, this article by Owen Duffy.  There exist cables with copper-coated aluminum (CCA) center conductors that have lower DC resistance that CCS cables, but they tend to be more fragile due to the tendency of the aluminum center conductor to become brittle with flexure.

The device itself (splitter, tap) is designed primarily for 75 ohms and this means that its performance will be somewhat degraded in a system that is "completely" 50 ohms (e.g. 50 ohm cables with F-connector adapters) but these effects are largely as follows:
  • The "through" loss may be slightly higher.  In the case of a 2-way splitter, the ideal loss will be 3dB - but even at the proper impedance, it will be slightly higher than this due to component losses, typically in the area of 3.5 dB.  Practically speaking, the main effect of using a 75 ohm splitter in a 50 ohm system was a slight change (only a few tenths of a dB) in the loss.
  • Reduced isolation between ports.  The most obvious effect on splitters was that the isolation between ports (e.g. the "out" ports of a 2-way splitter) was reduced.  Compared to some specialized splitters, the isolation of inexpensive, consumer-grade "TV" splitters is lower overall.  As can be seen from the graphs, below, operating in a 75 ohm system resulted in better isolation - sometimes over 40dB at certain frequencies - but this assumes that all loads and sources are well-matched to 75 ohms, something that is not likely to be the case in a real-world installation.  Typically, isolation reduced to something in the 20dB area when operated in a 50 ohm system.  In many cases, this is "good enough".
  • In splitters and taps, resistors are major components in determining their "native" operating impedance.  For example, a 75 ohm splitter or tap, depending on design, may have a 150 ohm or 37.5 ohm (2 times and one-half 75 ohms, respectively) resistor contained internally.  In theory, changing this resistor to a value appropriate for 50 ohms (typically 100 or 25 ohms) would optimize performance at 50 ohms - but doing this may or may not be worth the trouble. 
In short:

Unless your situation requires precision, the use of inexpensive, TV-type splitters and taps of the types described on this page will yield "reasonable" performance over the design frequency range - provided that the device is constructed as described by a reputable manufacturer.

The use of a (nominally) 75 ohm device in a 50 ohm system will require using connectors that are not normally used in 50 ohms systems (typically "F" connectors) which means that adapters of some sort will be needed - the expense, bulk and inconvenience of which must be considered in the overall design.

Finally, note that the above comments are for the general case:  Remember that your needs, requirements and results may vary and that you must do your own analysis and testing to verify that such components are appropriate in your specific case.

* * *

Plots of various devices:

Below are selected plots of devices representative of the types on-hand.  In general, devices with similar stated ratings performed in the same manner.  In all of these plots, the insertion loss is represented by the blue line while the complex impedance data is depicted on a Smith chart in the middle:  Numerical data at the frequencies indicated by markers is seen in the lower-left corner of the screen.  Again, remember that at higher frequencies, the nature of the 50 ohm test system, connecting cables and adapters will increasingly skew the results - particularly those depicted by the Smith chart.

The interpretation of a Smith chart will not be covered here, but there are many online resources that describe its use including this video in a series on this topic by W2AEW on his YouTube page.

A "satellite" splitter:
Figure 3:
The "through" loss of the HFS-2 splitter represented by the blue line
across the top.
Click on the image for a larger version.

This device - a "Tru Spec HFS-2" is representative of those intended for use on an (older) L-band system found in satellite receive systems, having on its label a "900-1500" MHz frequency range.  As noted above, the limitation of the measurement set-up made measurements above the 70cm amateur band (in the 440 MHz area) suspect - but the object here was to see if it was usable below that range.

At initial glance, the "through loss" of this device below 900 MHz (Figure 3) might seem to indicate that it worked below this frequency, but notice that at lower frequencies (below 50 MHz) indicates a loss less than 3dB indicating that it is not working as a proper 2-way splitter.  A look at the isolation plot (Figure 4) tells more of the story.
Figure 4:
Isolation between ports of this splitter.
Click on the image for a larger version.

As can be seen, at about 900 MHz and above, the apparent isolation between ports is reasonable but at 2 meters (146 MHz) it is only 3dB verifying the fact that at these lower frequencies, it less a proper splitter, but more equivalent to a device where the three ports are connected with a piece of wire.  The apparent isolation increase at low HF is more likely an artifact of its construction - the insertion loss being below 1 dB (in Figure 3) verifies this.

In short, these "Satellite only" splitters aren't really useful on TV and CATV frequencies or the amateur bands 70cm and below.

A "TV" splitter:
Figure 5:
The "through" loss of the Archer splitter.
Click on the image for a larger version.


I tested several splitters that were intended for general VHF/UHF/FM use - one of these being an "Archer" (Radio Shack) two-way splitter being typical of that type.  The implied frequency range is from at least 54 MHz to 700 MHz - the extent of the cable TV, FM broadcast, and off-air VHF and UHF TV frequencies at the time it was made.

Figure 5 shows the measured "through" loss in a 50 ohm system.  Compared to a plot done at 75 ohms (not shown, using resistive matching) the insertion loss barely changes across the frequency range.  In both 75 and 50 ohm systems, at least at 2 meters, down to 20 meters (14 MHz) seems to be "ok" - but the "dip" in the 3-4 MHz area - and the fact that the attenuation below it drops below 3dB - indicates that it's not likely acting like a splitter at these lower frequencies.

Figure 6:
Port to port isolation at 75 ohms for this splitter.
Click on the image for a larger version.
Figure 6 shows the port-to-port isolation at 75 ohms and we note that in the "low" and "high" VHF band (U.S. channels 2-13 - which more or less includes the 6, 2 and U.S. 222 MHz amateur bands, that the isolation is quite decent - well above 20 dB.

From this plot we can see that the "dip" in the 3-4 MHz area seen on Figure 5 is quite telling as the port-to-port isolation is pretty much gone below this frequency

Figure 7:
Port to port isolation in a 50 ohm system for this splitter.
Click on the image for a larger version.
The plot of Figure 7 shows what happens if the splitter is operated in a 50 ohm system.  The main effect is that the port-to-port isolation is reduced - being on the order of 15 dB or so from the 20 meter band through the 2 meter band (14 MHz - 144 MHz).

From this we can conclude that this splitter is quite usable from the middle of the HF spectrum through at least 2 meters - and is probably usable through 70cm.


A "TV/CATV/Satellite" splitter - preferred for HF use:

Figure 8:
Holland HFS-2P through loss in a 50 ohm system.
Click on the image for a larger version.
I have on hand several splitters that have on their label a frequency range that starts at (typically) 5 MHz with a high end of between 600 MHz and 2450 MHz.  The reason for this extended "low end" is likely due to their being designed for use in systems that have "Cable Internet" where the return (upstream) signal from the user's modem to the cable system are likely to be in the 5-50 MHz (or, possibly, a bit higher) range.  The plots included are those of a Holland Electronics HFS-2P which is a 2-way splitter/combiner that has a stated range of 5-2050 MHz and the results of this device are typical of that type.)

Figure 8 shows the "through" loss in a 50 ohm system showing a reasonable insertion loss (4 dB or below) from below 40 meters (about 5 MHz) through at least 70cm (440 MHz) - but again, the limitations of the measurement set-up make readings higher than this a bit suspect.

Figure 9:
Port-to-port isolation at 50 ohms.
Click on the image for a larger version.
Again knowing that the "isolation" measurement is the way to get the "true" story, port-to-port isolation in a 50 ohm system is depicted in Figure 9.

This verifies - to the extent that the test set-up can - the 5-2050 MHz range showing that the port-to-port isolation from 5 MHz to 1 Ghz is well over 15dB.  A port-to-port isolation measurement at 75 ohms (not shown) is slightly better (by a few dB) over the same range.

The combination of Figure 8 and Figure 9 show that this device may be usable down to the 160 meter band (1.8 MHz) provided that a slight amount of extra insertion loss (about 1dB) and lower isolation (approximately 12dB) can be tolerated.    (The Holland HFS-2D has characteristics similar to the HFS-2P down to 1.8 MHz.)

Figure 10:
The through loss, the other 7 ports being terminated with 75 ohm F-type
connectors.  The insertion loss is reasonable - between 10.5 and
11.5 dB over the range of 1.8 to 450 MHz.
Click on the image for a larger version.
An 8-way splitter:

The final splitter to be tested was the Holland Electronics GHS-8 8-way splitter-combiner.  Often, splitters with an even number of outputs greater than two contain multiple two-way splitters which means that this 8-way splitter might contain seven such devices - but I didn't break it open to check.
Figure 11:
The port-to-port isolation between two adjacent ports with the "in/out"
port and unused ports terminated with 75 ohm "F" loads.  The apparent
isolation is on the order of 35dB from 1.8 through 450 MHz - but this would
likely drop to something closer to 20dB.
Click on the image for a larger version.

Figure 10 shows the typical "through" insertion loss with the seven unused ports being terminated with 75 ohm "F" type terminators:  I don't have enough F-male to BNC-female adapters on-hand to terminate the 7 ports at 50 ohms - but if one were going to use one of these devices, it's probably more convenient to use F-type terminators on the unused ports, anyway.  The typical "through" loss is measured to be about 10.5-11.5 dB - slightly higher than the predicted "ideal" 9dB insertion loss, but typical for these devices.

The port-to-port isolation was also measured and the use of 75 ohm terminations on the other ports and the "common" in/out port likely improved this:  The isolation would likely be significantly worse if all ports were at 50 ohms, for the same reason as the other splitters tested.

Based on these readings, this device is useful down to 1.8 MHz and up through 2 meters - and probably 70cm.

Figure 12:
Coupling coefficient at 50 ohms for this tap
Click on the image for a larger version.
A TV-type signal "tap":

Likely unfamiliar to many, a signal "tap" is a very useful device in multi-drop TV installations found in hotels, hospitals and other larger buildings.  Unlike a splitter - which usually divides a signal equally to its output ports - a "tap" will siphon only a certain amount of signal off the cable and leave the majority of it intact - which is very useful for systems such as those in a hotel or hospital to distribute and split a signal hundreds of times to serve all of the devices.

In some ways it can be considered to be similar to a part of an SWR bridge where only a small amount of signal is sampled - and in only one direction - allowing the majority of the original signal to pass with minimal loss.  Several taps - all from Holland Electronics - were tested as they were what was on-hand and the "DCG-6SB" is represented in the plots. 
Figure 13:
The "reverse isolation" loss of the tap (e.g. turned "backwards") with a
50 ohm termination.
The reverse isolation is described as being the absolute amount of isolation
(e.g. that in the chart above) minus the coupling coefficient which means
that the actual forward coupling loss - which means that using Figure
12, we know that the actual reverse isolation is about 7 dB lower than
indicated by the graph above.
Click on the image for a larger version.

Figure 12 shows the "coupled" energy in a 50 ohm system:  Compared to the coupling in 75 ohm system (now shown) the insertion loss was slightly higher (about 1dB) but the frequency loss/flatness was about the same, being pretty consistent from about 1.8 MHz through 1 GHz.

Figure 13 shows the reverse isolation of the tap:  Rather than 6dB of coupling from the main line for signals going the "other way", the absolute is closer to 20dB - about 13dB lower.  (The actual reverse isolation is the absolute isolation minus the forward loss).  In a 75 ohm system (not shown) the reverse isolation was quite a bit better (closer to 30dB over the 5 MHz-1GHz range) - but this result is completely expected:  The reverse isolation is akin to measuring VSWR, and operating a 75 ohm device at 50 ohms implies a VSWR of 1.5:1 - a "return loss" of 14dB - very close to the values depicted in Figure 13 over much of the frequency range when the "forward" loss is taken into account.

On a tap there is yet another measurement to be taken - the loss between the in and out port.  Because we are measuring a 6dB tap - a device which siphons off about 25% of the signal - we would expect at least that amount (theoretically 1.25dB for 6dB) to be lost as it is coupled to the "tap" port. Figure 14 we can see that the measured loss is slightly higher than this between 1.8 and 200 MHz- a bit over 2dB.  Some of this "extra" loss is due to the intrinsic losses of the device, but a smaller amount is a result of the use of a 75 ohm device on a 50 ohm system.
Figure 14:
Through loss of the 6dB tap in a 50 ohm system.
Click on the image for a larger version.

This device - which is rated down to 5 MHz - may be useful through at 160 meters (1.8 MHz) - but the insertion loss goes up rather quickly at lower frequencies.

This device is NOT suitable for passing DC (e.g. for amplifiers, control signals) as it has a DC short across it - but that is not true of all taps.  For example, the Holland Electronics "HDCS" series does allow low frequency RF down to DC to flow through it - but like the DCG-6SB, its coupling coefficient deteriorates quickly below about 1.8 MHz.

* * *
General conclusions:

If you are going to use TV-type splitters for HF, make sure that you get devices that are explicitly rated down to 5 MHz.  Based on the (limited!) sample of devices that were tested, these devices can be expected to work into the 160 meter amateur band (down to 1.8 MHz).  While these devices may be usable thoughout the entire AM broadcast band (down to 540 kHz) expect performance to drop quickly in terms of added "through" attenuation and worse port-to-port isolation.

A "TV" type device - one that may indicate a start frequency of 5 MHz, or just any device that is claimed to work at TV (VHF/UHF) and FM broadcast frequencies will likely work from 6 meters through 70cm (50 MHz - 450 MHz).

Again, for general signal splitting and combining, these 75 ohm devices, used at 50 ohms, are quite usable for non-critical applications - provided that they be used at low power levels (a few 10s of milliwatts at most) and where one need not have precise 50 ohm matching and high port-to-port isolation.  Remember that most 50 ohm devices (receivers, amplifiers, filters) have only "approximately" 50 ohm source/load impedances - and filters in particular will, out of their design frequency range (outside the band-pass, on a notch frequency, above the low-poss cut-off, below the high-pass cut-off) will likely have anything but a 50 ohm characteristic impedance, so even a "proper" 50 ohm splitter/tap device would not necessarily yield any better performance in those situations.

For information about the design and use of splitters/combiners in general, a good reference is Mini-Circuits AN10-006, "Understanding Power Splitters" - link.

* * *

Far more data was gathered than was presented here, but that depicted above is representative of the devices that were on hand.

* * *

This page stolen from ka7oei.blogspot.com

[End]



Saturday, December 7, 2019

Automatically re-connecting low-voltage cut-outs for 12 and 24 volt lead acid battery systems

In a previous post I described a simple circuit that provided a low-voltage cut-out that could be used in a battery-operated system - see the article "A latching low-voltage disconnect for 12 volt lead-acid and lithium batteries".

That circuit - intended mostly for lithium-based batteries - required manual intervention to "reset" the device, intended for those situations where you wanted to provide manual intervention in resetting the circuit to prevent causing harm to the battery - and maybe the gear connected to it.

Figure 1:
The completed 24 volt low-voltage cut-out.  R5, the cut-off voltage
adjustment, is the blue 10-turn potentiometer near the middle and
R8 is the black, single-turn potentiometer near the top, to the left
of Q1 and its heat sink.  U1 is the lower transistor-looking device
near the left side of the board.
This device was installed in an outdoor enclosure that houses a relay
point to bridge a wireless Internet connection over a hill that has
a 24 volt back-up battery system.  In the event of an extended
power outage, this device prevents damage to the batteries,
particularly in the winter where this device will
prevent the batteries from being damaged by freezing.
The small heat sink is adequate for this FET and an operating
current of 5 amps.
Click on the image for a larger version.
This circuit is different from that described in the link above, intended for lead-acid battery based systems where an automatic reconnect of the battery is required when the battery voltage rises above a threshold voltage after application of charge current.  An example of where this would be useful is a system in which the battery is charged by a main-powered DC supply that is used to keep the battery charged where the battery will power the load in the event of a power failure.

Why would we worry about this?

Related to this, let's briefly talk about maximizing longevity of lead-acid batteries:
  • For best longevity, the shallower the discharge, the better.  A classic example of this is a solar-powered system where there is a daily discharge.  If the battery were sized too small such that it was discharged from "full" to 30% every day, it might last only a year or so at best, but by increasing the battery size such that it was only discharged to 75% worst-case, the cycle depth would be much shallower and the battery would last much longer.  This assumes, of course, that there is enough charging capacity to do this, which can be an issue for a solar charge system in the winter and/or periods of weather where there is little direct sun.
  • Lead-acid batteries can handle being run completely down - as long as you don't keep them in that state for very long (a few days at most - as little time as possible) and don't do it very often - and the more charge you can leave on them at the deepest point of discharge, the better.  In other words, if you run an otherwise healthy lead acid battery completely dead and immediately recharge it, little immediate damage is likely to have been done other than taking a bit of life off it farther down the road as most lead-acid batteries can only handle a few hundred of such cycles before they are significantly and permanently degraded.  (The previous statement applies only to "Deep Cycle" lead-acid batteries:  Automotive "starting" batteries do not fall into this category - They can be permanently damaged by as few as dozen or two "complete" discharge cycles in some cases.)
  • In cold-weather environments, while the degradation (primarily sulfation) caused by the battery being dead for extended periods is dramatically slowed, extremely deep discharge also reduces the specific gravity, raising the electrolyte's freezing point and increasing the possibility of the battery being damaged/destroyed at very low temperatures if it does get cold enough to freeze the electrolyte.  Preventing too-deep a discharge by disconnecting the load while there is still some capacity left can reduce the probability of this happening.  (This can happen with both flooded cell and AGM batteries.)
  • Important:  A general rule of thumb is that above approximately 70F (approx 20C) for every 10F (approx. 5.5c) increase in temperature, the expected lifetime of a lead-acid battery is approximately halved.  Carefully consider this if the battery is to be in a location that might get warm/hot.
For several of the reasons mentioned above, a low-voltage disconnect circuit is essential in systems where there is the distinct possibility that the battery could be completely discharged - both to prevent the loss of battery capacity due to over-discharge (e.g. sulfation if the battery is left "flat" for a long time) as well as preventing freezing of the electrolyte when the battery is deeply discharged where/when that is a concern.

A simple low-voltage cut-off circuit:

While there are many ways to build such a circuit, the circuit below - designed for a 24 volt battery system (values are included for use with a 12 volt battery) - demonstrates a simple, yet flexible device.

Figure 2:
A low-voltage battery disconnect, designed for 24 volts.  It has a useful adjustment range of between 22 and 28 volts.
For 12 volts:  R2=4.7k, R4=62k, R7=47k, R11=47k, R12, R13, R14=10k.
 Components LED2 and R14 are optional, used to indicate that the circuit has disconnected the battery from the load.  (Errata:  LED2 is shown connected backwards.)

Not shown is appropriate fusing on the battery side of the circuit to protect against short circuits and/or catastrophic component failures.
Click on the image for a larger version.
This circuit, somewhat similar to that described in the other article linked above, works as follows:

The battery voltage is measured using R4, R5 and R6 which form an adjustable divider to take the battery voltage down to the 2.5 volt comparison voltage required by U1, a TL431 "programmable" zener diode.  When the voltage at the wiper of R5 goes above 2.5 volts, U1 will conduct.

When U1 conducts it pulls down the voltage at the base of Q1 via R3 causing it to turn on.  This voltage, via R12, is fed to the gate of Q1 - an N-channel power MOSFET in the negative battery lead. In this way, as long as the voltage is above the threshold set by R5, Q1 is turned on, connecting the load to the battery.  While U1 is turned on and pulling the voltage at the bottom end of R3 close to ground, the voltage at the gate of Q3 is low, keeping it turned off.

If the divided battery voltage drops below the low-voltage cut-out threshold, U1 turns off and its cathode is pulled high by R2 with resistor R1 being used to allow Q2's gate to go high and turn it off.  When Q2 is turned off, the gate voltage on Q1 goes away, being pulled to its source potential by R11.

At this same time, the rise in voltage at U1's cathode also causes Q3 to be turned on and the voltage at the junction between R4 and R5 is pulled slightly down, requiring a higher battery voltage to turn U1 back on than was required for it to be turned off, thus providing a bit of hysteresis.  It is by adjusting R5 that the amount of hysteresis (e.g. the amount by which the turn-on voltage must be higher than the cut-off voltage) may be set.

There are some "extra" components included to make the circuit more robust:
  • C1, across the reference pin of U1, reduces this circuit's susceptibility to being triggered by power glitches (e.g. lightning) and stray RF energy.  If the circuit tends to trip out when a load is activated (e.g. voltage "dip" caused by battery/lead resistance)  this capacitor can be increased in value to slow U1's response - up to several microfarads.
  • Capacitors C2 and C3 are across their respective FET's gates to provide RF energy protection as well as to smooth transients that might cause false triggering or damage.
  • ZD1 is placed between Q1's source and gate to protect it against excess voltage that would damage the device - possibly caused by transients.  The maximum gate-source voltage rating of a typical power FET (Q1) is around 20 volts, so this is a required component for the 24 volt version.
  • Finally, C4 is placed across the drain and source leads of Q1 to provide both resilience to stray RF energy as well as to suppress transients that might appear across it when the power is off.
 
Also included in the circuit is the "power on" indicator consisting of LED1 and R13 which provides an indication that voltage is present between the "Load +" and "Load -" terminals.  The optional circuit containing LED2/R14 shows when Q1 is turned off (e.g the load is disconnected) by detecting when there is a voltage drop across it.  (Note:  LED2 was accidentally drawn "backwards" in Figure 2.)

Comments:
  • Important:  To simplify the design, it is the "Batt -" lead that is switched to disconnect the load.  What this means is that the negative battery lead must not be connected anywhere else or the battery disconnect will not work properly.  It is possible to design the circuit to use a P-channel FET in the positive lead, but high-current N-channel devices are more plentiful and less expensive.  Instead of a power FET, a slight modification will allow a relay to be used, instead - but at much higher quiescent current owing to the required current for the coil.
  • LED1 and LED2 may indicate correctly only when there is a load connected across the "Load +" and "Load -" terminals.
  • Even when the load is disconnected this circuit will draw several milliamps of current due to the quiescent load of U1, its voltage divider and LED2, the "Battery Disconnected" indicator.  Keep this in mind if you are going to use it in an application where there is a reasonable probability that battery charging may be unavailable for very long periods of time.
  • The current capability of this circuit is limited by the current-handling capability of Q1, the power FET, and the amount of heat-sinking attached to it.  When choosing a power FET, not only should the current rating be considered, but also the specified "on" resistance.  For example, if a 50 amp FET is used and it has an ON resistance of 0.01 ohms, at 20 amps it will dissipate at least (20 * 20 * 0.01 = )  4 watts and will drop (20 * 0.01 = ) 0.2 volts.  The heat sink shown in Figure 1 is capable of dissipating that much power under all but the hottest ambient temperature.
Comment about Q1, the main power FET:

The power FET is not specified, as practically any appropriate N-Channel power FET will work - and here are a few selection guidelines:
  • Pick a device with a current rating of at least 3 times the maximum that you expect to draw.  100-150 amp devices are fairly inexpensive, so don't be afraid to pick a far "heftier" device than you might otherwise select.  Note that while a particular FET may have a "150 amp" rating, if there is a short circuit downstream, it is possible that connections inside the FET itself may act as a quicker-acting fuse than an actual fuse intended to protect the circuit.
  • Pick a device with a voltage rating of at least 4 times the battery voltage, which would imply at least a 125 volt device for 24 (really around 28) volts and 75 volt device for 12 (really around 14) volts - and there is little penalty in going higher than that given a choice of devices with the same rated current.
  • With a properly functioning circuit, a heat sink is likely not required, but a small heat sink doesn't hurt - particularly if the circuit will be exposed to high temperatures.  While the "on" resistance of a saturated power MOSFET can be very low, it is not zero and at maximum expected current, the FET can produce a some heat.
    • Note that a FET's "on" resistance may be a fraction of an ohm, but it is not zero so some voltage will be dropped - and heat generated - by the FET.
    • Remember the formula - Power in watts = I*I*R where "I" is the current and "R" is the "on" resistance of the FET - and if power is more than 2-3 watts, consider a larger heat sink for Q1 - and after it is built, make sure Q1 isn't getting "hot" under the expected load.
The amount of current that can be handled by this cut-off-switch is essentially the limit of the chosen FET, the fusing, and the interconnection wiring.
Examples of FETs:
  •  The IRF540 is rated for 25+ amps at 100 volts and this would be an excellent device for a disconnect that was intended to handle 5-8 amps at 12 volts.  This device is inexpensive and readily available - even from surplus outlets, less than $1.50 from Electronic Goldmine.
  • The 45N20 is rated for 35 amps at 200 volts and would be a good choice for 10-12 amps and either a 12 or 24 volt disconnect - and is also available from Electronic Goldmine as the "SSP45N20".
(The Electronic Goldmine is used as an example of surplus electronic suppliers.  These parts are readily available from suppliers such as Digi-Key, Mouser, etc.  Always buy parts like this from reputable suppliers as counterfeit power devices are quite common.)

Adjustment:

Important:  Verify that the circuit works and is properly adjusted before placing it into service!

The procedure is as follows:
  • Set R8 for maximum resistance. 
  • Connect "Batt +" and "Batt -" to an adjustable DC power supply that is capable of providing between 20 and 30 volts.
  • Connect a known-accurate voltmeter to the DC power supply to measure the "Battery" voltage.
  • Connect a load to the output terminals ("Load +" and "Load -").  A small lamp of the correct voltage or resistor (1k at 1 watt or more) will do. 
  • Increase the voltage to 30 volts:  The load will turn on. (15 volts for the 12 volt version.) 
  • Decrease the voltage slowly:  The load will turn off below a certain voltage.
  • Increase voltage again to turn the load back on and then adjust R5 so that the load turns off at the desired voltage  The lowest cut-off voltage that is suggested is 1.92 volts/cell for a state-of-charge of 20-25% at "room temperature", which translates to:
    • For a 12 volt lead acid battery, 11.5 volts
    • For a 24 volt lead acid battery, 23.0 volts
    • Note:  At very low temperatures, the output voltage of a lead acid battery will drop given a constant charge level.  This means that in the case of the 12 volt battery that is cold (32F, 0C) a 11.5 volt cut-off would imply that the actual charge level of a battery would be higher than the 20-25% level indicated by this same voltage of the battery were at room temperature.
  • This concludes the adjustment of the cut-off voltage.  Adjusting R8 should not affect the cut-off voltage.
  • Set the power supply voltage below the cut-off voltage and slowly increase it until the load turns back on, noting the voltage at which that happens.
  • Adjust R8 so that the cut-in voltage is adjusted as desired.  The cut-in voltage is much less critical, but should be well above the cut-out voltage.
    • A typical cut-in voltage for a 12 volt lead-acid battery is 13 volts, or 2.17 volts/cell, but it may be set lower - to 12.5 volts (2.083 volts/cell) if quicker power restoration is required in a system that has limited charge capacity. 
    • A typical cut-in voltage for a 24 volt lead-acid battery is 26 volts, or 2.17 volts/cell, but it may be set lower - to 25 volts (2.083 volts/cell) if quicker power restoration is required in a system that has limited charge capacity. 
    • Note:  A higher cut-in voltage may delay restoration of the output load, but it will allow the battery to charge more before the load is restored.  An example where this may be important is in a solar-powered system where a low output from the PV array may be sufficient to slowly charge the battery, but may not be able to sustain the load:  This can happen in the morning, when the panels are covered with snow, and/or on very dark and cloudy days.
  •  Re-check the cut-off and cut-in voltages:  When adjustment is needed, remember to set the cut-off voltage first and then the cut-in voltage.

Comment:
This device could be used for similar-voltage Lithium-based batteries with appropriate adjustment of the cut-in and cut-off voltages.  Note that any Lithium-based battery system should already have a "protection" circuit that protects against over and under voltage.  Particularly with LiFeO4 batteries, a circuit to equalize the voltage across the cells should be used.
* * *

This page stolen from ka7oei.blogspot.com

[End]


Sunday, November 17, 2019

Homebrew construction of 2 and 4 port splitters/combiners for the LF-MF-HF(30 kHz-30 MHz) frequency range.

Note:

This is a follow-up of a previous article, "Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range" - link.

Comment:
All of the devices described here could also be used to combine signals from multiple sources.  Unless the signals being combined are "phase coherent" (e.g. from the same signal source) the insertion loss will be the same as that in splitter operation.

"Rolling your own" splitter for LF through HF (<30kHz-30MHz):

Unless you get the Mini-Circuits devices for cheap at a swap meet or via a surplus outlet, their cost may be a bit prohibitive for casual use in the shack.  How about making your own splitter that will work over the 30 kHz-30 MHz range?

Why would one want this?  There are a number of modern Web-Based SDR receivers that cover from (literally!) audio through 30 MHz - and in my case, I have a number of KiwiSDR receivers - link that are connected to an antenna system that is capable of intercepting signals over this range.  If one has several such receivers, it can be a challenge to find a splitter that works well over this range - particularly the low end - as described in the article linked above.

Figure 1:
A two-transformer splitter/combiner.  L1 transforms the impedance at
J1 to half that value and L2 splits the signal itself.  R is twice the system impedance
- 100 ohms in a 50 ohm system.  The tap on L1 is at 0.707 times of the total
number of turns:  A tap at 7 of 10 total turns is "close enough" while C is a high-
frequency compensating capacitor.  L2 is a bifilar-wound transformer with the
two sets of windings connected in series at the common point.
Click on the image for a larger version.
To do this, ferrite - rather than iron-powder - cores would be used, the most common types using mix 31, 43, 61, 73 and 75 - and the most useful of the lot for the low-frequency end are types 73 and 75.  Not having a complete assortment of the ferrite types on hand, I used what I had and the use of a binocular core with mix 43 and a ferrite with mix 75 is discussed here.

Two-transformer splitter/combiner:

A common splitter topology consists of two cores:  One to transform the impedance to half that of the characteristic system impedance and a second to split the signal two ways as depicted in Figure 1.  The inductance of L1 and L2 should be high enough to present a reactance of 3-10 times the system impedance at the lowest frequency.

Figure 2:
Measured insertion loss of a transformer using two BN43-202 cores wound
with 30 AWG wire:  10 turns, tapped at 7 turns for L1 and 10 bifilar turns
for L2.  R = 100 ohms and C = 62pF.
The required number of turns to achieve the desired low-frequency response
increases the stray capacitance, undesirably increasing loss at the
high-frequency end of the HF spectrum.
Click on the image for a larger version.
Figure 2 shows a tested version of this transformer, the details noted in the caption.  According to this diagram the insertion loss is a nominal 3-ish dB from 50 kHz to 1 MHz, dropping down to about 5 dB loss between 20 kHz and 10 MHz and nearly 6dB at 30 MHz.  Capacitor "C" was made variable, adjusted for lowest loss, improving the highest end by a bit over 1dB, and its value measured.

For LF and HF use, this splitter is just "OK" - the loss being an extra 3dB at the high end of the spectrum:  If preceded with amplification, this loss may be tolerable - but note that even the nominal 3dB loss of a 2-way splitter should be of concern at the higher HF bands as signals - and the natural noise floor - can be quite weak and additional loss can drop the receiver's noise floor below that, potentially causing the loss of reception of weaker signals.

Much of the high-frequency loss is due to the inter-winding capacitance.  Experimentally, versions were constructed using wire with PTFE ("Teflon") insulation and comparing it with another with the same number of turns of the 30 AWG enamel and the losses for the PTFE wire version were 1.5-2dB lower - but fewer turns could be passed through the core and low-frequency response suffered.

Figure 3:
In this form, the primary (connected to J1) has 1.414 times as many turns
as each of the two identical secondary windings.  The value of R is half that
of the characteristic system impedance, or 25 ohms for a 50 ohm system:
Parallel 51 ohm resistors were used for a nominal 25.5 ohms.
Typical turns values are 10/7 turns, 14/10 turns and 20/14 turns for the
primary and bifilar secondary, respectively.  For the center-tap, the windings
of the secondary are connected as if they were in series.
Click on the image for a larger version.
If a higher-permeability material (like 73 or 75 mix) were used for the core rather than 43, fewer turns could have been used to maintain the inductance and low-frequency response and it is likely that the high-frequency loss would be reduced.

Single-transformer splitter-combiner:

Another common splitter/combiner is the form depicted in Figure 3, using a single core - and potentially this can reduce loss compared with a device with two cores.

In this system the primary should consists of 1.414 times (e.g. the square root of two) as many turns as each of the secondary windings.

Figure 4:
The insertion loss of the described two-way splitter using 24 AWG wire on
an FT50-75 core:  It is well below 4dB over the range of 10 kHz to 60 MHz.
Click on the image for a larger version.

Both a binocular and toroidal core were tried and better results were obtained with the FT50-75 core than the available BN43-202 binocular core - both because the higher permeability improved the low-end response and larger wire could be used for the toroid:  Capacitance was reduced on the toroid because the turns could be spread out rather than being tightly overlaid as the case of the binocular core, and high-end losses were further-reduced by laying the turns of the secondary next to each other rather than the higher capacitance resulting from the two conductors being twisted as is commonly done with Bifilar windings.

The results of this work are visible in Figure 4.  For this transformer, two parallel secondary "bifilar" windings consisting of 14 turns each were carefully and neatly laid down using 24 AWG enamel wire with 20 turns of 24 AWG over the top.  As can be seen, the results are excellent:  The insertion loss is below 3.6dB from 10 kHz to 60 MHz and the overlaid Smith chart shows the VSWR to be pretty well-behaved, never exceeding 1.5:1 over this range.

Figure 5:
The port-to-port isolation is quite good over the range of 100 kHz to
30 MHz.  The peculiarly-flat isolation limit of the bottom "trough"
of the graph is a result of the value of "R" not being exactly 1/2 of the
impedance value of the system used for testing:  If R is made variable,
higher isolation may be obtained in the middle of the range - but the
difference between that and the fixed resistors used was only an
ohm or two.  In practice, high values of isolation can be obtained only
if the source and load impedances are purely resistive, but since
practical antennas, amplifiers, receivers and filters will not be perfect
sources and loads, such high isolation cannot be achieved in practice.
Click on the image for a larger version.
Additional tests were run to determine the port-to-port isolation of this splitter - the results being visible in Figure 5.  Over the range of 100 kHz through 60 MHz, the isolation exceeds 15dB, exceeding 25dB from abut 100 kHz through 30 MHz.  During testing, the same device had been constructed using smaller 30 AWG wire and the results were worse above 10 MHz (by 2dB at 30 MHz) - likely a result of the skin effect losses of this smaller wire.

A four-way splitter:

I happened to have a need to take signals over a wide frequency range and split it four ways - specifically, to several KiwiSDR receivers, stand-alone web-based receivers capable of reception over the 5kHz-30MHz range - so I decided to construct a splitter using the configuration described above.  To do this, I would need three splitters:  A pair of splitters to feed the four outputs and one more splitter to feed the aforementioned two splitters.  This splitter is depicted schematically in Figure 6:
Figure 6:
For a 50 ohm system, resistors "R" are 25.5 ohms (two 51 ohm resistors
in parallel) and capacitors "C" are 47pF NPO/C0G types used to "flatten"
the response to 30 MHz.
The as-built splitter uses FT50-75 cores wound with 24 AWG, the dual
secondary windings consisting of 14 turns and the primary with 20 turns.
The dual secondaries are laid parallel rather than twisted to minimize stray
capacitance.  The center-tap is connected as if the two secondary windings
were placed in in series.
Click on the image for a larger version

This splitter consists of three of the two-way splitters connected as described:  FT50-75 cores wound with 14 turns, each of two parallel 24 AWG conductors for the secondary overlaid with 20 turns of 24 AWG for the primary.  During testing it was observed that the addition of capacitors "C" slightly reduced (by nearly 1 dB) the insertion loss at 30 MHz at the expense of increased loss (about 2dB) at 60 MHz - but because the target high-end limit was 30 MHz, this was considered to be acceptable.

The end result was an insertion loss (see Figure 7) of less than 7 dB from 20 kHz through 30 MHz, rising to 8 dB and 9.3 dB at 10 kHz and 60 MHz, respectively, being under 6.3dB between 50 kHz and 10 MHz.  In testing port-to-port isolation, the worst case results were those obtained from the same transformer (e.g. T2 or T3) and this value was at least 15dB from 50 kHz to 30 MHz.

Figure 7:
This shows the typical insertion loss of the as-built four-way splitter
depicted schematically in Figure 17.  The insertion loss is less than 7 dB
from at least 20 kHz through 30 MHz with the VSWR being 1.5:1
or less over that same range.
Click on the image for a larger version.
This four-way splitter was built into a small die-case box for mechanical rigidity and electrical shielding.  Inside the box, pieces of plastic tape were affixed to the bottom and the lid to eliminate the possibility of inadvertent shorting of connections to the case:  Details of the mechanical construction may be see in Figure 8.

To reiterate:  It was determined that with the number of turns required to obtain good response into the LF range (e.g. below 30 kHz) that the use of twisted bifilar windings was NOT indicated:  Doing so resulted in excess loss (3-6dB) by the time one got to 30 MHz.  As indicated, the use of thicker insulation (e.g. PTFE versus enamel) reduced this loss somewhat, but using the smallest wire on hand with the only available 75 mix toroid, too few turns could be wound to afford the needed inductance for the desired low frequency respons:  The best-results with the materials on-hand were obtained by simply laying the "bifilar" windings parallel to each other.  In this case, 24 AWG enamel wire was used, a compromise between lower skin-effect losses and the ability to fit the required number of turns on the FT-50 core.

Comment:  There are other splitter topologies available that have their own sets of advantages and disadvanges.  While some of these may be discussed in (a) future article(s), they are beyond the scope of this article - which is the construction of a very simple, straightforward device that is suitable for the task at hand.

Conclusion:

If one needs a very wide-range splitter for broadband receivers that cover from LF through HF - such as some modern "Direct Sampling" SDRs (e.g. the KiwiSDR) there are some commercially-available devices that may be found that will work well - if you can find them surplus, or are willing to pay for them.  If you are willing, a perfectly suitable device may be constructed inexpensive using a minimal complement of components.

Figure 8: 
The as-built 4-way splitter. capable of useful operation from below 20 kHz to above 30 MHz.
As described in the text, the cores are FT50-75 wound with 24 AWG wire, wired "dead bug"
inside a small die-cast aluminum box.  The resistors "R" and compensating capacitors "C"
may be easily seen.  The bottom of the box and the lid (not visible) are insulated with a piece
of plastic tape - this this case, 1" (25mm) wide PET (Polyester) tape.  If I'd had some on hand,
I would have wound the transformers on slightly larger toroids to spread out the windings a bit.
Click on the image for a larger version.
* * *

This is a follow-up of a previous article, "Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range" - link.



Stolen from ka7oei.blogspot.com


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