Wednesday, April 19, 2017

A daylight-tolerant TIA (TransImpedance Amplifier) optical receiver

While the majority of my past experiments with through-the-air free-space optical (FSO) communications were done at night, for obvious reasons, I had also dabbled in optical communications done during broad daylight, with and without clouds.

The challenge of daylight:

Clearly, the use of the cloak of darkness has tremendous advantages in terms of signal-noise ratio and practically-attainable communications distances, but daylight free-space optical communications has some interesting aspects of its own:
  • It's easier to see what you are doing, since it's daylight!
  • Landmarks are often easier to spot, aiding the aiming.
  • Even in broad daylight, it is possible to provide signaling as an aiming aid, such as a mirror reflecting the sun - assuming that it is sunny.
  • The sun is a tremendous source of thermal noise, causing dilution of the desired signals.
  • Great care must be taken when one wields optics during the day:  Pointing at the sun or a very strong specular reflection - even briefly - can destroy electronics or even set fire to various parts of the lens assembly!
As you might expect the biggest limitation to range is the fact that the sun, with its irradiance of around 1kW/m3 (when sunny) can overwhelm practically any other source:  This is why the earliest "wireless" communications methods often used reflected sunlight, notably the Heliograph, where a mirror was "modulated" with telegraph code or the "Photophone", a wireless audio transmitter using reflected light, an invention of Alexander Graham Bell from the 1870s with earlier roots - a device that Bell himself considered to be his greatest invention.
Figure 1:
Optical communications during daylight.  In the center of this contrast-enhanced picture (the red spot) is the light from an optical transmitter using a 30+ watt LED at a distance of 13.25 miles (21.3km).  This image is from my own "modulatedlight" web site.
Click on the image for a larger version.

Means of detection:

While the modulated speech may be produced in any number of ways (vibrating mirror, high-power LED, LASER) some thought must be given on the subject of how to detect it.  While the detector itself need not be spectacularly sensitive due to the nearly overwhelming presence of the thermal noise from the sun, it is worth making it "reasonably" sensitive so that this is not the limiting factor.  An example of an un-sensitive optical receiver (e.g. one that is rather "deaf" and itself is not likely to be sensitive enough even for daylight use) is a simple circuit using a photodiode as depicted below:
Figure 2:
A simple phototransistor-based receiver (top).  This circuit was built by Ron, K7RJ, simply to demonstrate the ability to convey audio a short distance:  It is (intentionally) not optimized in any way and is not at all sensitive.  A similar, but slightly better circuit was found on the Ramsey Electronics LBC6K "Laser Communicator", which was also quite "deaf".  See the article Using Laser Pointers for Free-Space Optical Communications - link that more thoroughly explains this issue.  This image is from my own "modulatedlight" web site and used with the permission of Ron Jones, K7RJ.
Click on the image for a larger version.

The circuit in the top half of Figure 2 (above) depicts one of the simplest-possible optical receivers - and one of the "deafer" options out there.  In this case a biased phototransistor is simply fed into an LM386 audio amplifier and the signal is amplified some 200-fold (about 46dB.)  As noted in the caption, this was a "quick and dirty" circuit to prove a concept and was, by no means optimized nor does it take maximum advantage of the potential performance of a phototransistor.

As it turns out a phototransistor isn't really the ideal device because it is, by itself, intrinsically noisy.  Another, more practical issue is that its active area is typically quite small which means that it won't intercept much light on its own.  Of course, any half-hearted attempt to use any device for the detection of optical signals over even a rather short distance of a few hundred meters would include the use of a lens in front of the detector - no matter its type:  The lens will easily increase the "capture area" by many hundred-fold (even for a small lens!) and will effect noiseless amplification with the added benefit of rejecting light sources that are off-axis.  With the tiny active area of a phototransistor it can be difficult to properly and precisely focus the distant light onto that area and it is likely that unless very good precision in both alignment and focus can be maintained, the "spot of light" being focused onto the phototransistor will be larger than its active area, "wasting" some of its light as it "spill"s over the sides.

One of the biggest problems with a circuit like this is that there will be a level of light at which the phototransistor saturates, and when this happens the voltage across its collector and emitter will go very close to zero and the received signals will disappear, possibly "un-saturating" only briefly during those points in the modulation where the source light happens to go toward zero, resulting in badly distorted sound.  In broad daylight the phototransistor may be hopelessly saturated at all times unless an optical attenuator (e.g. neutral density filter) is used to reduce the total light level and/or more current is forced through it.

Introducing the TransImpedance Amplifier (TIA):

A much better circuit is the TransImpedance Amplifier, a simple circuit that proportionally converts current to voltage.  With this circuit (see Figure 3) one would more likely use a PIN Photodiode, a device akin to a solar cell, in which the output current is pretty much proportional to the light hitting its active area. This is quite unlike the manner in which a phototransistor is typically used where in the former case the impinging light causes a voltage drop across the device.

Figure 3:
A simple transimpedance amplifier.
(Image from Wikipedia)
In this circuit the junction between the inverting (-) and noninverting (+) inputs of the op-amp "wants" to be zero, so as the current from the photodiode increases in the presence of light, its output voltage will increase, sending a portion of that current through feedback resistor "Rf" until the overall voltage is zero.  What this means is that the output voltage, Vout, is equal to the current in the photodiode multiplied by the magnitude of resistance, Rf - except that the voltage will be negative, since this is an inverting amplifier.

As an example, assume that Rf is set to 1 Megohm.  Assuming no leakage and a "perfect" op-amp we can determine that if there is -1 volt output, we must have 1/1000000th of an amp (e.g. 1 microamp) attributed to Ip, the photodiode current.  This sort of circuit is often used as a radiometric detector - that is one in which its output is directly proportional to the amount of light striking the photodiode' surface, weighted by intervening optics and filters and the spectral response of the detector itself.

For more about the Transimpedance Amplifier circuit, visit the Wikipedia page on the subject - link.

This is OK when the photodiode is in complete darkness - or in near-complete darkness, but what about strong light?  We can see from the above example that if we have just 10 microamps - a perfectly reasonable value for a typical photodiode such as the BPW34 in dim-to-normal room lighting - that Vout would be -10 volts.  If this same circuit were taken outside, the diode current could well be many hundreds or thousands of times that amount and this would "slam" the output of the op amp against a rail.

One of the typical means of counteracting this effect is to capacitively couple the photodiode to the op amp so that only changing currents from a modulated signal get coupled to it, blocking the DC, but there is another circuit that is arguably more effective, depicted in Figure 4, below.

Figure 4:
A "Daylight Tolerant" Transimpedance amplifier circuit.
In this circuit the DC from the output is fed back to "servo" the photodiode's "cold" side so that its "hot" side (that connected to the op amp's inverting input) is always maintained at the same potential as the noninverting input, eliminating the DC offset caused by ambient light.  The disadvantage of this method is that it does not lend itself well to reverse-biasing the photodiode to reduce its capacitance, but between the high intrinsic thermal noise levels of daylight and the related photoconductive shunting of the device due to high ambient light, this is largely unimportant.  For the photodiode the common and inexpensive BPW34 may be used along with many other similar devices.
Click on the image for a larger version.

This circuit is, at its base, the same as that depicted in Figure 3, with a few key differences:
  • An "artificial ground" is established using R101 and R102, allowing the use of a single-polarity power supply.  This artificial ground is coupled to the actual ground via C102 and C103 making it low impedance to all but DC and very low AC frequencies.
  • The voltage output from the transimpedance amplifier section (U101a) is feed back via R104 to the "ground" side of the photodiode (D101) to change its "ground".  If there is a high level of ambient light, the voltage at the "bottom" end of D101 (at D101/C107) goes negative with respect to the artificial ground, setting the DC voltage at the non-inverting input of the op amp to zero, cancelling it out.
  • R104 and C106 form a low-pass filter that passes the DC offset voltage to the bottom of D101, but block the audio.  In this way the DC resulting from ambient light that would "slam" the op amp's output to the negative rail is cancelled out, but the AC (audio) signals remain.  The time constant of this R/C network is slow enough to be "invisible" all but the very lowest (subaudible) frequencies, but more than fast enough to track changes in ambient light.
  • By not placing any additional components between the "hot" end of the photodiode and the op amp, the introduction of additional noise from the components (including microphonic responses of the coupling capacitor) is greatly reduced.
In the above circuit the values of R103 and C104 would be chosen for the specific application.  In a circuit that is to be used at very high light levels where high sensitivity is not very important a typical value of R103 would be 100k to 1 Megohm:  Do not use a carbon composition but a carbon film or (better yet) metal film resistor is preferred for reasons of noise contribution.  While tempting to use, a variable resistor at R103 is also not recommended as these can be a significant source of noise.  If multiple gain ranges are used, small DIP switches, push-on jumpers or even high-quality relays - wired to the circuit - could be used to select different feedback resistances, knowing that these devices have the potential of introducing noise as well as additional stray circuit capacitance.  (Such a relay/switch would be wired on the "output" side of the op amp/relay of the feedback resistance(s).)

C104 is used to compensate for photodiode and other circuit capacitance and without it the high frequency components would rise up (e.g. "peak"), possibly resulting in oscillation and general instability.  Typical values for C104 when using a small-ish photodiode like the BPW34 are 2-10pF:  Using too much capacitance will result in unnecessarily rolling off high frequency components, but will not otherwise cause any problems.  A small trimmer capacitor may be used for C104, either "dialed in" for the desired response and left in permanently or optimally adjusted, measured, and then replaced with an equal-value fixed unit.

Again, the reason why the ultimate in high sensitivity is not required on a "Daylight Tolerant" circuit is that during the daytime, the dominant signal will be due to thermal noise from the sun - a signal source strong enough that it will submerge weak signals, anyway:  It need be sensitive enough only to be able to detect the sun noise during daylight hours.

The op amp noted in Figure 4 is the venerable LM833, a reasonably low-noise amplifier and one that is cheap and readily available (and actually works well down to 7 volts - a bit below its "official" voltage rating allowing the above circuit to powered from a single 9 volt battery) but practically any low-noise op amp could be used:  Somewhat better performance may be obtained using special, low-noise op amps, but these would be "overkill" under daylight conditions.

For nighttime use - where better sensitivity was important - a standard "TIA" amplifier that omits the DC feedback loop to cancel out the DC (potentially noise-contributing) components along with higher values of Rf would offer better performance, but for much better low-noise performance (e.g. 10-20dB better ultimate sensitivity) under low-light conditions than is possible with standard components at audio frequencies in a TIA configuration the "Version 3" optical receiver circuit described on the page "Gate Current in a JFET..." - link is recommended, instead.

Additional web pages on related topics:
The above web pages also contain links to other, related pages on similar subjects.


This page stolen from "".

Friday, March 31, 2017

A (somewhat convoluted) means of locking a "binary" (2^n Hz) frequency to a 10 MHz reference

DDS (Direct Digital Synthesis) chips are common these days with small boards containing an Analog Devices AD9850 board being available on EvilBay for a cost lower than one is likely able to buy the chip by itself!  While these boards are quite neat, they do have a problem (or quirk) in that you are not likely to be able to generate the exact frequency that you want - at least if it is to be an exact integer of Hz.

Let us take as an example one of those ADS9850 DDS boards available on EvilBay.  These come equipped with a 125 MHz crystal oscillator that will likely be within 10-20 ppm or so, but let us assume that it is exactly 125 MHz.

Other than the 125 MHz clock and some output filtering, the AD9850 DDS chip has nearly everything else that one would need to generate an output from DC to around 60 MHz - the precise limit depending on filtering - and this frequency is set using a 32 bit "tuning word".  The combination of the 125 MHz clock and the 32 bit tuning word means that our frequency resolution is:
  • 125,000,000 / (232) = 125,000,000 / 4,294,967,296 = 0.02910383045673370361328125... Hz per step - approximately.
For most purposes around 1/34th of a Hz resolution would seem to be good enough - and it probably is - but what if you wanted to be able to generate frequencies that were exact multiples of 1 Hz steps for frequency comparison purposes or to be able to generate precise, standard frequencies like 1, 5, 10 MHz, etc. - or even a very precise 1 kHz tone?

The quick answer to this is to pick a clock frequency that is an exact "power of two", and the closest 2n multiple to 125 MHz is 227 or 134.217728 MHz - slightly beyond the ratings of the AD9850, but it is likely to work.  (Depending on the high frequency requirements, half this frequency - 226 Hz, or 65.108864 MHz could be used instead:  Other frequencies that are 2n divided by an integer such as 2n/10 are usable, too as an example.)

What does this change in clock frequency gain for us, then?
  • 227 / 232 = 0.03125 Hz per step, which is exactly 1/32nd Hz.
In this way, very precise frequencies that are a multiple of 1 Hz (and a half-Hertz as well) could be produced.

(Where does one get a 134.217728 or 65.108864 MHz oscillator?  This would likely require a custom-made crystal/oscillator or it could be produced using another synthesizer such as an SI5351A that, itself, uses a VCXO.)
Locking the DDS synthesizer to a 10 MHz frequency reference

It would make sense that if you actually needed to be able to set your frequency to exact 1 Hz multiples that you would also need to precisely control the reference frequency as well - likely with a 10 MHz precise reference from a GPS Disciplined Oscillator (GPSDO), a Rubium frequency reference or something similar.  Unfortunately, 227Hz is an awkward number that doesn't easily relate to a 10 MHz reference.

The most obvious way to do this is to use a second DDS generator board (they are cheap enough!) clocked from the same 227Hz source with its output to exactly 10 MHz using a frequency word of 320,000,000d, comparing it to the local standard and applying frequency corrections to the 227Hz oscillator.

There is a less-obvious way to do this as well:
  • Take the 10 MHz output and divide it by 625 to yield 16.000 kHz
  • Multiply the 16.000 kHz by 32 to yield 512.000 kHz
  • Divide 512 kHz by 125 to yield 4096 Hz
  • Divide any 2n Hz frequency down to 4096 Hz as a basis of comparison
(Depending on one's requirements, the precise method could vary with other frequency combinations possible.  The frequency of 512kHz was used because it was well within the operational range of good, old-fashioned 4000 series CMOS circuitry.)

Why would anyone use this second method?  Back in the 1980s I built a DDS synthesizer that used a 224 Hz reference (16.777216 MHz) that used a 24-bit tuning word to provide precise 1 Hz steps, but I also needed to lock that same synthesizer to a high-quality 10 MHz TCXO.  While it would have been possible to have built another synthesizer, a 1980s solution to this problem meant that an entire synthesizer circuit (or most of it, anyway) consisting of more than a dozen chips - some of them rather expensive - would have have to be replicated to do just one thing.

This seemingly convoluted solution required required only 6 inexpensive chips - a combination of 74HC (or LS-TTL) and some 4000 series CMOS devices.  For example:
  • Dividing the 10 MHz reference by 625:  A 74HC40103 wired as a divide-by-125 followed by a 4017 counter wired as a divide-by-5 to yield 16 kHz.
  • The multiplication of 16 kHz by 32 to 512 kHz:  A 4046 PLL and a 4040 counter to form a synthesizer.
  • Division of 512 kHz to 4096 Hz:  Another 40103 wired as a divide-by-125.
  • Division of 16.777216 MHz down to 4096 Hz:  A 74HC4040 counter dividing by 4096.
The final step to lock the two frequency sources together was to use the venerable 4046 phase detector, outputting the correction voltage to the 16.777216 MHz oscillator.

It's worth noting that because the 4096Hz output from the divide-by-125 from the 512kHz source is a pulse rather than a square wave so it is not possible to use the "XOR" phase detector (Phase detector 1) of the 40406, but rather the flip-flip detector (Phase detector 2).  The "problem" with the flip-flop detector is that when the two frequencies are close, instead of having a constant train of pulses being output that are either at the reference frequency or twice the reference frequency, one will get occasional, brief pulses as the output of one of the flip-flops occasionally drops out of its high-impedance mode.  The problem with is that these occur (more or less) randomly and comparatively rarely, meaning that they they are at a rather low frequency and can get through the loop filter, causing extra jitter on the locked frequency - the 16.777216 MHz oscillator in this case.  The "fix" for this is to slightly bias the output of the phase comparator toward V+ or ground with a high-value resistor (100k-4.7 Meg, depending on the application) which will "pull" the output constantly toward one rail, forcing the loop to be corrected constantly meaning that instead of the occasional, narrow pulse, there will always be a string of pulses at a "high-ish" frequency that can be removed by the loop filter.
With the main 16.777216 MHz reference being a VCXO (Voltage-Controlled Crystal Oscillator) the above scheme worked very well, locking to the 10 MHz reference in under a second.  Back in the 1980s the most accurate frequency reference that I had was a collection of OCXOs (Oven-Controlled Crystal Oscillators) and TCXOs (Temperature-Controlled Crystal Oscillators) with the 10 MHz units being easily referenced to the off-air signal from WWV to provide both an accuracy and stability of around one part in 107 or better.  Because, in our example, we are starting out at a much higher frequency (e.g. 134-ish MHz) we would would divide this down to 4096 Hz using a combination of 74F or 74Axx logic and a (74HC)4040 counter.

(If our 134-ish MHz clock were produced using an SI5351A synthesizer, the PLL corrections in this scheme would be applied to its clock, which typically operates at around 27 MHz.)

Nowadays, with GPSDOs and second-hand rubidium references being affordable, the accuracy and stability can be improved by several orders of magnitude beyond this.

Having said all of this the question must be asked:  Is any of this still useful?  You never know!


This page stolen from

Tuesday, March 21, 2017

The 1J37B as a replacement for a 1L6?

The rarity of the 1L6:

Owners of the classic Zenith Transoceanic radios from the early-mid 50's will probably be aware of the pain involved if they have to buy a 1L6 tube (or "valve") for their beloved radio:  A "good" 1L6 - seemingly the tube that goes bad most often in these radios - can fetch up to $60 today, a significant fraction of what one might have paid for a second-hand radio to restore.

Figure 1:
The original 1L6 - a "not-too-common" tube even during its
heyday.  A "good" one like this is even rarer today!
Click on the image for a larger version.
One of the problems with the 1L6 is that there really aren't any good substitutes since this tube, a hexode  (a.k.a. "pentagrid converter") wasn't  commonly used in the first place, finding near-exclusive use in higher-priced battery-powered shortwave radios.  One of the few (almost) direct plug-ins that exists is the 1U6 which is apparently rarer than the 1L6 and requires some slight circuit modifications to accommodate its lower (25 mA) filament current.

There are other tubes that will plug in, but these simply don't work on the higher shortwave bands (e.g. the 1R5, intended for AM broadcast band battery portables and not for the higher shortwave frequencies) or, in the case of the European 1AC6, requires a bit of modification and has issues with radio alignment.  There is, of course, the electrically equivalent and comparatively easy-to-find 1LA6, but it's in a completely different form factor (e.g. a loctal tube rather than a 7-pin miniature) and requires either an adapter or a different tube socket.  Finally, there are the solid-state replacement options which are roughly comparable to the cost of a known-good 1L6 and while some work fairly well, they definitely lack that "tube" aura.

What now?

One of the sticking points is that the 1L6 serves both as the local oscillator and frequency converter:  One of the internal grids of this hexode is used as sort of the "plate" of the oscillator while a grid closer to the anode takes the signal from the RF amplifier stage and modulates the electron stream to mix it with the local oscillator to produce the 455 kHz IF - and it does this all with a filament that consumes just 50 milliamps at about 1.25-1.4 volts.  Without significant rewiring, this kind of rules out the use of pretty much any tube other than one that takes just 50 milliamps at 1.4 volts for its filament!

Having established that there really aren't any other 7-pin miniature tubes that are "close enough", what about broadening the scope to include something entirely different?

Figure 2:
The Russian 1Ж37Б "rod" pentode.  Approximately the same diameter
as a ball-point pen, it's overall length, minus leads, is about that
of a 7-pin miniature tube.  This specimen bears an early 1987 date
of manufacture.
Click on the image for a larger version.
This thought came to me at about the time I was first experimenting with some Russian Rod tubes as described in my December 31, 2016 posting, "A simple push-pull amplifier using Russian Rod tubes and power transformers" - link.

While that article discusses the use of a 1Ж18Б (usually translated to "1J18B" or "1Zh18B") pentode, there is another member of that family, the 1Ж37Б (a.k.a. 1J37B or 1Zh37B) that is also a pentode rated for operation to at least 60 MHz.  One property in its favor is that its filament voltage and current are "pretty close" to that of the 1L6:  Anything between 0.9 and 1.4 volts will work and the rated filament current is around 57 milliamps - a tad higher than the 1L6, but something that we can probably live with.

Doing a quick finger-count of the number of elements of a pentode and comparing that with the number of tube elements that one would need to simulate a 1L6 hexode immediately reveals a problem:  How would one use a pentode as a pentagrid converter when we are an element short?

The 1J37B to the rescue?

As it turns out, the 1J37B is a unique animal:  As a result of its construction using metal rods to form and modulate sheets of electrons rather having the grid-like structures of "conventional" tubes, it actually has TWO "first" grids that are pretty much identical - a construct that is often likened to that of a dual-gate MOSFET.
Figure 3:
The bottom-view pin-out and the internal diagram of the 1Ж37Б pentode.
Following the original nomenclature, the "grids" are referenced using the "C" designation - somehow appropriate even in English since this tube does not use "grid" structures at all, but control rods to alter the trajectory of sheets of electrons from the cathode.  As noted in the text there are two "first grids" that operate identically and (in theory) may be used separately, interchangeably or even tied together as a single "grid" with higher transconductance.  Because these tubes manipulate sheets of electrons, they are quite sensitive to magnetic fields!
Click on the image for a larger version.

The internal mechanical layout of the 1J37B is also quite interesting in that it is essentially two tubes in parallel, sharing the same cathode, screen "grid" suppressor "grid" and plate connections.  In the middle, the identical sheets of electrons from the cathode go in two directions, each controlled by its very own "C1" control rod (e.g. C1' and C1").  Beyond C1' and C1", the structures of the screen, suppressor and plate elements are physically mirrored and connected together.

In comparing the specifications of the 1L6 and the 1J37B, the important specifications  (e.g. transconductance, capacitance, filament voltage and current) weren't terribly far off.  Some of the voltage ratings for the 1J37B - particularly that of the screen, rated for 60 volts maximum - are below that which one would see when used as a 1L6, but those may be dealt with later.

What if we could use one of these two "first grids" and the "screen grid" as the basis of the local oscillator section and simply apply the input signal to be amplified and converted to the other "first grid"?  Because it was more like two tubes in parallel than one tube with multiple control grids I wondered if there was enough isolation to allow both oscillating and signal mixing functions to occur simultaneously.  I was a bit skeptical if this idea, even though I was the one that thought of it (as far as I know.)

I decided to try it.

Making the base
Figure 4:
Using masking tape, a "form" is made to set the shape and position of the
pins the pieces of 18AWG wire poking through two layers of masking
tape to protect the socket.  After dripping in the epoxy, the pins were
moved about to make sure that they were completely surrounded by
Click on the image for a larger version.

Rather than mess with the Zenith TransOceanic for the first attempt at this, a friend of mine (Glen, WA7X) rummaged through his collection of old radios and produced an old Motorola battery/AC radio that used 1 volt tubes - including the 1R5 which is (sort of) "pin compatible" with the 1L6.  Being a broadcast band radio I figured that if the concept was usable at all, the simple, nearly foolproof low-frequency circuits of such a radio would be the place to try it first:  If it worked there, there may be some hope that it would work in the ZTO.

I needed to make a fake tube base, but not having a dud 7-pin miniature tube immediately at hand - and remembering from my past how difficult it is to solder to the "bloody stumps" of the dumel-like wires on the carcass of a deceased tube' base - I set about making one.  I first covered the 7 pin socket in the radio with two layers of masking tape and then poking through this tape and into the socket seven lengths of bare, 17 or 18 AWG copper wire.  A ring of masking tape was then placed around the outside of these pins and some "5-minute" epoxy was dripped into the middle, carefully avoiding the copper "pins":  No doubt a small piece of plastic tubing or a taped-together ring of a sheet of plastic from a discarded "blister pack" would have made a nicer form than a floppy piece of masking tape, but it did the job.
Figure 5:
After the epoxy had started to set up, it was heated to speed up curing.  After
it had adequately set, it was removed from the socket:  Here it is before
the wires were trimmed and tape and excess epoxy were removed.
Click on the image for a larger version.

Working the copper pins back and forth to make sure that they were surrounded with epoxy I allowed the requisite "5 minutes" for the "fast curing" epoxy to (somewhat) set. I then heated the contrivance with an SMD hot-air rework gun on its lowest heat (212F, 100C) for several minutes which immediately caused the epoxy to set hard enough to work once it had again cooled.

Carefully removing the "base" from the socket and peeling away some of the masking tape I trimmed the seven wires underneath to lengths comparable to that of a typical tube and did similar to the top side.  I then had my 7-pin, solderable "tube base".

From this point on the wiring of the 1J37B to the base seemed pretty straightforward..

Wiring it up:

For the initial stab at replicating the function of a 1R5 the 1J37B was wired to the 7-pin base as follows:

1J37B Pin                  [7 pin base connection for the 1R5]
1 - Filament (-)                 [Pins 1]  Filament and suppressor grid
2 - "Grid" 1'                      [Pin 4]  "Oscillator Grid" (G1)
3 - Grid 3 (suppressor)     [Pins 1]  Filament and suppressor grid
4 - Filament (+)                [Pin 7]  Filament
5 - "Grid" 1"                     [Pin 6]  "Signal Grid" (G4)
6 - "Grid" 2 (Screen)        [Pin 3]  "Oscillator plate/grid" (G2)
Plate wire (top)                 [Pin 2]  Plate

Or, put another way:

7 Pin base connection   for the 1R5    [1J37B Pin connection]
1 - Filament (-) and Suppressor Grid    [1 - Filament (-) and 3 - Suppressor Grid]
2 - Plate                                                 [Top plate wire]
3 - 1L6 "G2"                                         [6 - Screen Grid]
4 - Oscillator Grid (1L6 "G1")              [2 - Grid 1']
5 - No connect (see text)                       N/A
6 - Signal Grid (1L6 "G4")                   [6 - Grid 1"]
7 - Filament (+)                                     [4 - Filament (+)]

Again, note that applying the word "grid" to the 1J37B, while descriptive of the function, is not accurate:  These "grids" operate more as control rods to deflect/direct the sheet of electrons being emitted from the cathode.

For replacing a 1R5:
Figure 6:
Right at home, the completed 7pin miniature tube base in the Motorola
"test" radio in the 1R5's position.
Click on the image for a larger version.

A bit of explanation about pins 1 and 5 is in order at this point.  For the 1L6, pin 5 connects to a pair of grids that surround the "Signal" grid (1L6 pin 6), but on the 1R5 the suppressor grid is internally connected to the "low" side of the filament using pins 1 and 5. Because the 1J37B is a pentode, the suppressor grid must be grounded which means that it would be connected to the filament low side as well.

Whoever made the radio could, in theory, use pin 1 and/or pin 5 for this connection and there is no real way of knowing without visually inspecting the socket.  Because of this it would be a good idea to connect both pins together when emulating a 1R5 unless you know for certain how this connection is made in the radio with which you are testing.

For replacing a 1L6:

When using a 1R5 as a "pinch hit" replacement for the 1L6 the voltage applied to pin 5, which is nominally at about 85 volts, is effectively shorted to "ground".  In the Zenith TransOceanic H-500 there is a 68k resistor in series with that line which means that the current will be around 1 milliamp or so, dropping the "85 volt" line - also used on the screen of the RF amplifier - by 3-5 volts, an amount likely not enough to be noticed.  If the intent is to never use this replacement in lieu of a 1R5 we would just leave pin 5 disconnected.

Trying it out as a 1R5:

For testing it out in the "1R5" configuration (e.g. 1R5 pins 1 and 5 connected together) in the Motorola radio I inserted a 10k resistor in series with the anode lead in order to monitor its current, but despite this inserted loss the faux 1R5 worked the first time.  The filament voltage across the 1J37B was 1.0-1.1 volts, well within its operational specifications and indicating that the tube in series with it across its 3 volt "A" battery (a 1S5) was probably seeing an extra 0.25 volts or so across its filament.
Figure 7:
The first prototype - the 1Ж37Б (a.k.a. 1J37B) wired to the 7-pin miniature
base as a "1R5".  The two 10k parallel resistors and 0.01 capacitor
were inserted into the plateto monitor current.  For this prototype the leads,
insulated with PTFE spaghetti tubing, were intentionally left at their original
length to facilitate rewiring and inserting other components (resistors, capacitors,
etc.) in the circuit during testing.  For a "final" configuration the leads would
be shortened considerably.
Click on the image for a larger version.

There was a minor problem, however:  At some frequencies the radio would start squealing - something that it did not do with the 1R5.  It is possible that there is a failing component in this radio somewhere, or it may also be that this faux 1R5 has enough extra gain to cause circuit instability, or a combination of both.  Despite this minor quirk, the results were encouraging as it is usually easier to dispose of extra gain than obtain it in the first place.

As a 1L6:

I then decided to try this faux 1R5 in my Zenith TransOceanic H500 with pins 1 and 5 still connected together.  While it seemed to work fine on the AM broadcast band, the radio got increasingly deaf with each higher band.  A quick peek with a spectrum analyzer on a service monitor showed that the oscillator was working on all bands, but it was always a low in frequency, causing mis-tracking of the RF filtering with the error increasing as one went up being low by about 600 kHz on the highest (16 meter) band.

There was another problem:  On 19 meters the radio started to oscillate, behaving like a regenerative receiver on the verge of oscillation and on 16 there was just solid hash, indicative of instability - likely because of excess gain.  Referring back to the 1J37B specifications, I'd noted before that the noted maximum indicated screen voltage was on the order of 60 volts - but nearly 90 volts was being applied in the TransOceanic.  Because of the rather low current pulled by the screen grid (being used as the "plate" for the local oscillator) and the still-within-specs amount of plate current (around 3 milliamps).  I wasn't particularly worried about violating this voltage rating as there is no actual delicate "grid" that can be damaged, but it occurred to me that the gain could be reduced a bit by lowering the screen potential.  With a bit of experimentation I determined that a 33k resistor paralleled with a 1000pF capacitor in series with pin 3 of the 1L6 socket reduced the screen voltage to around 65 volts - still a bit above its specifications - but this change resulted in unconditionally stable operation.

Disconnecting the now-unnecessary pin 5 and wielding an alignment tool I went to work re-tweaking the radio.  For all but the 16 meter band, the local oscillator adjustment was well within the range of the various coils and capacitors, but for 16 meters, removal of the local oscillator's slug only brought it to within about -400 kHz of where it should have been.

On the lower bands, particularly AM Broadcast, 2-4 MHz, 4-8 MHz and 31 meters, the radio's sensitivity was reasonably good - not quite up to that of the 1L6 on 31 meters, but perfectly usable nonetheless.  For the higher bands, 25 and 19 meters, I could still hear a bit of ambient atmospheric noise and those radio stations for which propagation was extant, but like 31 meters, the receive sensitivity was still a bit low indicating the need for yet more tweaking.

More tweaking and testing:

I later did a bit more experimentation, adjusting bias and re-dressing the leads, but I could not affect the 16 meter tuning range significantly enough to bring it back into dial calibration, nor could I make a "dramatic" improvement in the high-band sensitivity.  If I'd replaced the core in the 16 meter oscillator coil with an aluminum or brass slug I may have been able to drag it up to frequency, but I didn't try it.


I did prepare another 1J37B tube and wired it in an identical manner to the first shown in the previous pictures (but with shorter leads) but interestingly it behaved remarkably different than the first:  It seemed to be much more prone to bouts of spurious oscillation (e.g. broadband noise) and fitful, intermittent local oscillator operation - a state not dramatically affected by swapping the two "first grids" C1' and C1".  Otherwise, the tube seemed to be behaving about the same in terms of DC current as the first.

What this told me is that my initial configuration of using the "screen grid" as the oscillator plate and applying the RF signal to be mixed to the other "first grid" may not be the best approach, as was my initial hunch - particularly in light of the fact that two seemingly identical tubes, both with fairly similar DC characteristics, seemed to behave radically different in this circuit - a strong indicator of a "non optimal" circuit topology that required on "quirks" of each, individual tube!

In the future I may reconfigure the circuit a bit to see if configuring the tube in some sort of "Gammatron" configuration may yield better results - but that will have to wait until I get more free time...

* * *

Additional information about the 1J37B and the "Gammatron" mode of tube operation:

  • The 1J37B at the Radiomuseum - link (Includes discussions about operating the tube as a Gammatron.)
  •  Russian rod tubes at "Radicalvalves" - link (Information about the 1J37B and other "rod" tubes.)


This page stolen from

Friday, February 24, 2017

Fun with self-oscillating TV flyback transformer circuits, arcs and high voltage

A few weeks ago I ordered a few things from The Electronic Goldmine and one of the items that I picked up was a small flyback transformer (Stock #:  G20787, manufacturer part number BSH12-N406L) as would have been used in a small CRT (Cathode Ray Tube) television.  In perusing the internet I was able to determine that this transformer was originally intended for a small Black-and-White TV with a nominal anode voltage of around 12kV.

Figure 1:
Drawing a 1/2" (1cm) arc from the contraption.
Click on the image for a larger version.
Having a back-burner project that will need 8-12 kVDC at a very low current I decided to mess about with a simple, self-exciting oscillator circuit.  Before I go on, I need to throw out a few "weasel words":

This project deals with high voltages - possibly in excess of 12 kV.  While the current is low, it is still possible for the output to cause fire, injury - directly or indirectly - or even death.
Any experimentation or use of the circuit(s) described on this page should be done with extreme caution and only by persons familiar with high voltage safety.

You have been warned!
In a television these transformers are driven externally at a specific horizontal frequency - usually between 15.6-15.8 kHz - but with a small number of components a self-contained "power" oscillator can be assembled, operating over a wider range of frequencies and capable of producing high voltages.

The circuit and (my arbitrary) pin connection for this transformer is shown in Figure 2, below.

Figure 2:
Self-oscillating flyback transformer driver.  Like most modern flyback transformers, this unit contains a high-voltage rectifier - which may also be part of an internal capacitor-diode voltage multiplier.  Capacitor C1 is semi-optional, but is highly recommended to reduce the amount of switching frequency energy from appearing on the V+ line.  The pin-out diagram is specifically for the BSH12-N406L flyback transformer (Electronic Goldmine P/N:  G20787).
This circuit operated from about 3 to 15 volts with higher supply voltages yielding greater high-voltage output:  R1 and R2 would be tweaked for optimal operation at the desired supply voltage and the specific transistor used for Q1.
Click on the image for a different-sized version.

The pin-out in Figure 2 is specific to this particular transformer but similar arrangements may be divined with most other flybacks from solid-state televisions with an ohmmeter and the use of clip-leads to find the optimal connections.  What is common to most flybacks is that one or more of the pins on the bottom will appear to not be connected to anything else, but one of these will probably be the bottom end of the high voltage winding with one or more of the others may be used for focus voltage or similar.

The starting values for R1 and R2 would be 1k and 270 ohms, respectively, but this would be adjusted for best performance with the operating voltage, expected load, specific transistor and flyback transformer that was used.  In testing, these resistor values were found to work between 4 and 16 volts - albeit, not necessarily optimally.  The use of capacitor C1 is strongly recommended and it is suggested that a "Low ESR" type as found in switching supplies be used.

Transistor Q1 was a 2SC4130 pulled from a junked switching power supply and was used because it was free.  Because this is an oscillator and this transistor's original use was in a switching supply - and with its high voltage rating - it was particularly suitable for this application.  The specific transistor isn't particularly important and almost any NPN power device will work, preferably one that is rated for over 100 volts, but some seem to work better than others for reasons that aren't immediately obvious so it's worth trying a few different devices.  No matter which transistor you use it is a good idea to heat sink it if it will be operated under any load for more than a few seconds.

For what purposes would one use this sort of circuit?

Aside from making pretty arcs or producing coronas and lots of ozone, these sorts of voltage (6-12kV) at the low currents of which a set-up like this is capable could be used for "lighting up" an image intensifier (a.k.a. "night vision") tube, for "electrostatic wind" experiments, to mildly charge objects so that they are attracted to each other (e.g. paint, glitter, etc.), to "strike" and light small HeNe laser tubes (with the appropriate ballast resistor) or to briefly test gas discharge tubes such as neon displays to verify their seal integrity.

What's the voltage, Kenneth?

Voltages like this aren't particularly difficult to measure, rather they are awkward.  They are far too high for all but the most specialized of voltmeters (you risk damage if you try!) so the most appropriate tool for this would be a high voltage probe as is used to measure voltage on a cathode ray tube.  Usually around a foot (25cm) long and with a separate ground lead, these may be had second-hand, particularly now that cathode-ray devices are becoming a rarity.

It is possible to use resistors to make a divider to measure this voltage, but there's a catch:  Most common resistors are rated for only 250-1000 volts (at most!) drop across them, the rating depending both on how they are made and the wattage/physical size.  As an example, if you wanted to use 10 Megohm, 1/2 watt resistors, you'd need to wire at least twenty of them in series to achieve a nominal 10kV safety rating, assuming a 500 volt rating per resistor:  Check the specs!

In my case I rummaged about and found a bunch of 10-20 Megohm, 2 watt carbon composition resistors (safe for1000-1500 volts, each) and wired them in as a divider to get an approximate voltage measurement.  Even though the resistance was in the 100+ Megohm range, I could tell by the reduction of the arc length and the amount of current being drawn from the power supply that this was loading the output and significantly reducing the voltage meaning that with no load at all, the voltage was higher, still.

Remember:  For whatever purpose you intend to use it, be careful!


This page stolen from

Saturday, February 11, 2017

A novel APD-based speech bandwidth optical receiver

In a previous posting I wrote about a novel application of a JFET (Read about that in the article "Gate current in a JFET - The development of a very sensitive, speech-frequency optical receiver" - link) in which the flow of gate current was integral to the operation of a photodiode-based optical detector.  In testing this circuit, which included an indoor "photon range" and out in the field, it was observed that the sensitivity of this circuit was, at "audio" frequencies, on the order of 8-20 dB better in terms of signal/noise ratio than any of the more conventional "TIA" (TransImpedance Amplifier - read about that circuit here - link) circuits that had been tried.

In the analysis of this circuit it was determined that several factors contribute to the ultimate sensitivity, including:
  • The intrinsic noise of the JFET.  This can be minimized by hand-selection of the device itself for the lowest-possible noise as well as selecting a device that can operate at a higher drain current to reduce the "bulk noise" - or even the use of several JFETs in parallel.
  • The contribution of noise by other circuitry.  In the design this was minimized through the use of a cascode circuit topology as well as the use of a low noise, high impedance current source to supply the bulk of the drain current and the complete avoidance of other components being connected to the photodiode-JFET circuit junction.
  • The capacitance of parasitic circuit elements, such as capacitance (including the Miller effect) that reduces the amplitude of the signals from the photodiode, particularly as the frequency increases, effectively reducing the signal-noise ratio.
  • The contribution of the photodiode itself.
Of these factors, the majority of the noise would appear to be due to the JFET itself, particularly above the low audio frequencies frequencies (e.g. below 100Hz or so) where 1/F noise would dominate. One of the possible approaches to get better noise performance is to cool the circuitry, but this is fraught with difficulties related to condensation which would require that the device itself be sealed in an atmosphere (e.g. dry nitrogen) in a manner similar to that used to cool CCD imagers for astronomy.
Figure 1:
The outside view of the completed APD-based optical receiver.  Because
of its extreme sensitivity it must be well shielded to minimize the pick-up
of stray fields such as those from AC mains and radio transmitters/phones.
Click on the image for a larger version.

What else may be done to improve the performance?

Perhaps counter-intuitively, the use of a smaller photodiode can help a bit and provide at least as much signal output as a larger one, provided that the optics can focus the given amount of light from the distant source of light efficiently onto its active area:  A smaller device will have lower self-capacitance shunting a smaller amount of the AC currents being produced in response to the impinging, modulated light in addition to having a lower intrinsic noise contribution.  In the case of an optical receiver the active area of the device is less important than in some other applications as lenses and mirrors may be used to concentrate the light from the distant source onto the photoactive area.

When reducing the size of the device one must assure that the optics themselves will resolve the distant spot of light to an area that is not larger than the active area of the device as well as taking into account additional constraints with respect to the accuracy and stability of the aiming and pointing mechanisms.  For example, using reasonable-quality molded Fresnel lenses of common focal lengths (e.g. an f/D ratio of approximately unity) one can expect only to resolve a spot with a "blur circle" of approximately around 0.2mm at best while high-quality glass optics should be able to reduce this by an order of magnitude or better assuming a suitably-distant source, a corresponding small subtended angle and proper paraxial alignment and focus.  If the resolved spot of light is much larger than the active area of the device - perhaps due to the device being too small for the optics ability to resolve or due to the quality and/or misalignment of the lens(es) - there may be an additional loss of available optical energy and signal-noise ratio as some of the light from the distant source is being "wasted" when it spills beyond the active area of the photodetector.
For more information on "spot sizes" using inexpensive, molded plastic Fresnel lenses see the article "Fresnel Lens Comparison:  A Comparison of inexpensive, molded plastic lenses and their relative 'accuracy' and ability to produce collimated beams" - link.
Aside from the reduction of the size of the photodiode or cooling, where else may one eke out greater performance from this circuit topology?

The Avalanche Photodiode:

The Avalanche Photodiode (APD) is a type of photodiode that contains an internal mechanism for amplification.  Simply put, instead of a single photon having a given probability of mobilizing a single electron when it impinges the active area of a standard PIN photodiode.  In an APD, what might have been a single electron being loosed in a normal PIN diode that same electron event can cause the mobilization of many electrons via an "Avalanche" effect, providing amplification of the optical signal and hence the name.  The result of this intrinsic amplification is that the output signal from this diode from a given photon flux can be much higher than that of a standard PIN photodiode.

Because the signal from the Avalanche photodiode itself is amplified internally it is more likely to be able to overcome the effects of the capacitance on frequency response as well as the noise intrinsic to the JFET amplifier, support circuitry and components, providing the potential of producing a greater signal/noise ratio for a given signal. Typically an Avalanche photodiode is incorporated into a TIA (TransImpedance Amplifier) with good effect, but what about its use in the previously-described "Version 3" photodiode receiver circuit that utilizes JFET gate current?

The basic design:

From the previous article (link) one can see the basic topology of the "Version 3" circuit using a "normal" PIN photodiode depicted in Figure 2, below.
Figure 2:
A diagram of the "Version 3" optical detector that utilizes JFET gate current.  In this circuit Q1 and Q2 comprise a cascode
circuit with Q3 providing the majority of Q1's drain current while U1b is configured as a differentiator to compensate
for the low-pass effects of the intrinsic capacitance of D1, the photodiode and Q1.  Resistors R1 and R2 along with
C1 provide a filtered reverse bias for D1 which not only decreases its capacitance, but it also biases Q1 to
its operating state where it is drawing maximum drain current.  In this circuit the connection between the Photodiode (D1)
and the gate of the JFET is made in air and not on a circuit board to minimize capacitance, stray signal pickup and
most importantly a source of leakage currents and related noise.
Click on the image for a larger version.
In this design PIN photodiode D1, a BPW34, is reverse-biased via R1 and R2.  One of the main benefits of doing this is that the capacitance of D1 decreases from approximately 70pF at zero volts to around 20pF at the operational voltage, reducing the degree to which high frequency signal are attenuated by this capacitance.  A somewhat less tangible benefit of this is that in addition to photovoltaic currents produced by the impinging light, the bias also allows photoconductive currents to flow from the bias voltage, through the photodiode and into the gate of the JFET.  As noted in the original article it is the presence of the gate-source junction of the JFET (Q1) and its conduction that limits the gate-source differential to around 0.4-0.6 volts, permitting D1's reverse bias to become established without the need of any additional noise-generating or lossy components.  In this configuration the drain current of the JFET is still proportional to the gate-source voltage (but with an offset of drain current greater than the "zero bias" drain current) and like a bipolar transistor's base voltage and current, the relationship between gate voltage and gate current is logarithmic.

What about replacing D1 with an avalanche photodiode?

Testing with an Avalanche Photodiode:

Like its more-sensitive distant cousin, the Photomultiplier tube, the avalanche photodiode requires a rather high bias voltage in order to function at maximum gain.  Rather than requiring a kilovolt or so as is needed for a photomultiplier, typical photodiodes may operate with up to "just" a few hundred volts.  Like the photomultiplier, the current required for "dark" operation is minuscule - a few hundred microamps is more than enough.

In perusing the various component catalogs I noted that Mouser Electronics carried some avalanche photodiodes - but as expected, there was a price - literally:  Around US$150 at the time for just one APD.  In a compromise between size, availability and cost I chose the AD1100-8-TO52-S1 by First Sensor  (previously known as "Pacific Silicon Sensor") - a device with a round, 1mm2 (1.128mm diameter) active area - a reasonable compromise.  This device, which came with its own test sheet, indicated a maximum gain ("M" factor) of approximately 1000 occurring at 134 volts at a temperature of 25C.

In most ways using an APD is just like using a reverse-biased PIN photodiode - except that the reverse bias voltage will be much higher.  Perusing the literature and manufacturer's specifications one will note that many designs depict a temperature-compensated bias voltage supply, but further investigation reveals that this is necessary only if the device is being used at/near maximum gain (and maximum voltage) and/or if it is necessary to precisely maintain a precise gain over a wide temperature range.  For our application, we don't really care if the gain changes with temperature, so an arbitrarily adjustable high voltage supply is fine - and actually preferred.

In my initial research I noted that the internal action of any APD suffers an inevitable, but expected, effect:  As the gain goes up with increasing bias voltage, the intrinsic noise of the device itself increases at a faster rate than the gain.  What this means was that there is going to be a point above which a further increase of device gain will cause the signal to noise ratio to decrease even though the actual signal level continues to increase with bias voltage - but at what voltage might this happen, and would this "crossover" point occur at a point where we can expect the overall "gain+noise" to offer a net advantage over a PIN photodiode?

Building a prototype receiver similar to that depicted in Figure 2 I substituted an APD for D1 using a string of sixteen 9 volt batteries and a 1 megohm potentiometer with a 100k resistor in series with the wiper (and some bypass capacitors to ground on the "hot" side of the diode) in lieu of R1 to set the bias voltage.  Placing this prototype in my "Photon Range" - a windowless room in my house where there is an LED mounted to the ceiling that may be modulated - I compared the sensitivity of this prototype to both my "standard" TIA receiver (the VK7MJ design) and an operational exemplar of my "Version 3" design.

Varying the voltage from 10 volts to around 140 volts I noted that at a bias voltage comparable to the reverse bias applied in Figure 2 (approx. 8 volts) the apparent sensitivity was roughly on par with that of the Version 3 unit using a normal PIN photodiode after the signal levels were corrected to compensate for the smaller area of the APD as compared with the BPW34 (e.g. 1mm2 of the APD versus 7mm2 of the BPW34 - the larger size gathering proportionally more light in this lens-less system).  At around 130-135 volts, the output of the APD-based prototype was very high, but the weak, optical signals from the test LED were lost in the noise.  In the area of 35-45 volts I observed that while the overall signal levels, while significantly higher than they were at 8-10 volts, were a fraction of what they were at 130 volts but the signal/noise ratio was roughly 6-10dB higher than it was at the lowest voltage when the differences in active area of the APD versus the photodiodes in the test receivers were taken into account.

  • The test receivers used BPW34 PIN photodiodes with an active area of 7mm2 while the APD has an active area of just 1mm2.  Because there are no optics in front of the photodiodes there will be 7 times as many of the LED's photons hitting the larger device, resulting in an approximate 8.5 dB difference in signal/noise - assuming all other parameters being equal.  It is when using the device in this "lens-less" configuration that this factor must be accommodated.
  • While it is theoretically possible to use a photomultiplier tube (PMT) in lieu of an APD, there are several practical concerns.  Even though an "S-1" type of photocathode has a peak in the red-NIR area, its low quantum efficiency makes it a rather poor performer overall.  The "931A" PMT - widely available surplus - has a more typical blue/violet peak response (type "S-4") in which the longer red wavelengths suffer greatly in terms of quantum efficiency.  Field testing of these devices by British amateur radio operators has shown that they offered no obvious advantage over the "Version 3" PIN photodiode design for "red" wavelengths.  As of the time of this writing the use of PMTs with more exotic photocathodes (such as multialkalai and GaAs) that are better suited for "red" wavelengths (but much more difficult to find surplus!) have not been field-evaluated.
A practical design:  The high voltage APD bias supply:

First, a few weasel words:
Even though the currents are very low, there is some risk of injury with the voltages involved (e.g. several hundred volts) and it is up to you to educate yourself about high voltage safety!
If you wish to construct these circuits, be aware of possible hazards and always assume that any capacitors are charged, even after power is removed.

You have been warned!
Because it is not convenient to carry around a lot of 9 volt batteries, a simple high voltage converter was designed to provide the  microamp-level current required for the APD bias supply and is depicted below in Figure 3.
Figure 3:
High voltage supply for the APD receiver.  U101a is an oscillator that drives Q101 to produce a high-voltage, low-current bias for the APD.  The output is regulated via U101b and associated components to the voltage set by potentiometer R111.  R109 is used to set the highest voltage that may be obtained when R111 is adjusted for "maximum".  R109 is shown with the two "ends" grounded only because it was convenient to wire it this way when the prototype was built.
Click on the image for a larger version.
This design is a simple "boost" type switching converter using a high voltage transistor and an inductor to produce the needed bias.  In this circuit U101A forms an oscillator that drives the high voltage transistor Q101, and when Q101 switches off, the magnetic field of L101 collapses, producing a high voltage spike that is rectified by D101 and filtered and stored by C102, R106 and C103.  To regulate this high voltage a sample is divided-down by R108 and R109 and compared with a 5-volt reference from U102 that is made variable with R111:  If the output voltage is too high, U101b turns on Q102 to pinch off the drive for Q101.  Because I used an "ordinary" op amp with an output that could not go all of the way to the negative supply rail, LED101 was put in series with the transistor's base to provide a drop of around 2 volts to assure that Q101 could be shut completely off.
Figure 4:
Inside the high voltage (bias) supply for the APD receiver.  Potentiometer
R111 and the indicator, LED101, are mounted in the front of the
case.  Both the high voltage generator and the receiver itself are powered
from a single 9 volt battery.  The typical combined current consumption
for the both sets of circuits is less than 35 milliamps.
Click on the image for a larger version.

LED101 also provides two other features:  It functions as a "power on" indicator, and since it is in series with Q101's base drive it is modulated at approximately 6.5 kHz (determined by experiment to be the frequency at which Q101 and L101 produced the highest voltage with the best efficiency) and can be used as an optical signal source to verify that the receiver is working.  Worth noting is that R112 is placed across the "hot" end and the wiper of R111 to "stretch" the high voltage end of the linear potentiometer's adjustment range a bit to compensate somewhat for the fact that near the maximum voltage, the gain goes up exponentially with the bias voltage, making fine adjustments at this setting easier.

The APD (optical) receiver:

The actual optical receiver section is depicted in Figure 5, below:
Figure 5:
The optical receiver which works in a manner very similar to that depicted in Figure 3.  In this implementation
the high voltage bias is applied to the cathode of D201, the APD, which has its anode connected to the gate of the JFET,
Q201.  Q201 and Q203 comprise a self-biasing, AC-coupled cascode amplifier while Q202 provides the a high-
impedance source for the bulk of Q201's drain current.  The components in the sections marked "HV Filter"
and "LV Filter" are used to keep the residual switching frequency energy from being conducted into these circuits.
As with other circuits of this type, the connection from the photodiode to the JFET's gate is made in air and not via a
circuit board trace - this, to minimize capacitance, leakage currents and noise.
Click on the image for a larger version.

Not surprisingly this circuit looks very similar to the "Version 3" optical receiver of Figure 2.  Notable features include an R/C filter consisting of R201, R202, C201 and C202 to remove traces of the 6.5 kHz power supply ripple from the high voltage supply while L201, C211, R215 and C212 do the same for the 9 volt supply that the receiver circuitry shares with the high voltage generator.  The two sections - high voltage supply and optical receiver sections - are separate, connected by a 3 foot (1 meter) umbilical cable, both to provide isolation of the extremely sensitive optical receiver from the electrostatic and electromagnetic fields of the high voltage converter and also to remotely locate the controls on the high voltage supply away from the lens assembly on which the receiver portion is mounted so that adjustments can be made without disturbing it.
Figure 6:
Inside the receiver portion of the APD receiver.  This section is physically
separated from the high voltage converter to prevent the switching energy
from getting into these extremely sensitive circuits.  In the center is
a small sub-board with the APD and JFET that is mounted on short pieces
of 18AWG wire to allow its position to be adjusted in all three dimensions
to provide both paraxial alignment and focus.
Click on the image for a larger version.

The APD itself is mounted on a small sub-board along with Q201 (the JFET) and the other capacitors noted in the box in Figure 5.  Most of Q201's drain current is provided by Q202's circuit, a current source, that operates at high impedance while Q203 is the rest of a cascode amplifier circuit that is designed to be self-biasing at DC and to provide gain mainly to AC signals.

The output of the cascode amplifier is passed to U201b, a unity gain follower amplifier.  This signal then passes to the circuit of U201a, a differentiator circuit that is designed to provide a 6dB/octave boost to higher frequencies to compensate for the similar R/C low-pass roll-off intrinsic to the APD and JFET itself:  Without this circuit, higher frequency audio components of speech would be excessively rolled off, reducing intelligibility.  By design the frequency range of the differentiator and its surrounding circuitry is intentionally limited so that low frequencies (below several hundred Hz) are strongly rolled off to prevent AC mains related hum from urban lighting from turning into a roar as are very high frequencies - above 5-7 kHz - which would otherwise become an ear-fatiguing "hiss" were the differentiation allowed to continue to frequencies much higher than this.

An interesting property of this circuit is that the "knee" related to this 6dB/octave roll-off occurs varies somewhat with the bias voltage and thus amount of device capacitance and, to a certain degree, its gain.  Because of this the frequency response of the APD/JFET circuit and the differentiator don't match under all operating conditions but experience has shown that it is better to have a bit of extra "treble boost" than not when it comes to making out words when the distant voice is immersed in a sea of noise.

A sample of the output from U201b, before differentiation, is also passed to J20, the "Flat" output.  The audio taken from this point, lacking differentiation, will sound a bit muffled under normal low-light conditions and it is not subject to either the high or low pass effects of the U201a differentiator which means that it will pass both subsonic and ultrasonic components as detected by the APD amplifier itself.  On the low end, the sensitivity is limited by 1/F noise which becomes increasingly dominant below a few 10s of Hz while on the high end it is again the capacitance associated with the APD and JFET circuits.  In testing it was observed that at this "Flat" output it was possible to detect signals from an LED modulated up to several MHz, albeit with significantly reduced sensitivity.  The main purpose of this output is to provide a signal point suitable for both subsonic digital communications as well as ultrasonic for experimentation with low/medium rate data, FM carriers and SSB signals.

In this circuit the amount of drain current in the JFET will vary depending on the individual properties of the JFET itself, the bias voltage, and the amount of impinging light.  Under "dark" conditions the "standing" JFET current was set to approximately 7-10 milliamps by the current source and the drain-source voltage varied from around 0.21 volts when the APD bias was just 12 volt to around 0.155 volts when the APD was operating at its maximum rating of 135 volts.  The specified JFET, the BF862, is typically capable of handling more drain current than this - and to do so would likely reduce its noise contribution slightly - but it was set at this level (with R205) to moderate battery current consumption.

Circuit testing:

Although it may have risked component damage, the APD circuit was "torture tested" to check ruggedness.  In a completely dark room a xenon photo flash was set off just inches/centimeters away from the photodiode with the bias set at 135 volts.  While the receiver was deafened for a second or two, the time it took for the various circuits to recover (e.g. power supply, re-equalization of various capacitors, etc.), repeated tests like this did not do any detectable damage to the receiver sensitivity or its noise properties indicating that the APD and JFET were more than rugged enough to handle any conceivable event that might happen in the field, aside from directly focusing the sun on the photodiode!

This circuit has also been successfully used in broad daylight.  While the receiver worked, the background thermal noise from the sunlit landscape was the limiting factor for sensitivity and the recovered audio had quite apparent nonlinearity (distortion) with an altered frequency response because the ambient light and resulting photodiode conductivity effectively shunted the high voltage bias and device capacitance.  In short, in such high ambient light conditions this circuit has no advantage over other optical receiver topologies such as the original "Version 3" or even a more conventional TIA (TransImpedance Amplifier) but its ability to be useful under such conditions is indicative of its versatility.

The results of in-field testing:

This receiver was first field-tested on a 95+ mile (154km) optical path during the September 2012 segment of the ARRL "10 GHz and up" contest:  For detail on this communication, read the blog entry "Throwing One's Voice 95 Miles on a Lightbeam" - link
Figure 7:
My end of the 95+ mile optical path during the session where the APD-
based optical receivers were first field-tested.  As seen in the picture
the optical path passes over urban lighting which tends to slightly raise
the noise floor due to both Rayleigh and lens-related scattering
Click on the image for a larger version.

During this test the optical (voice) link was first established using the "Version 3" PIN Photodiode receiver depicted in Figure 2.

With the reasonably clear air and the moderately long path we noted that we could reduce the LED current to a tiny fraction of the maximum before significant signal/noise degradation was noted.  At this lower LED current each station at opposite ends of the path switched from the PIN photodiode to the APD receivers and after tweaking our pointing and reducing the LED current even more we observed what turned out to be between 6 and 10 dB improvement in the signal-noise ratio - about what was observed on the indoor "Photon Range" with the initial prototype circuit.  It is likely that the actual improvement in sensitivity was greater than this, but because our respective optical paths passed directly over populated areas (see Figure 7) our ultimate noise floor was degraded by light pollution which included a thermal "hiss" and a low-level, harmonic-rich 120 Hz hum.

As was determined in the lab, the best signal-noise ratio in the field occurred with the APD biased in the 35-45 volt range where the "M" (amplification) factor was in the area of 3-10 (approximately 10-20dB gain).  At this rather modest bias voltage the "Gain+Noise" from the APD itself was sufficient to overcome much of the intrinsic noise of the JFET.  At higher voltages the gain continued to increase but the signal-noise ratio decreased at a faster rate until the APD's own avalanche noise drowned out the desired signal.

* * *

For more information about (speech bandwidth) free space optical communication, check out these links from my "Modulated Light" web site (link):

Be sure to check out the "" web site's other pages as well!


This page stolen from "". 

Wednesday, January 25, 2017

An A/B Battery replacement for the Zenith TransOceanic H-500 radio, with filament regulation

A friend recently gave me an old Zenith TransOceanic (ZTO) H-500 and after re-aligning it to get it into proper working condition I decided that I wanted a battery pack for it - both for "completeness" as part of making it look as original as possible and to allow the radio to be used outdoors, away from interference sources.  While it might be said that the GoogleWeb is lousy with options to replace the obsolete "A/B" battery used to power the Zenith TransOceanic, that wasn't a deterrent for me to design and build yet another one.

While it is easy to use a lot of 1.5 volt cells to get the 9 and 90 volts required to operate the radio, I decided to make do something different.

Figure 1:
The faux A400 "AB" battery, installed and working in the Zenith Trans
Oceanic H-500.  Contained therein are eight "D" type cells and circuitry
to produce the 90 volt "B" voltage and a regulated 9 volts for the
filament supply.
Click on the image for a larger version.
I threw a computer at it.

While it might seem odd to wield a microcontroller to solve a relatively simple problem on an antique, tube-type radio, it does make sense in a few ways as I'll outline below.

Design goals:

There are several things that I decided that this voltage converter should do:
  • Automatically power up when the radio is turned on and shut down when it is turned off. 
  • Cause no interference to radio reception.
  • Consume minimal current when the radio is turned off.
  • Produce a regulated B+ voltage so that radio performance is consistent.
  • Regulate the filament voltage so that the radio functions properly even when the battery is mostly discharged so that maximum use can be made of its total capacity.
While I was at it I decided that it should be able to do a few other things as well:
  • If the radio is on for a long time (e.g. more than about 2 hours) do a "power save" shut down to (hopefully) prevent the batteries from being completely flattened.
  • "Lock out" the operation of the radio if the batteries are already extremely low.  Avoidance of completely killing the battery may reduce the possibility of their leaking.

Generating the "B+" voltage:

The "B Battery" (high voltage) needs of the ZTO are rather modest:  pproximately 90 volts at 5-20 milliamps.  Aside from using a battery of sixty 1.5 volt cells or ten 9 volt batteries in series there are two common ways to generate this sort of voltage electronically:
  • Use a step-up transformer to take the low battery voltage to the appropriate B+ potential, typically using a low-voltage mains transformer in "reverse" (e.g. applying drive to the secondary, rectifying high voltage from the primary.)
  • The use of a simple boost-type converter using a single inductor.
The first method has the advantage that it is possible to design it such that the switching of the driving transistors is "slow" enough (at a modest efficiency loss) that it does not produce harmonics that may be picked up by the receiver - even at the lowest receive frequencies, and without shielding.  If you are interested in a good discussion of this method visit Ronald Dekker's excellent page on the subject (link).
Figure 2:
Test circuit to determine the suitability of various inductors and transistors
and to determine reasonable drive frequencies.  Diode "D" is a high-speed,
high-voltage diode, "R" can be two 10k 1 watt resistors in parallel and
"Q" is a power FET with suitably high voltage ratings (>=200 Volts)
and a gate turn-on threshold in the 2-3 volt range so that it is suitable
to be driven by 5 volt logic.  V+ is from a DC power supply that is
variable from at least 5 volts to 10 volts.  The square wave drive, from a
function generator, was set to output a 0-5 volt waveform to
make certain that the chosen FET could be properly driven by a 5 volt
logic-level signal from the PIC as evidenced by it not getting perceptibly
warm during operation.
The second method - and the one that I chose - uses a boost-type converter as depicted in Figure 2.  The switching frequency must be much higher than one would use with an ordinary mains transformer, typically in the 5-30 kHz range if one wishes to keep the inductance and physical size of that inductor reasonably small.  With these higher frequencies and the typically "square" drive signals (which are rich in harmonic content) needed to obtain good efficiency there is a much greater likelihood that it will interfere with reception - particularly in the AM broadcast band.  While a bit of a nuisance, the interference potential may be easily mitigated by putting the entire circuit in a metal box and appropriately bypassing and filtering the leads in and out.

Raiding my inductor drawer I picked a few "power" devices (those capable of handling at least half an amp) in the range of 100μH and 1 mH and threw together the circuit in Figure 2, consisting of a high-voltage FET (Q), the inductor under test (L), a high voltage, high speed diode (D), a 22μF, 160 volt capacitor (C) and a 5.6k, 2 watt load resistor (R).  Connecting the FET's gate to the square wave (50% duty cycle) TTL-level output of a signal generator I measured each one in terms of output voltage, total output power and overall power conversion efficiency with respect to frequency.

As would be dictated by the plethora of design articles on the subject, not to mention data sheets of switching regulator chips, I noted that neither the value of the inductance or switching frequency was particularly critical to achieve the desired results.  In general, higher inductances produce a bit more output at the lower frequencies (a few kHz) while the lower inductances worked a bit better in the 10-30 kHz range, but all of the inductors did work over the entire range to a greater or lesser degree.  Settling on a decent-sized 330μH inductor - a value that is not particularly critical - I proceeded with the circuit design.
Figure 3:
Schematic diagram of the voltage converter.  See text for details.
Click on the image for a larger version.
The circuit:

Rather than go through a lot of theory I'll just describe the circuit that I designed and built - See Figure 3, above.

When the radio's power switch is turned on its filament circuit is connected and a voltage appears across the "Batt-" and "A-" leads and R7, a 10k resistor connected across switched-off FET Q4 which are in series with the filaments.  When this happens transistor Q3 is turned on, pulling the base of Q1, a PNP transistor in the high side of the BATT+ line, toward ground and turning it on and applying power to U3, a 78L05 voltage regulator, and microcontroller U1, a PIC12F683.  After a short initialization delay the microcontroller activates the "PWR_SW" line, turning on Q2 which assures that Q1 is always turned on even if the filament switch is turned off abruptly and Q3 turns off or, as we shall see, when the battery voltage is at or below the filament regulator's set point.

At this point the microcontroller enables interrupt-driven code to produce the high voltage (B+) output by monitoring it via resistor divider R18/R19/R20:  If the voltage is below the threshold, the duty cycle of the PWM signal output on the "SW_DRIVE" line is increased to force more energy storage in the inductor (L1) - up to a maximum limit of around 80%, set in software.  If the voltage is above the threshold, the duty cycle is decreased - down to zero and even into "discontinuous" mode (e.g. the PWM signal intermittently turned off and on) if necessary as would be the case if there were no load on the output.  In this way the output voltage is appropriately regulated, typically to 90 volts as set by R19.  In this circuit when the PWM signal turns off Q5, the high voltage FET, the magnetic field in L1 collapses and induces a high voltage across it.  The current resulting from this field collapse is rectified by high-speed, high-voltage diode D2 and stored and filtered by C8 and additionally filtered and smoothed by R21 and C9.
Figure 3:
The (mostly complete) converter board.  The high-voltage FET (Q5) is
in the lower left corner while the filament regulator FET is in the lower-
right corner.  In the upper right corner is U2, the rail-to-rail dual op-amp
that is part of the filament regulator.  Because of the very small amount of
heat being dissipated by any component, no heat sinks were required.
The high voltage filtering components and the optoisolator are in the
upper left corner.
No circuit board is available - but if you design one, I'd be happy
to post information about it and give you credit! 
Click on the image for a larger version.

Because the battery voltage could be as high as 16 volts if ten fresh "1.5" volt cells were used it is necessary to regulate the filament voltage down to something around 8.5-9 volts.  Op amp section U2b is configured as a "difference amplifier" (a.k.a. subractor) that measures the voltage difference between the "A-" and the "A+" lines (the filament supply to the radio) and this calculated voltage difference is output from U2b and applied to the inverting input of U2a via scaling potentiometer R14.  The voltage at the inverting input of U2a as set by this potentiometer is compared to the "reference" voltage applied to its non-inverting input and if the voltage is low, its output voltage is increased so that FET Q4, which is placed between the A- and BATT- connections, conducts more to increase the filament voltage.  Conversely, if the voltage is too high, the output voltage of U2a to Q14's gate is reduced, decreasing its conductivity.

The use of the circuitry of U2b is necessary because neither the A- or A+ (filament) leads are referenced to the circuit ground (e.g. they are sort of "floating") which makes it necessary to measure the difference between those two leads to ascertain the actual filament voltage.  If the battery voltage does get low enough that Q4 is completely "on", the voltage across R7 will disappear and Q3 will turn off:  It is because this can happen that we must have activated Q2 to keep the microcontroller's power turned on and this is also why we cannot use the voltage drop that we used to tell if the radio was turned on to also detect if the filament current has ceased to flow when the radio is turned off.

Note:  It would have been possible to have used a microcontroller to regulate the filament voltage in a manner similar to that in which the high voltage is produced, but a programming bug or crash could cause the fragile, expensive tubes to be exposed to the full battery voltage whereas a malfunction of the high voltage generator is unlikely to cause damage to the radio.

A short time after the high voltage converter is enabled the "FIL_SW" line is set high.  Because the microcontroller has low-impedance FET output drivers, this pin's voltage is essentially that of the 5 volt regulator and it is used as the filament voltage reference.  Similarly, if the microcontroller sets the "FIL_SW" line low (zero volts) this will shut off the filament supply.

With the use of a MOSFET (e.g. Q4) as the filament control device, the series regulation of the filament has a very low drop-out voltage - that of the voltage drop across the FET, limited by its own "on" resistance, and the wiring used to carry the filament current - and this drop can be as low a few 10s of millivolts.  What this means is that if the filament voltage is set to 9.0 volts by R14, as long as the "A" battery voltage exceeds that by a few 10s of millivolts, the filament will always be maintained exactly 9.0 volts but if the "A" supply (battery voltage) drops below 9.0 volts, Q4 will be turned fully on and the filament voltage will be within 10-20 millivolts of that battery voltage.  Compared with the operation of a typical "low dropout" regulator IC that has around 0.15-0.3 volts drop, the circuit used here offers a lower voltage drop and better radio performance in those situations, particularly when even a few tenths of a volt can make a lot of difference!
Figure 4:
Inside the completed voltage converter.  All leads going in and out are
bypassed with low-ESR electrolytic capacitors and further filtered with
series chokes as shown in Figure 3.  The use of a completely shielded
enclosure (top not shown) is necessary as direct E-field radiation from the
circuit will otherwise be heard on the radio.  This box is made from
cut pieces of circuit board material, soldered at the seams inside and out,
with cut-in-half nickel-plated brass standoffs soldered to the board being
used to support the circuit and at the corners to attach the lid.
Click on the image for a larger version.

A second or so after the application of the filament voltage - enough time for the tubes to warm up - the microcontroller starts to "look" at the current drawn on the B+ lead as detected by U4, an opto-isolator that is in series with this supply.  Once the tubes warm up and begin drawing current, U4's internal LED turns on, activating its internal phototransistor which then pulls the "HV_IMON" (high voltage current monitor) line low, indicating to the microcontroller that the radio is now operating.  At this point the microcontroller is in a mode where it will repeatedly check to see that current is drawn by the radio on the high voltage line.

When the radio is turned off the current on the B+ line will disappear due the loss of the tubes' emission caused by the filaments being turned off and, possibly, the B+ line being disconnected.  When this happens the LED in optoisolator U4 will turn off, its phototransistor will stop conducting, and the "HV_IMON" line will be pulled high indicating to the microcontroller that the radio has been turned off.  After a short "debounce" period to verify that this loss of current wasn't due erroneously detected, the microcontroller will shut off the high voltage generator, set the "FIL_SW" line low, powering down the filament regulator, and then set the "PWR_SW" line low which then disconnects the microcontroller's power source from the BATT+ line, removing load from the battery.

Why use eight 1.5 volt cells rather than just six to get the filament voltage?

Why not just use six 1.5 volt cells to get "9 volts" for the filament string?  As it turns out only a set of six fresh 1.5 volt cells will actually produce 9 volts - and the voltage drops from there.  If one consults the manufacturers' specifications for alkaline cells it will be noted that the majority of the useful life of typical "1.5 volt" cells occurs with their voltage being in the range of 1.2-1.3 volts and it isn't until a cell gets all of the way down to 1 volt (for a total of 6 volts to the filaments for a six cell battery) that just 80% of the cell's capacity has been exhausted.

In this radio I noted that below an "A" battery potential of 8 volts (e.g. 1.33 volts/cell for 6 cells) the sensitivity started to drop and by the time it has dropped to around 7.5 volts (1.25 volts/cell for 6 cells) the radio was practically deaf with the oscillator abruptly stopping just below this.  Poking around inside the radio I noticed that at 9 volts, the series voltage drop across each of the tubes' filaments was very close to that shown on the schematic diagram in the service manual, but by the time it dropped to 7.5 volts it had become unequal, with the 1L6 converter tube being disproportionately affected and its filament voltage at or below 1 volt.  Interestingly this drop-off in sensitivity did not appear to be related to frequency:  The radio still worked at all frequencies with a filament voltage just above where it cut off, but it was just as deaf on the low bands as it was on the high.  Because the 1L6 tube is the component in this radio that is the most difficult to find, it would also make sense to construct the battery supply in such a way that it would allow the best operation from a "weak" tube, anyway.

For this reason I decided to use the battery voltage of eight 1.5 volt cells rather than the "9 volts" obtained from six cells for two important reasons:
Figure 5:
Inside the faux "AB" battery box for the Zenith TransOceanic.  Eight
"D" cells are used in four holders (one 4-cell,  one 2-cell and two 1-cell) which,
along with the converter box, are screwed down to some plywood (3 layers of
3.2 mm "luon") which itself is glued to the bottom of the box.  The cover,
made from the same circuit board material as the box containing the circuits,
has both of its surfaces electrically connected using thin, copper foil soldered
to each side to assure that an electrical connection is made to the box
itself when the cover screws are tightened.  The authentic-looking replica
battery box and radio connector were obtained from "".
Without having made the voltage converter smaller, there is room only for
eight "D" cells in the box.
Click on the image for a larger version.
  • The higher voltage of eight 1.5 volt cells (12+ volts when fresh) would allow the total filament potential ("A" voltage) to be regulated down to 8.0-9.0 volts.  (For longest filament tube life, one should run the filament string in the 8.0-8.5 volt range - the lower end being somewhat preferred if the radio's performance is still acceptable.)
  • The use of an extra two cells will allow the use of more of the battery capacity.  For example, with 8 cells discharged to 1.0 volts each, around 80% of the cell's useful life has been utilized with the ending voltage still being 8 volts.  Contrasting this to the use of just six cells, at a total "A" voltage of just 7.75 volts (approx. 1.3 volts/cell for 6 cells) 40-60% of the life of the cells will remain, but the radio will likely be getting deaf or even may not be usefully operational!
  • In theory, ten 1.5 volt cells could be used.  Because the voltage of a "fresh" 1.5 volt alkaline is around 1.6 volts, this could expose some of the devices, particularly the electrolytic capacitors and U2, to voltage at or above the official maximum rating.  Practically speaking these devices will likely survive this, particularly since the voltage will very quickly drop under the load presented by the radio into the "safe" range.  The use of one or two additional 1.5 volt cells (e.g. 9 or 10) won't add more than 10-15% of "run time" to the radio so it is not likely to be worth using more than eight 1.5 volt cells.  (Nine cells would work as well, provided there was space in the battery box and that you were OK with using an odd number of cells.)
  • The typical filament current of this radio is on the order of 50 milliamps.  At a battery voltage of 12 volts where 3 volts is dropped by the series regulator, approximately 150 milliwatts is dissipated as heat - about 25% of the total filament power, or around 8% of the radio's total power consumption.  Were a switching regulator used for the filament its efficiency would likely be in the 85-90% range and increase of efficiency over the linear regulator would likely not be worth the added complexity.  Considering that the average voltage of the battery over its life will be around 10-10.4 volts (approx. 1.25-1.3 volts/cell) with a regulator dissipation of only 70 milliwatts, the difference in loss will be even lower.
With a fresh set of eight 1.5 volt "D" cells the current consumption was measured at 140-150 milliamps at very low volume and peaking to well over 250 milliamps when the volume was set to maximum on a strong station (lots of audio distortion!) with the filaments accounting for around 50 milliamps of the total.  While it has not been empirically tested (it's not particularly cheap to buy eight "D" alkaline cells just to run them down!) the estimated run times at "room" temperatures and normal receive volume to 1 volt per cell for various sizes of alkaline cells, based on manufacturers' data sheets are:
  • For "AA" size:  15-20 hours with reduced performance for an additional 1-2 hours.
  • For "C" size:  30-40 hours with reduced performance for an additional 3-5 hours.
  • For "D" size:  70-90 hours with reduced performance for an additional 6-10 hours.
If just six cells were used the filament voltage would drop below 7.5 volts in about half the time noted above and by then, the radio's performance will have likely diminished considerably.  In contrast, using eight cells and a filament voltage regulator the performance will remain essentially unchanged until the cells are about 80% discharged (around 1 volt/cell) and the radio's performance will drop from there.

Note that this circuit can be powered directly from a 12 volt supply or battery - just heed the warnings below about NEVER allowing the "Batt-" line to come in contact with the "A-" lead - or any part of the radio's chassis.  Because this radio has no AC (mains) power transformer, its chassis could be "hot" with mains voltage and as such it is already capacitor-isolated.

Additional comments about the circuit:

It should be noted that the "BATT-" and "A-" lines are isolated from each other.  These two lines should never be connected to each other as that would prevent the closure of the filament switch from being detected when the radio is turned on and it would bypass the filament regulator, exposing the tubes' filaments to the full battery voltage, likely destroying one or more of them!  The reason for putting the filament regulation in the negative lead is allow an N-channel FET to be used and to avoid the use of a P-channel device in the "high" side and the complications required in driving this device and keeping its circuit stable (e.g. avoiding spurious turn-on events and momentary loss of voltage regulation) when the unit is powering up or down.

Even more circuit comments:
  • Resistors R8 and R17 are used to bias their respected FETs "off" by default.  This is necessary as the outputs of the microcontroller are high-Z unless/until it is operating and these FETs could randomly turn on due to leakage currents without them.
  • Similarly R15, on the "reference" voltage for U2's filament regulator circuit from the microcontroller, pulls that output down before the processor initializes its outputs from their default "Hi-Z" state, eliminating a possible "glitch" of the filament voltage during circuit start-up and shut-down.
  • U2, the filament voltage regulator, MUST be a rail-to-rail input and output op amp.  An "ordinary" op amp such as the '1458 or '358 WILL NOT WORK PROPERLY under all conditions.  Some parts suggestions for suitable op amps are included in the schematic diagram of Figure 3.  In other words, if you use a "normal" op amp it is possible that this circuit will misbehave and expose the filaments to excessive voltage.
  • Resistor R9, a 470 ohm resistor in series with the output of U2a and FET Q4, isolates Q4's gate capacitance, preventing instability of the op-amp while C6 provides frequency compensation for the regulator circuit.
  • When powered down the quiescent current of this circuit is approximately 7μA and is a result of the battery voltage (minus the drop of D1) always being applied across the B+ voltage divider string R18, R19 and R20.  This amount of current is comparable to the self-discharge rate of modern alkaline cells and can generally be ignored.  If this amount of current were to really bother you, the  voltage converter circuit could be powered from the "V+_SW" line and transistor Q1 could be replaced with a P-channel power FET as noted on the diagram.
  • LED1 and LED2 are optional.  LED1 will glow when the microcontroller activates the "PWR_SW" line and can be used for troubleshooting.  For example, if no current is being drawn from the B+ line - or the converter is not working - the software will continually cycle:  It will turn on the high voltage, wait for current to flow and when not seeing it, it will turn off the high voltage again and retry after a few seconds causing LED1 to turn on and off.  LED2 is also optional and is used to indicate when the circuit is powered up by Q1, either by the microcontroller turning on Q3 and/or Q2 being activated by the voltage drop across R7 when the filament switch is closed.  If desired, it may be mounted to the battery box so that it is visible when radio's back cover is open.
  • Transistor Q5, used in the high voltage "boost" converter, must be rated for at least 200 volts and it should have a "logic level" gate threshold appropriate for turning the FET (more or less) fully on at just 5 volts:  Some suggested device types are noted on the diagram (Figure 3).  An additional device worth considering is the ON Semiconductor NDD02N40-1G, a 400 volt, 1.1 amp FET that has a suitably low turn-on threshold - and it's pretty cheap.
  • Components TH1, a 1 amp self-resetting fuse, and diode D1 protect the circuit against shorts or accidental reverse polarity by limiting the current to a reasonable value should this occur.  TH1 may be replaced with a 0.75-1 amp fast-acting fuse if so desired.
  • The PWM (switching) frequency is approximately 15.625 kHz and is based on the microcontroller's internal 8 MHz clock.  Both 7.8125 and 31.25 kHz were tried and the conversion efficiency was slightly lower (by approx. 1-5%) with the 330 μH inductor value chosen - an indication that the actual value of L1 isn't particularly critical.
  • The value of L1 may be anything from 220μH to 470μH - and even a bit beyond this range.  Make sure that the inductor used has a current rating of at least a half an amp or else internal resistive losses will significantly impact conversion efficiency.  If available, a toroidal inductor or other shielded type is preferred as it better-contains its magnetic field than solenoid styles.
  • The measured efficiency of the boost converter of the prototype was greater than 80% despite the power lost in R21, the "filter" resistor in series with the B+ output.
  • The 15 volt maximum supply voltage limit is set by the voltage rating of op amp U2 and possibly the ratings of the electrolytic capacitors exposed to the battery voltage.
  • If one chose to use just six 1.5 volt cells instead of eight, never supplying more than 9 volts, the "FIL_SW" line would be connected directly to the gate of Q4 and the circuitry related to U2 would be omitted.  Do note that six "fresh" 1.5 volt alkaline cells could initially produce a bit over 1.6 volts/cell and expose the filament string to over 9.6 volts.
  • The diagram and pictures show the use of feedthrough capacitors (4000pF) to pass the voltages through the shielded box.  Feedthrough capacitors are somewhat difficult to get, but good results may be obtained by using good-quality monolithic ceramic (NOT disk ceramic) capacitors instead, placing them - using very short leads to a solid ground plane - where the wires pass through the hole in the shielded box.  These capacitors are typically square in shape and rather compact and available in both leaded and surface-mount form.  Remember that for the B+ output a capacitor with a rating of at least 100 volts must be used:  Any value from 0.0022μF to 0.1μF may be used.
  • If you build this sort of circuit make absolutely certain that you simulate the filament string with a 150-200 ohm 1/2-1 watt resistor and the B+ load with a 10k, 1-2 watt resistor and verify that the circuits are working properly BEFORE connecting it to a radio.  While a brief bit of over-voltage on the B+ line (to perhaps 130-150 volts) will likely not harm the radio, more than 9 volts on the filament line, even for a moment, will probably ruin one or more of the fragile and expensive tubes!
  • About that "auto power save" feature mentioned at the top of this article?  After two hours of uninterrupted operation the microcontroller will modulate the filament line with an intermittent tone and drop the B+ voltage to about 50% causing the radio to partially mute with the alarm tone sounding in the speaker.  This "beeping" will continue for about a minute before the microcontroller turns off the filament and high voltage supplies, dropping the current consumption from around 150 milliamps to about 6-12 milliamps - the quiescent current of the remaining circuitry.  Turning the radio off for 5-10 seconds and then back on will reset this timer at any time.  The down-side of this is that if the radio shuts down in this way, one may forget that the radio is even on, still drawing a few milliamps, except for the fact that the front lid of the radio will still have been in its upright position!  If the battery voltage is less than around 7.5 volts (0.9375 volts/cell) the radio will be "locked out" and will not even turn on, but at this voltage the batteries are not only quite discharged, but their internal resistance will be rapidly increasing as well and little run time would have been left.
Figure 6:
A handy "map" showing where the various RF adjustments may be found.
This doesn't really have too much to do with the article, but since I made it
when I was aligning the radio I thought that I might as well post it here!
Note that locations of some of the trimmer capacitors - particularly those
in the lower-left corner - will vary with different production runs.  Some of
the alignment points shown in this picture are also omitted in the
"official" H500 service manual and thus have no parts designations:  These
adjustments are peaked at the frequencies indicated on the drawing.
Click on the image for a larger version.
How well does it work?

As can be seen in Figure 1 the circuit board and the eight "D" cell battery is concealed in a replica battery box that is situated exactly where an original "AB" battery would be placed.  Then the power switch is turned on it takes a bit over a second for the computer to power up, do its checks and for the tubes to warm up and the radio begins playing while the power-off is detected within two seconds of the radio power switch being turned off.

With the shielding of the circuity and bypassing of its leads there is no detectable interference caused by switching voltage converter.  With the filament and B+ voltage being regulated to the same as a "fresh battery" or AC mains voltage, the sensitivity and audio output capability are maintained until the battery is more than 80% depleted.

In other words, it works just as it should!

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If you are interested in the code for this (written in "C" using the PICC compiler) or just a .HEX file so that you can program a PIC12F683 yourself, or if you are interested in getting an already-programmed PIC12F683, let me know via a comment.

And before you ask:  Sorry, but I can't build you one at this time...


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