Setting/calibrating signal and noise levels for signal/noise level monitoring using the "wsprdaemon" script

The WSPRDaemon script, a work largely of Rob, AI6VN, is designed to automate WSPRNet reporting of signals transmitted using the WSPR protocol - (The script may be found on GitHub, here).  While the WSJT-X program, itself, can report the signals that it hears in terms of callsign, Maidenhead grid locator and apparent signal-noise ratio, the WSPRDaemon script leverages the multi-receiver capability that can exist in various hardware configurations - including the KiwiSDR - which, with the appropriate platform, can receive as many as 15 signals on 15 different frequencies simultaneously.

Comments:

While this discussion is focused on methods and measurements exclusively using the KiwiSDR as that is the hardware with which I am most familiar, where similar capabilities exist, these same methods can be applied to other receive platforms.

Similarly, the methods discussed on this page need not be specific to the software mentioned and may be applied any time receiver signal level measurement is needed.

Care has been taken such that at the Northern Utah WebSDR, all of the WebSDR's receivers' S-meters are calibrated to within a dB or so in the center of their target band using the methods described below.

HF noise floor measurements:

In addition to facilitating the decoding and reporting of WSPR signals, the WSPRDaemon script can also divine the apparent noise floor within the receive bandwidth on some of these platforms.  In the case of the KiwiSDR - where an "S-Meter" reading is available via the remote interface - this reading can be used to derive absolute measurements and such data, taken from multiple locations over disparate geographical locations, can be used to glean information about the noise on the HF bands over time.

In practice, accurate, absolute signal level readings are quite difficult to obtain owing to the issue of meaningful calibration - and the more complicated the antenna system is, the more difficult it may be to get such measurements as one must consider things like:

• Cabling losses
• Splitter losses
• Filter losses
• Amplification
• Calibration of the receiver
• Gain of the antenna (e.g. dBi, dBd)

Fortunately, all but the last on the list may be accounted with a single measurement using a known-accurate signal source.

The "short" version:

The ultimate goal of this exercise is quite simple:

Input a signal of known amplitude into your antenna system - say, -60 dBm - and have the noise logger report that same signal level, +/- your known antenna gain.

The steps below are included in the event you may not be familiar with the methods/techniques involved.

An example receive system:

As an example, the KiwiSDR-based portion of the system in use at the Northern Utah WebSDR (a browser-enabled remote HF receiver found at sdrutah.org) will be discussed and a simplified block diagram of the signal path may be seen below:

Figure 1:
Simplified KiwiSDR Signal Path used at the Northern Utah WebSDR showing several gain blocks and filters.
As noted, the directional coupler is connected in "reverse" as it is being used to inject rather than extract a signal from the signal path.
Click on the image for a larger version.

The components of the receive system depicted above in detail:

As with any antenna system, it begins with an antenna - but for the moment, we will ignore any intrinsic gain - or loss - that the antenna may have.

• Directional coupler:  At the Northern Utah WebSDR, the Mini-Circuit Labs ZFDC-20-3 is used at the main antenna input.  This device has minimal insertion loss (<0.25dB) on the "through" line and there is a separate port (labeled "CPL") that is coupled at 20dB from the through line.  This sort of device is typically used to extract a small sample present on the line while minimally affecting it and "in" and "out" ports are labeled for such purposes, but here it is being used in reverse to inject a signal into the receive signal path.

This device is convenient in that it has the advantage that it may be left in line all of the time with negligible effects at HF and it can put test signals into the receive signal path without affecting off-air performance.

This device is optional in that the same measurements may be obtained without it by connecting the signal source directly to the antenna port - the obvious disadvantage being that the signal path is interrupted - something that we try to avoid on a busy, multi-user system such as a WebSDR.

• Calibrated signal source:   It should be no surprise that having a known-accurate signal source is an absolute necessity when it comes to calibrating the signal level readings with available levels in the range of -60 to -30dBm being the most useful.  Having access to such devices (e.g. Communications Test Set or RF Service Monitor) is helpful - but other means of obtaining a usefully-accurate signal include:

• An "inexpensive" calibrated signal source like the Elecraft XG-3 - the >$300 price tag (when fully equipped) being the reason for the quotes.

• A "transfer" measurement from a signal source with a consistent level.  Some antenna analyzers - including the popular "NanoVNA" - can output a level that is fairly consistent.  Coupled with outboard attenuators (a switchable step attenuator and/or an assortment of fixed attenuators) one can use known-accurate test equipment (an RF power meter, signal level meter, spectrum analyzer) to obtain a table of signal power readings for each of the HF amateur bands.

• Because of the harmonic content of the NanoVNA, it is preferred that a selective means of measuring RF power like a spectrum analyzer be used rather than radiometric detector like a power meter - but the difference is unlikely to be significant.

• Important note:  The amplitude of the NanoVNA's output port is too high to be directly used as a calibration signal - especially if amplification is used in the signal path.  Expect to need attenuation in the range of 20 to 50 dB to provide a usable signal level:  It's recommended that one use a switchable step attenuator and/or a series of fixed attenuators to get reasonable signal levels.

• A transfer measurement from a piece of equipment that is likely to be "pretty close" - such as the KiwiSDR itself:  With the default "factory" S-meter calibration level of -13dB the KiwiSDR to which I have access have typically been within a couple of dB when comparisons were made.
• For calibrating, a signal source (e.g. NanoVNA) would be connected directly to the input of the receiver - via the attenuator(s) - and the S-meter levels noted for different settings on each of the amateur bands.

• AM BCB Filtering:  In many cases, strong AM broadcast band (mediumwave) signals will be present on the antenna port.  Even if these signals are not strong enough to overload the RF amplifier that follows, such signals can overload the receiving device downstream.  As mentioned later, there are advantages to strategically attenuating strong signals to minimize the total power impinging on the A/D converter of any direct-sampling receiver.
  

The AM broadcast-band filter used at the Northern Utah WebSDR is a rather complicated affair (and is described in this article:  Managing HF Signal Dyanmics and preventing overload... (link)).  Its main purpose is to reduce the amplitude of several very strong (50kW) local transmitters to prevent the "wasting of amplifier power and A/D bits" on very few signals.

• Amplification (post AM/BCB filtering):  In general, a 15 dB system noise figure is sufficient to allow the detection of a 10 meter signal above even the lowest expected level of background noise found at this frequency.  For this reason it is suggested that an RF amplifier be placed after the AM BCB filter (to minimize the probability of overload and intermodulation/mixing products).  Remember:  Once loss appears in a receive system due to loss/attenuation, no amount of amplification can restore the system noise figure beyond that point.

Such an amplifier should also be placed prior to any RF splitting as much as is practical as any loss - which can be from the feedline, in filters or in a splitter - will directly contribute to the the system noise figure.  In cases where very strong signals may be present on the antenna system (local AM BCB, strong Shortwave Broadcast signals - when the bands open) it is preferable to have an amplifier at this point that has only modest gain - in the area of 12-15 dB:  A high-gain amplifier (>20dB) may be more prone to overload in such conditions and the typical means of preventing this (addition of an attenuator in front of the amplifier) is not recommended as this will increase system noise figure.

• Limited attenuation high-pass filter:  Any receive system is capable of handling only a certain amount of total signal input power - and the direct-sampling receiver of the KiwiSDR is no exception.  The overload level of the KiwiSDR's 14 bit A/D converter is approximately -13dBm, so the total power arriving at its antenna port should be kept below that level and strategic, selective attenuation of local, strong carriers - often those of AM/mediumwave broadcast transmitters - should be considered - but there is another problem:

The sensitivity of a KiwiSDR is approximately -155dBm (in a 1 Hz bandwidth) over its frequency range which means that in a unity-gain HF antenna system that at 10 meters, the KiwiSDR will be "short" by at least 6-8dB from hearing the textbook "rural quiet" noise floor on that band, requiring 10-15 dB of overall amplification to place the 10 meter noise floor comfortably above that of the receiver's in that environment.  This amplifications has a cost:  With the addition of a 12 dB RF amplifier, the KiwiSDR will now overload at -25dBm.  It is not uncommon for the absolute power level of a single high-power shortwave broadcast station to exceed this level under good band conditions - and there will be many such signals on the bands - not including the wide-band energy from lightning static - that can combine in overall signal power and exceed this overload level.

What system noise figure is required for HF reception? This is a tricky question to answer because conditions on the HF band vary so much - most strongly on a per-band basis. Assuming quiet (e.g. "dead"), bands the lack of storm static. the absence of solar noise and the receiver being located in an area completely absent of man-made noise, noise figures equal to or lower should be attained for the receive system to have a "natural noise" limit: 1.8 MHz:  45dB NF 3.5 MHz:  37dB NF 7 MHz:  30dB NF 14 MHz:  24dB NF 21 MHz:  20dB NF 28 MHz:  15dB NF 50 MHz:  9dB NF 144 MHz:  2dB NF These numbers are approximate, but are a general guideline for system design.

Were the KiwiSDR being used over a very narrow range of frequencies - such as the case for most commercial direct-sampling receivers like the Icom IC-7300 and IC-7610 - this could be managed by gain adjustment in the signal path, but if the KiwiSDR is to be used over the entire HF spectrum, this cannot readily be done.  Instead, one can selectively insert attenuation where the signals+noise are likely to be the highest, namely below approximately 10 MHz and adjust calibration accordingly.

The use of this "limited attenuation" high-pass filter leaves signals above 12 MHz alone but offers significant attenuation (about 12 dB - around 2 "S" units) to signals below 8 MHz.  Because of the relatively high noise levels in the lower HF bands, additional attenuation - and the commensurate increase in system noise figure - can be tolerated without the loss of useful sensitivity.

Note:  This same issue of overload will occur with similar devices - such as the Red Pitaya - if operated as wideband HF receivers, without input band-pass filtering.  Simple math will indicate that even if the KiwiSDR had a 16 bit A/D converter - which would theoretically yield another 12 dB of dynamic range - one would still need to take these same precautions.  It is for this reason why all well-designed direct-sampling receive gear has narrowband filtering that precedes its A/D converter.

Unfortunately, the "limited attenuation" filter fits a niche requirement as is not a commercially-available device, but one may be built without too much difficulty:  Two versions are described here:

• A limited-attenuation high-pass filter for the KiwiSDR - link. This filter starts to attenuate below 12 MHz, offering about 12 dB below 8 MHz.

• Revisiting the limited-attenuation high-pass filter - link.  This version is similar to that above, but its attenuation pass range is limited  to the range of approximately 1700 kHz to 12 MHz, allowing the passage of signals in the AM broadcast band and below for the reasons discussed in the article.

• Because of the simplicity of these filters they exhibit a low return loss (high reflectance) at lower frequencies and  it is recommended that they be preceded with an amplifier to present the input feedline with a more consistent "50 ohm" impedance.

• Amplification (post high-pass filter):  Where high signal levels are likely to exist it is prudent to distribute the signal gain across several blocks rather than have a single high-gain amplifier near the "front" of the signal path.  An amplifier placed after the "limited attenuation" high-pass filter will see lower signal levels from the very strong lower HF-band signals and be less subject to overload.

• Splitter:  A 4-way splitter is depicted in figure 1.  A real-world 4-way splitter will have 6.5-7.5 dB loss between the input and any of its output ports which means that for the higher-frequency bands it is particularly important to have established the system gain and noise figure prior to the splitter.  Like any other block of gain or loss, the impact of such a device must be considered when evaluating the system overall.

Analyzing your receive system:

The procedure following this point assumes the availability of a known-accurate signal source:  See the heading "Calibrated Signal Source", above, for more information.

Again, we are characterizing the signal path beyond the antenna and ignoring any gain/loss of the antenna itself for the moment.

A procedure for obtaining the calibration of the receive signal path is as follows:

• Input a known signal level at the input of the receive signal path.  This signal would be input at the antenna port - or via the "coupling" port of the directional coupler.  This signal should be input at a frequency in the band of interest as calibration values should be obtained for every amateur band to be used.  For WSPR operation it is desirable that one generates test signals within the WSPR passband.

• Tune in the signal and note the signal level on the receiver.  A known signal level of at least 20dB (30dB is better) above the receiver noise floor and other signals in the receiver passband is suggested to minimize effects of the noise and signals - but the level should be no higher than necessary to avoid overload/compression of any amplifiers - or the receiver itself.

• If using a directional coupler as suggested, make sure that it is oriented correctly by comparing the signal levels when the generator output is fed directly into the receive signal path and again with signals fed via the coupler:  If its "in/out" is reversed the coupling level will be inaccurate. (Again, for a coupler used for extracting a signal from a coax, the connection will be reversed.)

• In the case of the 20 dB coupler suggested above, the signal levels reaching the receiver will be 20 dB lower due to the coupling loss than that if the signal generator coupled directly and this should be remembered when taking measurements.

• If the levels of the test signal are low (less than 20dB S/N) and cannot be increased, narrowing of the receive passband can minimize the effects of noise - just observe the S-meter while making such adjustments to verify that the test signal remains within the passband.

• Because we are measuring the amplitude of a coherent signal within a finite bandwidth rather than trying to measure the noise power in a specific bandwidth, we must make sure our test signal is strong enough that the power of the background noise is a small percentage of the total.

• While it is necessary to compensate for detection bandwidth when measuring the noise within that bandwidth, one does not do so for coherent signals as all of the power of that signal is (theoretically) all on a single frequency.  It is for this reason that the measurement obtained of the coherent signal should match the actual power of that signal.

• Knowing the signal level being input to the signal path, observe the S-meter reading on the KiwiSDR on a per-band basis.  It is not important that the S-meter read correctly - except for the benefit of users connected to the KiwiSDR and looking at the meter - as the levels reported by wsprdaemon will be adjusted in its calibration file.

• If you are running wsprdaemon and are producing a signal within the WSPR passband, the signal will show up on the graphs after several minutes as a constant level - and this level can be used to verify calibration.

• The level displayed on the graph will be that of the KiwiSDR plus the amount of attenuation/amplification specified in the wsprdaemon.conf file for that band.  Use either raw data or the interactive Grafana data to read the signal levels precisely.

• At the Northern Utah WebSDR, the KiwiSDR S-meter calibration setting has been adjusted so that for bands above the roll-off of the high-pass filter (e.g. 20-10 meters) the S-meter read the same as the input signal level.  At lower HF bands, the S-meter reads low by the amount of attenuation of the filter.  Since there is currently only one S-meter calibration point that applies to all frequencies, there is nothing to be done to correct this.

Any adjustment of the KiwiSDR's S-meter calibration should be done before completing the steps below.  If it is changed, the final per-band calibration values will need to be adjusted accordingly.

• Record, for each band, the amount of RF you are applying at the antenna port (taking into account the coupler, if used) and the S-meter reading on the KiwiSDR.

• The difference between these two readings will be the amount of gain - or loss - in your signal path.  Again, we are not concerned about the absolute gain of the antenna itself at this point.

• If the reading on the KiwiSDR is above that of the absolute input level, signal gain is indicated at that frequency.

• If the reading on the KiwiSDR is below that of the absolute input level, signal loss is indicated at that frequency.

• We have now characterized the gain/loss of the signal path.  If our antenna had unity (0 dB) gain we would use the numbers that we'd just obtained as an offset in the wsprdaemon.conf file.  For example:

• If -50dBm were present at the input of the signal path and the KiwiSDR read -38dBm on 10 meters, we would know that we had 12 dB gain at that frequency.  To compensate we would put -12 db in the entry for 10 meters in the wsprdaemon.conf file.

• If -50dBm were present at the input of the signal path and the KiwiSDR read -52 dBm on 80 meters we would know that we had 2dB attenuation at that frequency.  To compensate we would put 2 dB in the entry for 80 meters in the wsprdaemon.conf file.

• Again, to verify our calculations we can place a carrier of known signal level within the WSPR passband for the band being tested.  If the measurements have been done correctly the levels on the graph should match those of the level of the RF at the input of the signal path.

Compensating for coax cable losses:

The next step would be to include the losses of any coaxial cable in the system before the point in the signal path where we injected our test signal.  This can be directly measured with a wattmeter and dummy load being used to measure the power lost over its length or - presuming a fairly low VSWR - can be estimated using the manufacturer's loss values on a per-band basis.

These loss values would be added to the per-band numbers in the wsprdaemon.conf file.

Compensating for antenna gain:

With simple equipment, calibrating the signal path for an absolute level is pretty easy - but the gain of the receive antenna at a given frequency is more difficult to determine as very few people have access to a known-accurate antenna range - so we will have to make an educated guess.  Because many HF antennas have definite patterns of of lobes and nulls one will have to decide if the peak gain of the pattern will be used, or some sort of averaged gain value.

Passive antennas:

For simple wire antennas, the gain can be estimated with reasonable (+/- a few dB) accuracy.  For example, a half-wave dipole above typical ground will likely be in the area of 1-2 dBi gain peak while the gain of a commercially made vertical or Yagi antenna might be available from the manufacturer's data sheet.

For multi-wavelength wire antennas such as long wires or rhombics, antenna simulation programs are suggested and may be the only reasonable means of estimating gain - but again, one will have to make a decision on what gain value to apply in light of the likely existence of lobes and nulls.

Active HF antennas:

If the feed from a conventional antenna (beam, dipole) includes an amplifier, the steps mentioned above may be used to estimate its gain and one need only add the loss of the feedline and gain of an amplifier to obtain the desired offset for the wsprdaemon.conf file.

Electrically-short active antennas such as the PA0RDT mini-whip pose a particular problem.  They are essentially a capacitive plate (or wire) that couples RF from the "aether" - but the amount of coupling (which correlates with the apparent gain of the antenna) can vary widely, the determining factors including:

• Height above "ground".  In general, the higher above the ground - earth or metal - the more signal such an antenna will intercept.

• Type of "ground".  The ground above which the antenna is mounted could be earth (dirt. soil) which is lossy or it could be a metal roof (much preferred!) to which the feedline's ground is bonded.

• Size of the "antenna".  The traditional PA0RDT antenna uses a PC board plate for coupling - but there are many variations on what size of coupling conductor - wire or plate - is used, all of which can affect the apparent gain.

• Gain response of the amplifier with frequency.  In most - if not all - implementations of an E-field whip are subject to gain drop-off with higher frequency and this is not easy to measure directly and determining a suitable number is almost entirely guesswork.

• Unfortunately, such drop-off conspires with the high RF levels at low-medium HF frequencies.  With a broadband, direct-sampling receiver like the KiwiSDR, overload will surely occur if enough gain is added to the system to attain the natural noise floor at the higher HF bands (e.g. 10 meters) unless strong compensation (e.g. something like the high-pass filter described above) is used.

However one might obtain the antenna gain, this value would be applied to the wsprdaemon.conf file:  Positive gain would mean a reduction of this number by the dB value while negative gain would mean a commensurate increase in this value.

What to expect:

In theory, the noise floor one sees should generally reflect that predicted in the ITU-R data, reproduced in figure 2, below:

Figure 2:  "Typical" noise floor for various radio environments.  Because the above chart is based on a 500 Hz bandwidth, one would subtract 27dB from its vertical axis to scale to an equivalent 1 Hz noise bandwidth when measuring noise power.
Click on the image for a larger version.

To be sure, the values depicted in Figure 2 are very generalized - and one would be lucky to approach the values of "quiet rural".  In general, these value depicted above would be most representative when the respective frequency band is "dead" (e.g. daytime for the lower bands, nighttime for the higher bands) with a "quiet sun".  While usable only as a general guideline, one should be suspect of their calibration settings if the "quiet band" noise floor values they measure on their system is markedly lower than those of the chart above.

Cross-checking noise level measurements with the receiver:

It is possible to check the noise level measurement provided by wsprdaemon using the receiver in question, but to do this two conditions must be satisfied:

• You must know the receive bandwidth.  For most SDRs this is pretty easy:  The "bandwidth" setting - which is typically at the -3dB or -6dB points - is "fairly close".

• The passband must contain ONLY noise and no other signals.  On the WSPR frequencies this may not be possible, but measuring on a "nearby" frequency (up to a few 10s of kHz away, provided that such measurements aren't affected by narrow band-pass filters that might be present).
• Wider (SSB) bandwidths are fine, but it may be necessary to user narrow bandwidths to avoid signals within the passband.

 The method of making this measurement is approximately thus:

• Note the average signal level reading (in dBm).
• Since we are looking for what amounts to a rolling average over time, the readings of occasional noise bursts must be considered which means that a "fast" AGC should be used and one should not be looking at any "peak" readings.
• Since we are likely doing a "sanity check" simply staring at the signal meter and guesstimating the average.
• It is best to do this during "quiet" conditions on a given band - that is, when it is dead:  Trying it on 80 meters during the night where there may be a lot of summer static is not recommended.

•  Note the receive bandwidth.
• On an SDR, this is likely to be the "bandwidth" figure which is probably taken from either the -3dB or -6dB points - which will be "close enough.

• Calculate the bandwidth noise power ratio.
• Because wsprdaemon's noise power measurements are scaled for 1 Hz detection bandwidth, the reading obtained on any receiver with a wider noise bandwidth than that will be much higher and the reading (in dB) must be offset by that amount.
• As an example:  If you are using a 500 Hz receiver bandwidth (e.g. a CW filter) you need to account for a 500:1 power ratio which is:
• 10 * log10(500) = 26.99 = 27 dB - that is, one would subtract 27 dB from the average level that you observed.
• In other words, to scale to the 1 Hz wsprdaemon bandwidth:
• Your signal level - (10 * log10(RX bandwidth in Hz))

Conclusion:

While it is unlikely that the typical amateur will be able to be absolutely confident of the accuracy of their absolute HF noise floor measurements, the steps above will greatly limit the uncertainty of the system's measurement to the the properties of the antenna itself.

[End]

 This page stolen from ka7oei.blogspot.com

A (semi)-typical suburban E-field whip receive system for the 630 and 2200 meter amateur bands

Even though the general availability of the 630 meter (472-479 kHz) and 2200 meter (135.7-137.8 kHz) bands to U.S. amateurs is a recent phenomenon, I've had interest in these frequency ranges for about as long as I can remember.  Back in the "old" days (the 1980s, for me) I would listen in these low-frequency ranges (10kHz to 530 kHz) using my modified Drake TR-7 which has an "LF input" on the back panel.

From the very beginning, I discovered a few things that did not work well for receiving these frequencies:

• Simply connecting an end-fed random wire to the "Low Frequency" input.
• Using my 40 meter dipole.
• Anything that was indoors.

Attempts to do any of the above resulted in either no audible signals other than a racket of power mains "buzz" that would drown out anything that I could hope to hear.  I quickly realized that there were a few signals that I could hear without too much trouble - mostly the very high-power VLF transmitters between 17 and 30 kHz and WWVB at 60 kHz, which is only few hundred miles/km away - and I knew that unless I could hear those signals really well that there would be little hope of hearing anything that was actually weak.

The "discoveries":

Figure 1:
The LF-400B active e-field whip on my
roof.  The antenna is about 5 feet (1.5
meters) above the roof, mounted to a vent
pipe.  The red ground wire connected to
the coaxial cable's shield at the bottom
of the antenna can be seen along with
choke.
Click on the image for a larger version.

Even as a teenager with limited experience and knowledge in such things I realized that at such long wavelengths even a rather long piece of wire as a receive antenna would be akin to putting a paper clip in the antenna connection of an HF rig and expect to "hear the world" - but I also knew that it was possible to hear low frequencies quite well on a very short antenna:  The short whip on my car could hear the entire AM broadcast band pretty well - so it was possible if done correctly.

These realizations told me several things:

• I would probably have to match the "short" antenna to the receiver input to be able to hear anything.  I determined that this could be done with a series inductor or some sort of high-impedance amplifier - or a combination of both.

• When in a car, I could be well-away from interference sources - such as power lines and noisy appliances - and could hear weak AM stations.  Somehow I had to keep the interference from things in the house from finding their way into my receiver.

Rummaging around in my junk box I found a large, variable inductor - probably from a scrapped TV - that I placed in series with my wire antenna and receiver - and over a limited frequency range (dictated by the adjustment range of this inductor) I noticed a dramatic improvement in the signal strength at about the frequencies that the combination of the coil and antenna provided a semblance of matching - although the noise was still substantial.

The next "breakthrough" was to wind a simple 1:1 transformer on a chunk of ferrite - probably the flyback transformer core of an old TV -  that allowed only magnetic coupling between the radio and its chassis, and the antenna and a connection that went directly to my kludgy system of buried ground rods.  By doing this, the "noisy" ground of my receiver - which was connected throughout the house with its noisy devices - was no longer referenced to the antenna.  Because the antenna must have a "ground" of some sort to "push" against I knew that if that "ground" was the radio itself, which was connected to the noisy house wiring, that this noise would, in effect, appear on the wire antenna.  This transformer effectively decoupled the two, using, instead, the comparatively "pristine" ground rod for the antenna to "push" against.  (For a depiction of this method see the external link to a paper by DL1DBC at the bottom of this page.)

Between the above two tricks an entirely new world opened up as I could now hear the (now defunct) Omega transmitters between about 10 and 14 kHz and a myriad of "NDBs" (non-directional beacons) and similar signals in the range from 190 through just below 530 kHz.  To be sure, I had to do most of my listening at night when TVs and lights were turned off, but that's when most of these frequencies propagated best, anyway!

The "LowFER" band:

Somewhere around this time I learned of the so-called 1750 meter "LowFER" band - a spectral slice from 160 through 190 kHz where legal, unlicensed operation (according to FCC §15.217 - read more here) could occur with some very strict limitations (e.g. an antenna that was, at most, 15 meters "long" and a maximum of 1 watt of input power.) but the challenge of both transmitting a usable signal with these limitations and receiving it via conventional techniques (e.g. CW) had its appeal.

It was at about this time - in the mid 1980s - that I purchased an LF Engineering LF-400B - a commercially-available active E-field whip antenna that seemed to have decent reviews in the various longwave-related newsletters to which I then subscribed.  This antenna, with a built-in amplifier and a strong low-pass filter to remove signals above 500 kHz, was much more convenient than trying to string a long piece of wire and matching it as it was rated from "3 kHz to 500 kHz".  One slight disadvantage of this - or any active antenna - is that it needs power, supplied in this case by a "power inserter" that ran from an external power supply or a pair of contained 9 volt batteries.

Being an E-field whip antenna it was still sensitive to the direct radiation of interference from the household and neighborhood wiring and appliances, but provided that I located it away from the house and "decoupled" its cable by winding as many turns as could fit on the core of a flyback transformer from a scrapped TV and grounding the shield at the antenna, it seemed to hear the background static very well - and if I could hear the background noise, there was hope that I could hear the weak signals buried within.

It was during this time period that I actively listened on the LowFER band, managing to hear a number of stations that were 200-700 miles (about 300-1100 km) away and, on one winter evening, hearing a station halfway across the continent - about 2000 miles (3200 km) away.   I also set up my own LowFER beacon that, although very modest, was occasionally heard, on CW, up to 700 miles (1100 km) away.

Comment:

Another antenna to consider for MF/LF/VLF reception is a shielded H-field loop.  By its nature, it is less-sensitive to nearby E-field energy - often that which emanates from electrical devices' interference radiating from wiring.

Another advantage of a loop is that it has a "figure-8" pattern with two nulls, allowing the possibility of rotating it such that one of these nulls is oriented toward an interference source.  The obvious disadvantage is that a loop should have provisions for rotation to steer it into the null for the worst interference - or take care of those instances where the desired station happens to be in the direction of the null.

Shielded loops are available and they can be constructed fairly easily, typically using a piece of coaxial cable.  Unless they are rather large and/or actively tuned to the receive frequency, they - like a short E-field whip - must have an amplifier that is externally powered.

Fast forward to the 21st century:

As it happens, I still have the L-400B and it has been outside, on a roof, for most of the time since the mid 1980s.  Other than having to repair it a time or two (usually due to condensation and related corrosion) it still works as well as it ever did.  While I had not been as active on LF as I once was, I'd been maintaining that receive antenna system and with the recent availability of the 630 and 2200 meter bands, interest has been rekindled.

To this end, I decided to document my receive antenna installation, showing what "works for me."

The antenna on the roof:

I will admit to a luxury that most others will not have:  My house has a metal roof.

Figure 2:
A close-up of the coax choke at the antenna.  This
choke consists of 10 turns wound on a large
ferrite bar.  The coax used is solid-dielectric RG-58.
The use of a solid dielectric rather than a foam
dielectric - such as that found in RG-6 - allowed
a very tight radius winding without worrying much
about the center conductor "migrating" and shorting
to the shield.  A cable like RG-174 would have also
been usable, allowing a tight radius and more turns.
At these frequencies, the loss of the coaxial cable is
insignificant.
Click on the image for a larger version.

The metal roof not only acts as an excellent ground plane, but it is also an effective barrier between what is "inside" my house and the "outside world".  This means that at VLF and LF frequencies, things in my house that generate noise (light dimmers, switching power supplies, TVs) are fairly effectively isolated from this antenna on the roof - at least in terms of direct radiation of energy from these devices.

If you are not "blessed" with a metal roof on your house - but you are willing to go through a bit of hassle - you could lay down a suitable ground plane:  Many people have been known to put down a layer of chicken wire on the roof or an interconnected grid of wires to act as an effective shield.  Practically speaking, it need not cover the entire roof, but if the radius of this plane is 1-2 times the height of the antenna over the roof, it will probably have reasonable effectiveness.

Somewhere this plane must be grounded and it is best that this is done via its very own ground system - which could be as simple as a ground rod - which is preferred over tying into the house's "noisy" electrical ground.

As can be seen from the picture in Figure 1 the whip antenna is mounted to a vent pipe at a height of approximately 5 feet (1.5 meters) above the roof - which happened to be the length of the piece of aluminum that I'd found to mount the antenna.  When experimenting with mounting this antenna I found that if I placed it just above the metal roof, it was very quiet and relatively insensitive - but much of that was due to the fact that the very E fields to which the antenna is sensitive decrease significantly with proximity to "ground".  By raising the antenna above the roof the signals increase very dramatically, but still seemed to be within the "cone of silence" afforded by the metal roof.

Decoupling the coaxial cable at the receive antenna:

At the time that I bought the LF-400B antenna it was offered only with a permanently attached RG-174 feedline, but after about a year of use, often hauling it into the wild to listen, away from the city, the coaxial cable fatigued and broke, so I carefully disassembled it and installed a BNC connector (later versions of this antenna have a choice of connectors as an option.)  This modification allowed me to connect a ground directly to the bottom of the antenna.

Figure 3:
A block diagram of the antenna and receive system showing the grounding and coax chokes.
Note that the "roof ground" - which is, in my case, the metal roof itself, but it could be a grid of wire or fencing material laid on the roof and is grounded elsewhere - is connected directly at the shield of the e-field whip itself, "before" the coaxial choke.  At the ground level, in close proximity to the building entry is another connection to a "clean" local ground such as several ground rods and/or some buried ground radials.
The "Inside coax choke" has the most inductance and does most of the isolating of common-mode noise currents that could otherwise "light up" the antenna with electrical noise that would be conducted from the radio system's ground connection to the power mains.  The DC power inserter puts DC on the coaxial cable to the antenna - but not on the coax to the receiver - to provide power.
Click on the image for a larger version.

As mentioned earlier, one of the "tricks" to a quiet E-field antenna is to prevent electrical noise from being conducted from the receiver and "lighting up the ground" of the antenna itself - a problem that is arguably worse than the antenna itself picking up noise, directly.  One of the better ways to to do this is to "decouple" the coaxial cable between the antenna and receiver using a large amount of inductance on the feedline - and I chose to do this several ways.

As can be seen from figure 1 there is a (red) wire connected directly to the antenna's connector that, in turn, connects to the "local ground" - that is, the metal roof itself.  By doing this, the "ground" of the antenna and the roof are at the same RF potential and the interception of "local" interference by the antenna is reduced.

Also visible in figure 1 - and in more detail in figure 2 - is an inductor in the form of a portion of the connecting coaxial cable being wound around a large ferrite rod from a discarded AM radio.  The location of this inductor places it between the antenna and the receiver and its inductance adds common mode impedance to signals that would be conducted along the coaxial cable, but will not affect the desired signals within the cable itself.  A better choke for this location would be like that depicted in Figure 4 (and described below) as it has higher effective resistance at the frequencies of interest, but since I'd already installed this one, I left it in place.

Figure 4 shows the other end of the cable just after it enters the house.  Just as it enters through the window there is another BNC connector, and connected to the shield at that point is a wire that goes directly to a grounding system that is located immediately outside the window.  Between this grounding point and the inside of the house where the connection to the radio is made the coaxial cable is wound around a much more substantial choke - this one consisting of as many turns of the RG-58 coaxial cable as will fit on a TV flyback transformer ferrite core that was scavanged from a discarded CRT TV or computer monitor.  The details of the locations of these chokes and the grounding points is detailed in Figure 3.

It is this second "inside" choke that does most of the work:  Consisting of about 20 turns, it has a measured inductance of about 15 millihenries.  In running the math we can see that this large amount of inductance is what is required to effectively isolate the coax at LF and VLF frequencies, as in:

Where inductive reactance is calculated using the equation:

Z = 2 * Pi * F * L

Where:

Z = Reactance in ohms
F = Frequency in Hz
L = Inductance in Henries

Because we are dealing with milliHenries and kHz, the "10s" parts cancel out, so:

At 500 kHz:


500 kHz * 15 milliHenries * 6.28 = 47100 ohms

Because this is a linear equation, we can then re-run the numbers which tells us that at 50 kHz, the reactance is 4710 ohms and that at 5 kHz it would be 471 ohms.


What this shows us is that even at very low (VLF) frequencies, the impedance of our rather substantial inductance is still effective, so it will work nicely at both 630 and 2200 meters - and everything in between!

Figure 4:
The indoor coax choke consists of 20 turns of  RG-58 wound on a TV flyback transformer core.  This choke, with a measured inductance of about 15 milliHenries, provides excellent isolation even down below 10 kHz.  If a flyback transformer core cannot be found a suitable choke can be wound on a high-permeability ferrite core using smaller (e.g. RG-174) coaxial cable as described below.  Note that to be effective at these frequencies this choke really does need to have at least several milliHenries of inductance!   See the links at the end of this page for sources of suitable ferrite devices.
Click on the image for a larger version.

Obtaining the inductance:

While "current-mode" 1:1 baluns that isolate the feedline in the manner we desire are readily available, unless they were specifically designed for LF and VLF use they do not have enough reactance to operate effectively at these low frequencies!  What this means is that unless a suitable product is offered by one of these companies that is has been designed for LF and VLF use, they will not work well!  This means is that you will probably need to construct your own coaxial choke.

Using flyback transformer cores:

Many years ago it was pretty easy to scavange flyback transformer cores from old CRT-based TVs or computer monitors, but these are getting harder to find - but this is mostly a good thing since these transformers were part of the very device that caused a lot of interference at VLF and LF frequencies!  Once one manages to get the core out of an old flyback transformer in the first place (sometimes a trick in and of itself!) the fact that these cores are in two pieces makes it easy to wind the coaxial cable over one half and then assemble it.  When I come across a flyback transformer, I often resort to putting it in a toaster oven and heating it so that the glue softens.  Often, the core breaks - but ferrite typically breaks very cleanly and the two pieces can be rejoined using a drop of cyanoacrylate (e.g. "super") glue with little change in performance.  Before using the ferrite core, make sure that any plastic or fiber shims between the two halves are removed:  These were important in the role as a flyback transformer, but they will reduce the inductance of our choke and aren't helpful to us in our quest to quash QRM.

If a flyback transformer core is not available, what can be used, instead?

Using high-permeability toroidal cores:

While not as convenient as a flyback transformer core - which can be disassembled during winding - a ferrite toroidal core can be used, instead.  To maximize the number of turns, smaller coax such as RG-174 would be used and the connectors installed/connected after winding was complete.

Take, for example, a common ferrite material designed for low frequencies - "Mix 75" (sometimes called "Mix J") with a typical permeability of about 5000.  A reasonably large toroidal core would be the FT-240 (the complete part number would be either "FT-240-75" or "F240-75").  Note that the ferrite mixes that one would normally use for things like HF baluns aren't ideal for this purpose as they have lower permeability.

Extrapolating from a data sheet and rewriting the equation we can see that if we can manage to wind 30 turns on this particular toroidal core, we can expect:

L = Al * (T/1000)²

Where:

L = Inductance in mH
T = Turns
Al = mH per 1000 turns from the spec. sheet - 6850 for an FT-240-75

So,

6850 * (30/1000)² = 6.165mH

Clearly, this is a bit less than half as much as I'd measured on my discarded TV flyback, but if we use the equation above we still get 194 ohms at 5 kHz and over 5 kohms at 2200 meters - a respectable amount of reactance!  Using this size of core (an inside diameter of 1.4 inches/3.5cm) it is likely that more than 30 turns of RG-174 could be wound on it - and if you make this type of core, by all means, put as many turns on at as you can!

Unfortunately, the "Mix 75" toroids are not as easy to find as typical toroids designed for higher (HF) frequencies and if we use a more common type such as Mix 31 the result will be between a quarter and a fifth of the inductance for the same number of turns whereas "Mix 77" will, for the same number of turns, yield about 1/3 of the inductance as Mix 75, but this would still imply between 1 and 2 kohms at 2200 meters - still quite good.

Where does one get this sort of toroid?   Toroids can be found at a number of places, including:

• Palomar Engineers (link) 
• Amidon Associates (link). 
• Another distributor of some of these devices is the web site kf7p.com - link.

Again, while "Mix 75" is preferred, "Mix 77" is the second choice - and cores may be stacked to increase the inductance for a given number of turns.

Another possibility - Common-mode chokes:

While a coaxial-based choke is preferred, there are other devices - possibly in your junk box - that may be suitable:  A common-mode choke used for power supply filtering.  The best place to find these is from scrapped switching power supplies - such as those used in computers.

Figure 5:
An assortment of power line filtering chokes and devices.  In the upper-left
is a self-contained AC line filter, but it is not suitable for this purpose as it
is designed to block all RF - both differential and common-mode.  All of
the other devices are dual-winding common-mode chokes that allow
differential currents to pass, but will block common-mode currents - but
not all of these devices are suitable for our purpose - see text.
Click on the image for a larger version. 

Figure 5 shows an assortment of typical devices - but not all of them are suitable.   As noted in the caption, the self-contained power line filter (upper-left) blocks all RF and wouldn't work, but the other devices allow differential currents to flow while blocking common-mode currents - which is what we want.

In order for these devices to be suitable for our purpose, they need to have:

• Adequate inductance.  As we noted above, we need milliHenries of inductance to effectively choke out interference at LF and VLF frequencies.  The smaller toroidal chokes shown - typically those wound on toroidal cores - have hundreds of microHenries of inductance which may be suitable at 630 meters, but could be marginal at 2200 meters.  For example, a choke with 100 microHenries per winding will offer about 295 ohms of reactance at 630 meters, but only 86 ohms at 2200 meters.  Because we want as much reactance as possible - at least in the many hundreds of ohms - we would hope to do better!
• Good balance.  All of these chokes consists of two identical windings and the idea is that if a common mode signal appears across both windings, they will be suppressed.  If, however, the two windings are not identical, this suppression will be incomplete.  It is likely that the "transformer-looking" chokes (e.g. those that do NOT look like toroids) will have reasonable suppression at 2200 meters - and maybe even 630 meters - but as one goes up in frequency even more, the imbalance will grow.
• Low loss to differential signals.  The reason that we can pass a signal through a coaxial cable wound on a large piece of ferrite without affecting the signal being carried by that cable is that the coaxial cable, by its very nature, is fairly low loss to the signals carried within where the signal on the inside conductor of the coax is precisely equal and opposite to that carried on the shield.  If one has separate windings, each carrying an equal and opposite signals, imperfections in these two windings - sometimes the same as those that cause imbalance - can cause degradation of those signals.  As one goes up in frequency these ferrite cores - which are formulated to block low frequencies - can start to get lossy - and this doesn't include the self-capacitance of the windings which can cause other things to happen, such as strange resonances or coupling.  In other words, they may work find at low frequencies, but "fall apart" at higher frequencies such as 160 meters (1.8 MHz) and up.

Figure 6: 
An example of how a bifilar (or similar) choke would be
connected to a coaxial cable.
The diagram above depicts how the two windings would be
connected, keeping straight which is the "center", and that
which is the shield of the coaxial cable.  The dots indicate "phasing" -
that is, same ends of the two windings connect to the antenna side and
the other ends connect to the receiver side.  On the "antenna" side's
coax shield would be connected our "quiet ground".
Click on the image for a larger version.

In short, the suitability these devices for our purpose is best determined experimentally.

How it is would be connected:

Figure 6 shows how such a device would be connected to coaxial connectors.  Note that the winding for the shield on one side of the choke connects to the same shield on the other side.  In theory, this wouldn't matter at RF, but because we may need to conduct DC to power the active antenna, we would also need to preserve the polarity.

Not also that both sides of the input and output coaxes connect to the same "side" of the dual winding choke as indicated by the dots - in other words, the two windings are in phase with each other:  Were either one of the windings (ground or center conductor) "flipped", this choke would do exactly opposite that which we desire - that is, the signal on the coax would be blocked, leaving only noise!

For an inductor such as that depicted in Figures 5 and 6 that is not wound with coax, it doesn't matter which side is the shield and which is the center - just as long as the windings are "shield-to-shield" and "center-to-center".

Another example of feedline choking and grounding:

Figure 7:
RG-174 coaxial cable wound on a TV flyback core.  About
55 turns fit on this core yielding a measured inductance
of a bit over 1.5 milliHenries.  If this core
had equal gapping in both of its "legs" (see text)
the inductance would have been higher.
Click on the image for a larger version.

Figure 7 shows another example of how an E-field antenna's feedline (not mine) was isolated.

In this case - which just happened to be another LF-400B - RG-174 coaxial cable was permanently attached to the antenna.  This cable was cut, leaving about 20 feet (approx. 6 meters) of it still attached to the antenna and, leaving a "service" loop of about 1.5 feet (35cm) the remainder was wrapped on the flyback core of a discarded computer monitor.

It is worth noting (again) that these flyback cores are usually "gapped" - that is, a small - usually plastic - insulator is placed between the two halves of a core to prevent it's being saturated.  On some of these cores there are two equal gaps - one on each of the two mating surfaces and if these are removed, the two ferrite surfaces mate closely.  In the case of the core in Figure 7, only one of these mating surfaces had a gap, meaning that one side mated closely while there was a gap on the other side:  Sometimes the cores are symmetrical and one can be "flipped" to eliminate this problem, but that was not the case with the core in Figure 4, so a slight gap was inevitable and this has the inevitable result of reducing the total inductance of the core.

In the case of the flyback pictured in Figure 7 the plastic gapping material was carefully retained and a very thin layer of epoxy was put on the two sets of mating surfaces and the metal bail holding the two together was reinstalled, the cores being worked back-and-forth to squeeze out extra epoxy.  Once this was done epoxy was applied to the wire bail itself to keep it in place.  After the epoxy was allowed to cure, the remaining RG-174 coaxial cable was wound on it, filling it up.

Figure 8:
A wire attached to the shield to permit grounding, necessary
because the coaxial cable was permanently attached to
 the antenna, preventing a connection from being made
at then antenna, between it and the choke.  See text.
Note that the ground wire emerges from the "downstream"
side of the coax.  When installed, this ground wire will
face down to reduce the probability of moisture ingress.

Click on the image for a larger version.

Figure 7 shows this core mounted in a plastic "pull" box intended for non-metallic electrical conduit.  On one end of the box is mounted an "F" connector to which the end of the RG-174 is soldered while the other end - connected to the whip antenna - emerges through a plug:  If you do it this way you will surely want to pull the coax through the plug before winding it on the core!  After it was assembled, it occurred to us that we should have put the "F" connector on the same side as the plug so that they could both be faced downwards.  Between the receiver and the choke box, ordinary RG-6 TV coax will work:  The impedance mismatch/loss is unimportant at this frequency, in this application.

Figure 8 shows a bit of detail about the grounding of the antenna.  This particular antenna has a permanently attached cable and the owner didn't wish to modify the antenna to add a coaxial connector to it.  That which follows was done before the coaxial cable was wound on the ferrite core.

The ground connection needed to be made directly to the cable's shield and this was done by carefully baring a bit of the shield by removing a small amount of the outer jacket and then using a hot soldering iron to quickly make the connection without melting the inner dielectric - a bit of a trick to do if one isn't skilled in the art of soldering!  To make this weather proof the connection was covered with a thin layer of thermoset (e.g. "hot melt") glue and a small piece of heat shrink tubing was slid over the joint and shrunk.  Over the top of this a thin layer of RTV ("Silicone") sealant was spread over the entirety of the connection and another, slightly longer piece of heat shrink tubing was installed and shrunk - and then another thin layer of RTV and slightly longer heat shrink tubing.

While this sounds like overkill, it should prevent moisture from finding its way in between the jacket of the coax and the tubing.  Finally, this connection should be oriented at the time of installation such that water runs away from it - which is to say, the part with the wire coming out from underneath the tubing should be facing down.

More information about interference reduction:

While the above techniques will go a long way to reduce the amount of noise picked up by an E-field antenna - and, to a degree, any antenna - it is too-often the case that there will be some device that simply radiates a lot of noise.  While at HF frequencies and higher it is possible to reduce this noise with the application of large ferrite devices on cables, power cords, etc. this tactic simply does not work well at VLF/LF/MF frequencies because it takes so much reactance (inductance) to introduce enough effective resistance in the wire conveying this noise and a "snap-on" choke simply cannot do this.  Even if a device contains "good" noise suppressing components (not all do!) they simply may not be very effective at VLF/LF/MF frequencies.

If you are interested in listening on the LF and MF amateur bands, the necessary first steps are outlined above:  Do what is necessary to prevent noise from being conducted out, onto the antenna in the first place.

Once that is done, you may need to "seek and destroy" devices that are particularly egregious when it comes to generation RF "grunge" - and the typical suspects are switching power supplies, light dimmers and some brands of LED lights.  Plasma TVs are notoriously bad interference generators, but since they are no longer being made, their contribution to the miasma of QRM is slowly decreasing as they die off.

The best way to find noise that you can do something about is to power the receiver from a battery (NOT including an inverter!) and turn off all of the power to the house - including shutting down any UPSs that you might have.  If the noise decreases or goes away, turn on one circuit at a time until it returns and upon finding the circuit, isolate the specific device that causes the problem.  If the noise is just the same with your power off as it is on, there may be a noisy power line nearby and/or a neighbor may have a noisy device - and how you deal with those two entities is up to you!

If you find a device (or devices) that generate lots of interference, they might either be replaced with "quieter" ones or modified to be quiet.  Unfortunately, the latter can be a challenge and the links below include techniques for doing this.  If your goal is interference reduction at VLF/LF/MF - and you are constructing better filtering - remember that the higher-inductance chokes will be best!

How well does my receive antenna system work?

In the late evening and overnight, I can easily hear the "band noise" - that is, the sounds of the ionosphere and propagated storm static.  During the day time the noise level is typically lower as it seems as though propagated noise from a wide geographical area is suppressed somewhat - possibly by the formation of the ionospheric E-layer.  During the "busy" hours - particularly from, say, 5 to 11 PM, there can be a bit of interference from other peoples' TVs, appliances and whatnot, but it is usually not severe enough to completely quash reception.

In my ham shack I have some track lighting over the workbench that is equipped with LED floodlights and is controlled by a light dimmer.  While I do not "hear" the LED's switching power supplies, I do get a significant "buzz" on 2200 meters from the dimmer itself - but I don't hear it on 630 meters.  The work-around for this is to use a smaller work light near the workbench - both of them being fluorescent - one having an iron ballast and the other electronic - but neither of them causing detectable interference on either 630 or 2200 meters.

For the past 5 years or so there have been a number of Canadian amateur stations (who have had access to the frequencies around 630 and 2200 meters for a while) plus some U.S. based "experimental" stations that have also operated on a number of other frequencies and in this time, I've been able to "receive" these stations which are typically using a digital mode like WSPR or a more analog-like mode like QRSS (slow-speed Morse code) - both typically being detected by computer.  The operational frequencies of these stations has varied from above 500 kHz to below 30 kHz, depending on the authorized frequencies of the various experimental stations and I've generally been able receive such signals including a number of stations operating in the 470-500 kHz range across the U.S. and an experimental station operating near 29 kHz (yes, 29 kHz!) from New York state to my QTH in Utah - a distance of about 2000 miles (approx. 3200km).

In the relatively short time since U.S. amateurs have been allowed to operate on the 630 and 2200 meter bands I've heard several stations on both bands - some well enough to have copied using Morse code via ear and, possibly, even SSB voice.  As the northern hemisphere descends into winter - and as more amateurs receive authorization and put their systems on the air - I expect to hear even more stations.

One device in my arsenal is a "Line Synchronous Noise Blanker" - that is, a device that will mute the antenna signal when an interfering pulse - which is usually in sync with the power mains - comes in.  This devices is adjusted manually and can go a long way to knocking out this type of noise.  This device is described on this page:  A Line-Synchronous Noise Blanker for VLF/LF/MF use - link.

Links to other articles about power supply noise reduction:

• Completely containing switching power supply RFI - link.  Sometimes it can be difficult to quiet a switching power supply, so it may be necessary to put it in a box with strong filtering on all of the conductors that enter/leave.
• Minimizing VHF (and HF) RFI from electronic ballasts and fluorescent tubes - link.  Electronic light ballasts, like many switching power supplies, operate in the LF frequency range so "cleaning them up" at VLF/LF/MF frequencies can be a challenge.
• Quieting high current switching power supplies used in the shack - link.  This page describes techniques that can be used to reduce the amount of RF energy produced by switching power supplies that you may be using to power your radios.  Again, higher-inductance chokes may be required at VLF/LF/MF frequencies.
• Reducing switching supply racket - link.  This describes techniques that can be used to beef up the filtering for switching supplies in general.

In case you get intermodulation distortion (e.g. overload) from nearby AM broadcast stations:

•  An effective low-pass filter for 500 kHz and below - link.

Other information about the use of active antennas at VLF, LF and MF frequencies:

• Discussion from the DL1DBC web site about active antennas, including their operation and installation - link.
• Construction and installation of a PA0RDT whip by VK6YSF - link.
• Discussion of E-field whip antennas by PA3FWM - link.

Once again, here are some links to suppliers of suitable ferrite devices:

• Palomar Engineers (link) 
• Amidon Associates (link). 
• Another distributor of some of these devices is the web site kf7p.com - link.

While "Mix 75" is preferred, "Mix 77" is the second choice - and cores may be stacked to increase the inductance for a given number of turns.

Final comments:

The L-400B still seems to be available - at about twice the price as it was when I bought mine in 1986-7.  The page with information on this and similar products may be found here - link.

In addition to the L-400B, there are now other active whips, including the AMRAD active whip, the PA0RDT "Mini-whip" and variants on those designs which may or may not include a low-pass filter to remove mediumwave signals.  All of these are reported to work well, but be aware that some receivers have difficulty dealing with signal from strong, local AM broadcast transmitters.

I have built my own copy of the PA0RDT mini-whip and I tried it only briefly:  When I put it up, it was badly overloaded by my own HF WSPR transmitter - so I took it down and put the LF-400B back up as its built-in low-pass filter makes it pretty immune to being overloaded nearby HF transmissions.

[End]

This page stolen from ka7oei.blogspot.com