Making LED headlights RF-quiet

TL;DR

 If you have LED headlights that are causing RF interference, ferrites alone will probably not be enough to completely solve the problem:  You will probably need to put the offending switch-mode LED controller in "RF Jail" as described below.  While VHF/UHF is the target of the efforts noted below, they should be equally effective at HF.

* * * * *

We live in a world of RF - and an increasing amount of this is from devices that are not intended to produce radio-frequency energy.  These devices have proliferated in the past several decades and surround us at home, work and in our cars.

Figure 1:
The vehicle in question with LED headlights -
which are now RF-quiet on all bands.
Click on the image for a larger version.

Generally speaking, modern internal combustion vehicles are in RFI terms "pretty quiet" - especially compared to their electric and hybrid counterparts - and someone in such a vehicle will experience less RF noise when they are "out and about" in a rural-ish area than they will at home.  For amateur radio operators, this is a good thing as mobile operation often includes weak signals - whether this is on HF, or on VHF/UHF with weak signals from distant repeaters or during simplex operation, with intervening terrain.

Effective mobile operation is therefore contingent on a vehicle that is intrinsically "RF-quiet", but this also means that any accessories that you might add to this vehicle also be RF-quiet as well.  These days there are any number of things that you might throw in your car that can spoil an otherwise-clean RF environment and the short list includes USB chargers¹ , GPS receivers and extra lighting ², to name but a few.

Real-world case - Aftermarket LED headlights

A friend of mine recently installed aftermarket LED headlights in his older Honda CRV ³.  It took a while to correlate the cause, but he eventually noticed that when the headlights were turned on, he lost significant weak-signal sensitivity on 2 meters:  Around town and with stronger repeaters, the effect wasn't really noticeable, but when the repeater was distant - or when communicating simplex (e.g. direct - with no repeater) with a weak/distant station - turning on the headlights dramatically reduced received ⁴ signal quality and range.  As it happens, he has more than one VHF/UHF radio in his vehicle and although both were affected, the one with its antenna mounted to the front fender - much closer to the LED headlights - was more severely impacted.

Retrofit LED headlights typically consist of two modules:  The LED module itself and the "controller" - which itself is a switch-mode power supply to regulate the current to the LEDs as well as switching arrays of LEDs for high/low beam - and even color temperature in some models.  Sometimes these units are combined, but in this case, they were separated with a short cable, the "controller" being separate from the LED module that is mounted in the headlight housing.

Even knowing that it was futile, we tried putting clamp/snap-on ferrites on the cables to/from the LED modules and its controller, but all that we could manage was a slight reduction in interference that was hard to quantify, taking the problem from being "terrible" to just "awful".  This was not unexpected:  Under the very best conditions ferrites alone may provide 15-20dB of reduction in conducted energy (2-3 "S" units) but at VHF/UHF getting anywhere near that much attenuation is very difficult - and measurements indicated that even if we did achieve 15dB reduction across the board, the "jamming" of weak signals by the headlights' switch-mode controllers would still be significant:  It would be like taking an "10-over S-9" interfering signal down to just "S-8" - still pretty bad!

Methods of filtering

As noted earlier, simply putting ferrite devices on the conductors can reduce the amount of conducted energy, but their effect is typically limited - likely 15-20dB in the best case when this is the only method employed and ferrites alone are likely to be effective only when the interference is very slight to begin with.  Ferrite devices - such as beads - simply add inductance (and thus loss or impedance) to RF energy while leaving DC and low-frequency signals alone, but these devices have limitations:  Properties such as self-resonance and the permeability of the magnetic material vary wildly with frequency and high levels of attenuation are difficult to attain - particularly at high frequencies (e.g. VHF/UHF) where even short conductors carrying RF currents can radiate with reasonable efficiency - particularly when the noise-generating device and the receive antenna are in close proximity.

As discussed previously on this blog ⁵ one sure-fire way to quash such interference is to put the offending device in "jail" - that is, enclose it completely in a metal box and use both inductive and capacitive filtering on each and every wire to prevent RF energy from being conducted and the metal enclosure can prevent direct radiation from leads themselves - particularly important at VHF/UHF but less-so at HF.  Done properly, this method can be "completely" effective ⁶ in preventing interference.

Figure 2:
Simplified diagram of the method of filtering.  The inductance - provided by the ferrite beads ("L")
provide "choking" impedance to the RF currents being carried on the wires from the LED controller
(the "noisy device") before they connect to the feedthrough capacitors ("FT") in order to maximize
their efficacy.
Click on the image for a larger version.

Figure 2 shows a very effective method of dealing with this problem and it involves inductance ("L") and capacitance in the form of "feedthrough" capacitors (marked "FT").  The inductance is in the form of a ferrite bead installed on each of the conductors between the "noisy device" and the capacitors that increase the impedance at radio frequencies on that conductor.  The capacitors are then used to shunt the remaining RF energy to the local "ground" which, in this case, is the partition on which the feedthrough capacitors are mounted (this will be discussed shortly) and also the metal enclosure in which the noisy device is mounted.

The intent here is to prevent RF currents flowing through to the "external connections" where the wires themselves will act as antennas to radiate the RFI generated by the noisy device.  By shunting RF to the partition - and the metal box itself - the remaining RF energy will be minimal and confined within the enclosure.  The above configuration is easily capable of attenuating RF energy from HF through UHF by 30dB or better (more likely 50-60dB at some frequencies) - a value far higher than ferrite alone.

The metal box containing the electronics and filtering offer another important benefit:  As even short conductors a few inches/cm long can radiate at VHF/UHF, placing the noisy device and its conductors within the metal enclosure will prevent this.

An important feature of the design is that the inductances ("L") are located between the noisy device and the feedthrough capacitors.  As these inductances (ferrite beads) offer 10s to 100s of Ohms of impedance to the RF signal, this allows the very low impedance of the feedthrough capacitors at those same frequencies (which is likely an Ohm or less at higher frequencies) to more-effectively shunt that energy.  If the ferrite beads - or some sort of inductor - had not been installed, the shunting of the low-impedance RF energy from the noisy device would have been far less effective.

Figure 3:
An assortment of feedthrough capacitors.  The top two
rows are of the "screw-in" type, typically mounted to chassis-
walls and bulkheads while those on the bottom row are the
"solder-in" type as used in the partition in this project.
Click on the image for a larger version.

In this case, we were preventing RF from a noisy device from leaving the enclosure - but if we were trying to protect a sensitive device FROM RF energy from a nearby transmitter we would place the inductances on the conductors coming from the outside world as well to allow the capacitors to better-perform their function.

Feedthrough capacitors and the partition

A bit more needs to be said about "feedthrough capacitors".  Even if you are "into" electronics, you may not have seen these devices for the simple reason that they are a bit esoteric - and, perhaps, they are not quite as prominent as they have been in the past.  Figure 3 shows an assortment of feedthrough capacitors:  The top two rows are chassis-mounted types that are held in a pre-drilled hole by a nut while the three on the bottom row are of the "solder-in" type.

A feedthrough capacitor has a wire that passes through its center with the "capacitance" surrounding this wire over the length of the of its body and the other "plate" of this capacitor is the body of the feedthrough capacitor itself.  By being constructed this way, there are no wires or leads between the "capacitor" part  and either the signal or ground wires and as such, any series inductance - which would reduce the efficacy of the capacitor - is minimized.

Figure 4:
Solder-in type feedthrough capacitors soldered
to the brass partition.  This large sheet of metal
provides a low-impedance RF path to the
common "ground" (e.g. case) to contain RF
entirely within the metal case.
Click on the image for a larger version.

Compared to a "normal" capacitor with wires, such a capacitor is far more effective at bypassing RF energy to "ground" and it also suffers much less from parasitic issues like self-resonance - a property in which the capacitor and its internal inductance form a resonant circuit can cause it to practically "disappear" from the circuit (e.g. cease to be effective) at certain frequencies.  For this application - where it's particularly important to reduce RF interference at VHF and UHF - the use of feedthrough capacitors is - along with some inductance - a nearly foolproof method of attenuating such energy without resorting to surface-mount components and/or a specially-designed PC board. ⁷

Figure 4 shows nine feedthrough ⁸ capacitors soldered to a brass partition (the soldering to the plate is on the opposite side) and as can be seen from the photo, these are capacitors that have a wire that runs through them.  As such, they have no "ground lead" aside from the outside of the body of the device and all of them are tied together on the same piece of metal.  This method assures a low impedance RF path between all of the capacitors and since the partition itself is bolted to the aluminum case (see Figure 5), it, too, is well-bonded.

Putting it in the box

Figure 5:
The LED controller in the box w/filtering.  By
containing RF currents within the box, both
common-mode and differential RF currents
on the leads are reduced to near zero.
Click on the image for a larger version.

To eliminate direct radiation from even the very short leads, the LED's controller which can cause problems at VHF/UHF, ferrite beads ⁹ and feedthrough capacitors are contained within a box as seen in Figure 5.

Toward the top of the image we see the switch-mode controller for the LED headlights, bolted inside the case (which also helps dissipate heat) and farther down we see that all nine wires (three for power, the remaining six to the LED module itself) connect to the feedthrough capacitors on the partition. Each of these wires has its own ferrite bead and these wires go directly to their respective feedthrough capacitors on the brass partition, which is held in the case with screws.

Below the partition (see Figure 5) are the wires that connect to the outside world:  On the right are the three wires that go to the power supply (e.g. the original connector to the headlights) while the gray cable on the left goes to the LED module.  The original LED retrofit had very short leads - on the order of 5" (13cm) for the headlight connector and another set of similar length to the LED module - and this made the installation a bit challenging as there was just enough wire to make the connection between the controller and the capacitors on the partition and install the rather large ferrite beads.

The box containing everything is die-cast aluminum and it's a bit larger than necessary - but it was the only size for which I had two identical cases and also large enough to accommodate the LED controller and the filtering.  As this box is quite a bit larger than the original controller, rather long wires had to be used to allow it to be placed where there was room in the rather crowded engine compartment, somewhat away from the headlights.

Figure 6:
As the original cables were very short (about 5", 13cm) longer
wires had to be spliced to allow placement of the large die-
cast box.  This shows the male headlight connector and the
LED module spliced to the cable and covered with nylon web.
Click on the image for a larger version.

At the opposite end of the wires it was necessary to splice the added cable to the LED unit's headlight connector and LED module and this was carefully done using soldered connections insulated with head-shrinkable tubing, all of which was covered by woven nylon braid for protection and a neater appearance.

It's worth noting that there is no effort to "ground" the box itself to anything.  While the LED controller happens to be connected to the box with a mounting screw, the fact that RF currents are contained within the box by means of the bypass capacitor means that there are no such potential differences outside the box.

The result

With the added length of the cables, there were nooks and crannies into which the die-cast box with the LED controller could be placed within the engine compartment of the Honda CRV.

The real test came when a distant 2-meter repeater was keyed up to cause it to send its ID:  The return signal was very weak and noisy - as hoped and expected - but there was no difference in the way that it sounded when the lights were switched on and off.  While admittedly unscientific, this test tells us pretty much everything that we needed to know:  Whatever RF interference there is that might be escaping the box and its filtering is well below the level at which it can be detected and the problem is considered to be solved!

* * * * *

Footnotes:

1. The topic of "very noisy" plug-in USB chargers effectively "jamming" VHF/UHF reception was discussed on this blog several years ago - see:  "How USB car power power adapters can ruin 2 meter mobile reception" - link and its follow-up article:  "A 'quiet' 5 volt USB car power supply" - link.
   
2. This same friend frequently volunteers in public service events involving runners and cyclists on roads where it is required that yellow/amber lights be used to minimize hazards.  Certain makes/models of these lights have been observed to produce tremendous amounts of RF energy that effectively quashed all 2 meter reception,so they were sent back to the seller until he found a unit that was "quiet".
3. Check your local regulations regarding retrofitting of headlights with equipment other than that of the type provided by the original manufacturer.
4. A low-level increase in the noise floor in the proximity of the vehicle would have absolutely no effect on the transmitted signals, but the result of this interference is that the station in his vehicle became an "alligator" - all mouth, no ears - meaning that he was able to "talk" much farther than he could hear.
   
5. Whereas simple capacitor (shunt) or inductor (series - and this includes ferrite devices) may reliably attenuate an offending signal by 15-20dB or so at best (very generally speaking) combining both types of reactance - "L" (inductor) and "C" (capacitor) - appropriately can provide many 10s of dB of attenuation if done properly - easily 30-60dB for simple circuits.  This greater amount of attenuation is far more likely to be able to put the interference from the device well below the noise floor of the receive system.  This is the technique used in footnote #1 (above) and explained in some detail in the blog entry "Completely containing Switching Power Supply RFI" link.
6. "Completely" eliminating conducted RF is actually impossible, but reducing it by 30-60dB is likely to attenuation it below the level of detection.
7. The use of surface-mount components - like capacitors - with their lower parasitic reactance than their counterparts with leads - can be used very effectively to filter RF, but several cascaded stages of such capacitors and inductors - and careful layout of a PC board - are likely to be required to obtain sufficient attenuation.  "Feedthrough"-type surface-mount capacitors are also available - which have excellent performance - but these, too, require a properly-designed PC board.  With the solder-in feedthrough capacitors, a brass partition was used as it was easier to drill and solder to than one of copper.  If screw-in feedthrough capacitors were used, the partition could have been aluminum.
8. Nine wires are needed for the headlights:  Three wires are from the vehicle itself (low beam, high beam and "common") with two more wires for the fan and two each wires for the pair of LED emitter arrays.  As an aside, this and many other retrofit LED headlights are polarity-insensitive (e.g. it doesn't matter if the "common" lead is at V+/battery+ or ground) since different vehicle manufacturers connect the headlight's "common" pin to either V+ or ground.
   
9. The ferrite beads used in this project were Fair-Rite 2643000801.  These use "43 mix" ferrite and are 0.295" O.D., 0.297" long and 0.094 I.D. (7.5x7.55x2.375mm) and are able to accommodate the wires + insulation of the conductors from the LED controller.  This material has a typical impedance of about 94 ohms at 100 MHz and cost about $0.24 each in single quantity at the time of writing.  I used them primarily because they were on hand.
   

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This page stolen from ka7oei.blogspot.com

[END]

Locking the Icom IC-910H to an external 10 MHz (GPS) reference

In late 2009 my friend Bryan, W7CBM, came to me with a project that he had in mind:  "Can we lock my Icom IC-910H to my 'Z-box'?" - in other words, could the 10 MHz output from his Z-3801 GPS Disciplined Oscillator - known to be accurate to better than one part in 100 million - be used to lock his tri-band (2 meters, 70cm and 23cm) all-mode radio to frequency?

Figure 1:
The front panel of the modified Icom IC-910H.
Click on the image for a larger version.

During the initial discussion he'd brought with him an article where Rex, VK7MO, had done a similar thing (see the article on the VK3HZ site from the web archive - link) using an external box to provide a precise version of the radio's 30.2 MHz reference - but he wanted it to be contained entirely within the radio.  

In looking at the requirements and designing the circuit in my head, I decided that we could make it simpler, smaller and easier to use - and with these ideas in mind, I wrote down the specifications for a 30.2 MHz fundamental-mode crystal and he sent an order off to International Crystal.

About 2 months later - in early 2010 - we got back together in my ham shack, crystal in hand, and it was then that I decided that I'd better get around to designing the circuit, so I scribbled the vestiges of a schematic onto a piece of paper and built several circuits that would fit into the aluminum box that Bryan had milled out. At the end of about 3 hours we had a circuit that would faithfully lock the 30.2 MHz crystal oscillator to a 10 MHz external source.  This circuit was fairly small and consisting of two boards:  The amplifier/counter/PLL section wired on prototype board while the VCXO itself was constructed "dead bug" on to a piece of copper-clad PC board material as seen in Figure 3.

"Patience is a virtue - but this is ridiculous!"

And that was where it stopped.  In a case of "out of sight, out of mind", "other fish to fry" - or any number of other excuses - the partly-completed lock unit stayed on a shelf in Bryan's ham shack for a decade, in plain sight.  When I'd go over to his shack, I'd see it as a reminder of a project yet to be completed, but it had become a fixture and was often overlooked.

Until recently.

As it happened, we both had more time available with the onset of winter and we carved out Wednesday evenings to get together to work on various projects and this, being the most senior and nearest completion, came to the top of the pile.  Over the course of a couple evenings we worked on it, having to pause occasionally to get a part, modify some aspect the circuit's implementation, do some physical machine work, or because we ran out of time - but it is now complete!

How it works:

The schematic diagram is depicted in Figure 2, below.

Figure 2:
The schematic of the lock unit for the IC-910H.  This schematic is a reverse-engineered  version of the (now lost) originals and is likely to be mostly correct.
Click on the image for a larger version.

The VCXO:

The heart of the unit is Y201, a 30.2 MHz fundamental mode crystal in a Colpitts oscillator.  Using D201, a varactor diode (approx. 5-20pF) its frequency is made variable, the center of the electronic tuning range being adjusted by trimmer capacitor C201.  The output of the oscillator is buffered by emitter-follower Q202 to isolate the oscillator from the load.

The 30.2 MHz output goes two places:  To Q103, the 30.2 MHz amplifier, and also to a low-pass filter consisting of C208, L201 and C209 which is then output to the IC-901H's synthesizer.

The 10 MHz chain:

Figure 3:
The lock unit under test prior to installation in the case.
Unfortunately, this is the only picture that I got
of the oscillator portion.  The small PCB is the RF sense
circuit, built using SMD components by Bryan.
Click on the image for a larger version.

The 10 MHz input - which can come from a GPS Disciplined Oscillator (GPSDO), a 10 MHz oven-controlled oscillator (OCXO) or a Rubidium source - is input to and amplified by Q101 to a logic level and buffered by U1a, one section of a 74HC86 quad XOR gate.  

The output of U1a is also applied to a 74HC40103 which is wired as a divide-by-50 counter to yield a 200 kHz output - and this is applied to U2a, a 74HC7474 divide-by-two counter to yield a 100 kHz square wave.

The RF sense circuit:

A sample of the 10 MHz signal from U1a is also applied to the input of Q301, which amplifies it:  This RF gets rectified to DC by D301 and D302 and its presence turns on Q301 which pulls R303 to ground and turns off Q301 which is connected to U301 - a 5 volt regulator that is connected to the +5 volt lead of the original TCXO in the IC-910H:  In this way, the internal oscillator in the IC-910H is enabled when there is no 10 MHz signal, but disabled when it is connected.

Also connected to the emitter of Q301 is PNP power switch Q203 which, when R208 is pulled to ground when Q301 turns off, applies power to U201 - a 9 volt regulator - to power up the 30.2 MHz oscillator when the external 10 MHz source is applied, preventing both oscillators from being turned on at the same time.

The Harmonic mixer: 

A sample of the 30.2 MHz signal applied to Q103 is amplified and applied to U1b, another XOR gate buffer, which is then applied, along with the 10 MHz from U1a, into U1d - yet another XOR gate.  This gate acts as a harmonic mixer:  By virtue of the multiplying action of the XOR gate, the 3rd harmonic of the 10 MHz input mixes with the 30.2 MHz input and at the output of this gate is a small amount of the difference frequency - 200 kHz - which easily is filtered by L101 and C103 and amplified by Q102.

Figure 4:
The unit in place - final test.  SMA connectors are used for
10 MHz input and 30.2 MHz output and
feedthrough capacitors are used for the 13.8 volt DC
input and the switched 5 volts for the TCXO.
Click on the image for a larger version.

The use of a harmonic mixer is a very old technique and it has an advantage of simplicity over a more "conventional" digital divider network - albeit more "analog".

A more "conventional" way of doing this might be to divide both the 30.2 MHz and 10 MHz signals down to a common sub-multiple - say, 200 kHz - but to do so would require both a divide-by-50 (to take the 10 MHz down to 200 kHz) and a divide-by-151 (to take the 30.2 MHz down to 200 kHz).  This method works, but adds the a bit of hardware (an additional divider) and, more importantly, these divider steps and subsequent comparisons reduce the PLL loop gain.

By contrast, directly using the 3rd harmonic of the 10 MHz reference to mix with the 30.2 MHz, the 200 kHz difference (ultimately 100 kHz - see below) may be used directly - and loop gain preserved, potentially improving PLL performance and simplifying the design.

The comparison with the reference frequency:

The 200 kHz "difference" signal from the harmonic mixer, filter and amplifier is applied to the divide-by-to circuit U2d to yield to yield a 100 kHz square wave.  The 100 kHz square wave from the divided-down 10 MHz reference signal and that from the 100 kHz "difference" signal are applied to U1c, an XOR gate, which is used as a phase detector.  As the phases of the 100 kHz from the reference signal and that of the difference signal "slide" past each other, the voltage - smoothed by R107 and C107 - will vary from 0 to 5 volts.  If, as an example, C201 in the 30.2 MHz crystal oscillator is adjusted so that 2.5 volts applied to the "VCXO Tune" line, this will cause the crystal oscillator to lock to the reference when the two signals are 90 degrees apart, being steered back onto frequency if they start to drift apart.  

I chose to use an XOR gate as a phase detector over a conventional phase/frequency detector because other than the desired DC component, the lowest-frequency component from its output cannot be lower than the comparison frequency - 100 kHz in this case, with the vast majority of the energy being 200 kHz and harmonics.  In comparison, many of the flip-flop phase/frequency detectors tend to output "occasional" pulses at very low frequency when at/near lock, which are nearly impossible to filter out.  There is a minor penalty, though:  An XOR gate phase detector requires use of 50% duty cycle square waves to work most efficiently, so each of its inputs is divided-by-two by a single 74HC74 dual flip-flop.

Figure 5:
Power connection to the original TCXO - L511 was removed.
Click on the image for a larger version.

Interfacing to the IC-910:

Switching the internal oscillator:

Bryan's IC-910 has the standard TCXO - X512 (the "CR-452") rather than the "High Stability" option (the "CR-293").  Either unit operates at 30.2 MHz, but there is a difference:  The standard TCXO operates from 5 volts while the high stability unit operates directly from the 13.8 volt supply.  Because the internal oscillator must be disabled when another source is applied, one will need to do one of two things, depending on how the radio is configured:

• Because this radio had the standard TCXO (CR-452 a.k.a. X512), inductor L511 (on the IC-910H's PLL board) was removed to make the power externally switchable and L510 and C501 (the "L510" on Figure 2, above) was connected to power X512.  It is this voltage that is switched by Q303 and regulated by U301 to provide switchable 5 volts.

• If the "High Stability" option ("CR-293") had been present (as described in the VK7MO case) we would have interrupted the 13.8 volt supply at C511/C512 (on the IC-910H PLL board) and switched it using Q303 directly rather than regulated to 5 volts by U301.  Comment:  It is unknown how much current the high stability oscillator consumes so a slight modification of the Q302 circuit might be required to do this.
  

Another difference between the way the two oscillators are interfaced appears to have something to do with the output level.  The standard TCXO outputs an RF signal of about 1.2 volts peak-to-peak while it can be seen from the IC-910H service manual that R515 is in series with the output of the high stability oscillator - presumably to reduce its level.

Injecting the locked 30.2 MHz signal:

Figure 6:
Connection of the 30.2 MHz to the PLL board showing
 the added D501 and L502.
Click on the image for a larger version.

Initially we simply connected the external 30.2 MHz in parallel with the output of the original TCXO, hoping that it would go "Hi-Z" when it was powered down - but that did not work:  When the original oscillator was powered down, its output was effectively shorted to ground, dropping the 30.2 MHz signal down to about 100 millivolts, so this signal was applied, instead, to the junction of variable resistor R570 and R572, using R570 to isolate it from the powered-down oscillator.  For this reason, diode D501 was implemented:  This diode - and L502 to provide a DC return - are connected directly at the junction of R570/R572:  When the external reference is activated, diode D501 is biased via R207, turning it on and connected the output of the 30.2 MHz VCXO to the IC-910H's PLL circuit.

If the external reference is not activated, Q203 - the VCXO power switch - is off and no voltage is applied to diode D501 via R207 and it remains "off", effectively isolating the original oscillator from the powered-down VCXO:  By placing the diode at the end of the coax, farthest from the external reference, there is minimal effect on the signal by that coax to the IC-910H's internal oscillator when the external reference is not being used.

Mechanical installation within the IC-910H:

Figure 7:
The back panel of the modified IC-910H.  The added BNC
connector is in the lower-left corner - the location of the
original ground screw, now relocated to the opposite corner.
Click on the image for a larger version.

Bryan had machined the box out of a chunk of aluminum back in 2010, sizing it to just fit (in all three dimensions) on the lid of the PLL unit.  As originally equipped, there are two brackets screwed down to the lid - apparently for the mounting of an optional voice synthesizer and DSP board - but these brackets were removed to make room.  Two SMA connectors were then mounted to the new box - one for the 10 MHz input and the other for the 30.2 MHz output, into the PLL board.  A pair of 1000pF feedthrough capacitors provide passage for the DC power into the box and the switched 5 volt output to the original TCXO on the PLL board.

Not shown (because I forgot to take the photo) is the connection to the switched 13.8 volt supply:  This was connected to the same point on the PLL board as depicted in the VK7MO document mentioned above - except, of course, that the trace did not need to be cut as would have been necessary to switch the power if the high-stability oscillator had been fitted.

The hole on the rear panel for the ground post was drilled out to permit mounting of a single-hole BNC connector with an already-fitted cable with attached SMA connector as can be seen in Figure 4.  This location for the BNC connector was slightly problematic as it somewhat blocked the screw to hold down the cover, but maneuvering of the connector, the use of tweezers and a small-diameter screwdriver permitted its installation.  In the opposite corner (the far-right in Figure 7) a new hole was drilled and tapped for the grounding post.

Spectral purity:

There was a small of concern that the spectral purity of the transceiver with the new reference oscillator would be worse than the original as I'd made no attempt to construct a very low noise oscillator (e.g. a lightly-loaded Butler or similar) so I compared the spectrum with both the internal oscillator and the "new", externally-locked oscillator on the various bands  - particularly on 23cm.

Figure 8:
Transmitter spectrum +/-500 kHz of a CW
carrier on 23cm as seen on an HP-8562A.
Click on the image for a larger version.

On 2 meters and 70cm, very weak (-70dBc) spurs at +/- 200 kHz - the main component of the output of the phase detector - were noted, barely above the broadband noise floor of the transmitter itself - but these were pretty much absent on 23cm as can be seen in Figure 8.  If these had been of concern, it would have been easy to further-improve the loop filter - which is currently a very simple R/C design as evidenced by Figure 2.  The fact that the plot in Figure 8 was made with the analyzer's resolution bandwidth set to 300 Hz should be an indication as to how low these 200 kHz components really are!

Another possible concern was closer-in phase noise:  Would various noise sources of the new circuits (VCXO phase noise, counter jitter, 1/F noise from regulators, loop noise, etc.) cause notable degradation?

Figure 9 gives a clue:  For this test, trace "A" is the original TCXO and trace "B" (the slightly fainter one corresponding with the peak on the right) was produced using the new, externally-locked reference.  As can be seen, the "close-in" phase noise performance of this radio isn't super great, anyway, but the two "noise humps" on either side of the carrier appear to be identical.

This trace also shows a slight difference in frequency, with the original TCXO (the peak on the left) being slightly low in frequency compared to the GPS-referenced, externally locked version - both showing identical amounts of phase noise indicating that the IC-910H is not degraded by this addition.

Figure 9:
A comparison of the close-in phase noise
using the original TCXO (left peak) and the
new, externally locked oscillator (right peak).
Click on the image for a larger version.

Conclusion:

Even thought it has been a long time in the making, this project is complete - and working as well as we hoped that it would.  I'm gratified that a mere decade ago, the circuit that I scribbled onto a piece of paper - and then built in one evening works just as it was expected, with no significant modification!

* * *

 P.S.  Alas, if you wanted to order a crystal, yourself, International Crystal Mfg. is no more, but custom crystals are still available via Quartslab - link, and Krystaly - link - to name but two places.

NOTE:  "Quartslab" stopped doing business in 2021 and "Klove" appears to have acquired the business - link.

(You would have to check with your chosen manufacturer to see if they will make a 30.2 MHz fundamental crystal, though - either that or modify the oscillator to use a 3rd overtone crystal.)

Comment:  Since this article was originally published, devices like the Leo Bodnar GPS reference link have become available that can produce the 30.2 MHz reference required by this radio directly.  This device uses GPS-based timing to set an oscillator to the desired frequency that is programmable - typically with accuracy in the range of 10E-10 or so:  In other words, it can directly synthesize the 30.2 MHz frequency needed by the IC-910H.

If you wish to generate a 30.2 MHz from a stable 10 MHz source that you already have, the Bodnar device really won't help you.

* * *

This page stolen from ka7oei.blogspot.com.

[End]

 

Using TV (F-connector) 75 ohm splitters and taps in 50 ohm systems on the amateur HF, VHF and UHF bands

I recently posted several articles about using commercially-available splitters link - and making one's own splitters - link - particularly for the HF frequencies and below (e.g. 30 MHz, down to a few 10s of kHz).  A comment was posted asking about how useful inexpensive 75 ohm "TV and satellite" type splitters might be for amateur radio use.

Figure 1:
The assortment of 75 ohm TV and satellite splitters and
taps tested in this article.
Click on the image for a larger version.

Implied by this question is the use of these devices in receive-only or small signal applications:  They cannot be used for transmit purposes as putting even 100 milliwatts through one of these devices is likely pushing its power-handling capability.

I've used these devices in 50 ohm circuits before - typically for VHF and UHF (2 meters, 70cm) where, along with some attenuators, combined the outputs of multiple signal generators to do "multi-tone" testing of receivers - but the question seemed to be a good one.  Rummaging around, I gathered a bunch of devices of various manufacturers and decided to test them for insertion loss and port-to-port isolation.

Note:

Please do not ask questions like "How well does a 'brand X' splitter work over the [fill in the blank] frequency range?"

There have been thousands of makes and models of these devices sold around the world over the past several decades and I simply am not able to find, locate, and measure more than the tiniest fraction of devices that have been sold.  The information given here is expected to be generally representative of the devices available from reputable manufacturers and distributors - but your mileage may vary.

Limitations of the measurements taken:

Because my VNA (DG6SAQ WVNA) was constructed for use with 50 ohm systems (the changing of  both internal hardware components and software would be required for "proper" analysis of a 75 ohm system) I was able only to analyze them in that context - but because the question was about using them in amateur radio service - which presumes a nominal 50 ohm system - I believe that the results are still useful within the limits noted in this article.

Because the emphasis of the question was interpreted as being for amateur-band frequencies likely to be encountered by the average user, the measurement range was limited to frequencies below 1 GHz - in some cases down to 100 kHz.  The nature of the equipment and methods (e.g. 50 ohm test equipment and cabling, the use of inter-series adapters, etc.) used to test the splitters and taps increasingly limits the usefulness and accuracy of these measurements at frequencies above that of the 70cm amateur band (above 450 MHz).

The variety of splitters and taps available:

There are literally thousands of brands and models of TV/Satellite splitters and taps available on this planet - some of them from recognizable names, but most not.  For those devices from sources that might be suspect (e.g. not "name" brands from reputable suppliers) you are on your own to determine the suitability of those devices for your purpose.

Although not intended as an endorsement per se, it has been observed that devices marketed by Holland Electronics appear to consistently meet their stated specifications and is one of the few brands that is likely available worldwide from a number or different sellers - including Amazon - and major suppliers of electronic components and TV/satellite supplies.

Over the years I have seen many dozens of brands and models of these devices - and the vast majority of them are what they are purported to be, but I have run across some devices that claimed to be splitters, but were simply a box with wires connecting the ports together.  In many cases, the casual user would not have noticed anything amiss, but using several of these faux devices in a larger system would certainly result in cumulative signal degradation (e.g. "ghosting" of analog signals, degrading of quality - but not necessarily signal strength - of digital signals).

General types of devices:

There seem to be three general types of these devices out there:

• "TV" and/or "VHF/FM/UHF" and/or "CATV" - These devices are typically designed to operate over the range of off-air TV stations across the world and the frequencies typically found on receive-only cable TV (with no Internet), encompassing the frequency range of about 40 MHz through 700 MHz, more or less.  While useful for use on the amateur bands from 6 meters through 70cm, inclusive, their usability on HF or above this range is limited as noted in the testing, below.

• "Satellite" splitters - These devices are typically designed to operate starting at about 900 MHz, often extending to 1500 or as high as 2500 MHz, depending on the vintage and intended use.  These devices are not usable on the 70cm amateur band frequencies and below.

• "TV/CATV/Satellite"- These devices are of a bit more recent vintage and are designed to accommodate a very wide range of frequencies - often from about 5 MHz through and above 2000 MHz - a band that includes off-air, cable and "L-Band" satellite signals - plus the "reverse" channels (sometimes called the "T" channels) often used by "cable Internet" modems that reside below 45 MHz.  These are the most useful to amateur service and can often be used on HF through 70cm.

If you do not see a specific frequency range noted on the device itself, assume the worst-case, smallest frequency range that covers that usage category - unless you can test them yourself.

* * *

General findings

For the TL;DR types, here is a summary of the results of the measurements described in more detail farther down the page.


Using 75 ohm devices in 50 ohm systems:

The most obvious issue is that TV-type consumer devices are almost universally equipped with type "F" connectors which means that one must use either an adapter or use a cable with an attached "F" connector.

Figure 2:
Left to right:  Two BNC female to male F connecitrs with an
F-type 75 ohm terminator on the right.
Click on the image for a larger version.

For receive-only systems, it's not too uncommon to simply use a 75 ohm cable like RG-6 - which is quite low loss and very inexpensive - to connect a 50 ohm antenna to a 50 ohm receiver.  The effects of this apparent "mismatch" are typically minimal as most receivers are only "approximately" 50 ohms, anyway.  In theory, the use of 75 ohm cable on 50 ohm devices will result in a 1.5:1 mismatch and commensurate losses, but this sort of mismatch is commonly observed on many antenna systems that are ostensibly designed to operate at 50 ohms and is usually of minor consequence.

When using an inexpensive cable like RG-6, it's worth noting that most of these cables use copper-coated steel (CCS) center conductors which may have implications for DC resistance of power is being sent on this cable (for a preamplifier, converter, controls) as this type of cable will have far more total resistance than one with a solid copper center conductor.  Copper-coated steel center conductors may also have implications in terms of skin effect at low frequencies (low HF and below) - but this is beyond the scope of this article - see, instead, this article by Owen Duffy from the Internet Archive.  There exist cables with copper-coated aluminum (CCA) center conductors that have lower DC resistance that CCS cables, but they tend to be more fragile due to the tendency of the aluminum center conductor to become brittle with flexure.

The device itself (splitter, tap) is designed primarily for 75 ohms and this means that its performance will be somewhat degraded in a system that is "completely" 50 ohms (e.g. 50 ohm cables with F-connector adapters) but these effects are largely as follows:

• The "through" loss may be slightly higher.  In the case of a 2-way splitter, the ideal loss will be 3dB - but even at the proper impedance, it will be slightly higher than this due to component losses, typically in the area of 3.5 dB.  Practically speaking, the main effect of using a 75 ohm splitter in a 50 ohm system was a slight change (only a few tenths of a dB) in the loss.

• Reduced isolation between ports.  The most obvious effect on splitters was that the isolation between ports (e.g. the "out" ports of a 2-way splitter) was reduced.  Compared to some specialized splitters, the isolation of inexpensive, consumer-grade "TV" splitters is lower overall.  As can be seen from the graphs, below, operating in a 75 ohm system resulted in better isolation - sometimes over 40dB at certain frequencies - but this assumes that all loads and sources are well-matched to 75 ohms, something that is not likely to be the case in a real-world installation.  Typically, isolation reduced to something in the 20dB area when operated in a 50 ohm system.  In many cases, this is "good enough".

• In splitters and taps, resistors are major components in determining their "native" operating impedance.  For example, a 75 ohm splitter or tap, depending on design, may have a 150 ohm or 37.5 ohm (2 times and one-half 75 ohms, respectively) resistor contained internally.  In theory, changing this resistor to a value appropriate for 50 ohms (typically 100 or 25 ohms) would optimize performance at 50 ohms - but doing this may or may not be worth the trouble. 

In short:

Unless your situation requires precision, the use of inexpensive, TV-type splitters and taps of the types described on this page will yield "reasonable" performance over the design frequency range - provided that the device is constructed as described by a reputable manufacturer.

The use of a (nominally) 75 ohm device in a 50 ohm system will require using connectors that are not normally used in 50 ohms systems (typically "F" connectors) which means that adapters of some sort will be needed - the expense, bulk and inconvenience of which must be considered in the overall design.

Finally, note that the above comments are for the general case:  Remember that your needs, requirements and results may vary and that you must do your own analysis and testing to verify that such components are appropriate in your specific case.

* * *

Plots of various devices:

Below are selected plots of devices representative of the types on-hand.  In general, devices with similar stated ratings performed in the same manner.  In all of these plots, the insertion loss is represented by the blue line while the complex impedance data is depicted on a Smith chart in the middle:  Numerical data at the frequencies indicated by markers is seen in the lower-left corner of the screen.  Again, remember that at higher frequencies, the nature of the 50 ohm test system, connecting cables and adapters will increasingly skew the results - particularly those depicted by the Smith chart.

The interpretation of a Smith chart will not be covered here, but there are many online resources that describe its use including this video in a series on this topic by W2AEW on his YouTube page.

A "satellite" splitter:

Figure 3:
The "through" loss of the HFS-2 splitter represented by the blue line
across the top.
Click on the image for a larger version.

This device - a "Tru Spec HFS-2" is representative of those intended for use on an (older) L-band system found in satellite receive systems, having on its label a "900-1500" MHz frequency range.  As noted above, the limitation of the measurement set-up made measurements above the 70cm amateur band (in the 440 MHz area) suspect - but the object here was to see if it was usable below that range.

At initial glance, the "through loss" of this device below 900 MHz (Figure 3) might seem to indicate that it worked below this frequency, but notice that at lower frequencies (below 50 MHz) indicates a loss less than 3dB indicating that it is not working as a proper 2-way splitter.  A look at the isolation plot (Figure 4) tells more of the story.

Figure 4:
Isolation between ports of this splitter.
Click on the image for a larger version.

As can be seen, at about 900 MHz and above, the apparent isolation between ports is reasonable but at 2 meters (146 MHz) it is only 3dB verifying the fact that at these lower frequencies, it less a proper splitter, but more equivalent to a device where the three ports are connected with a piece of wire.  The apparent isolation increase at low HF is more likely an artifact of its construction - the insertion loss being below 1 dB (in Figure 3) verifies this.

In short, these "Satellite only" splitters aren't really useful on TV and CATV frequencies or the amateur bands 70cm and below.

A "TV" splitter:

Figure 5:
The "through" loss of the Archer splitter.
Click on the image for a larger version.

I tested several splitters that were intended for general VHF/UHF/FM use - one of these being an "Archer" (Radio Shack) two-way splitter being typical of that type.  The implied frequency range is from at least 54 MHz to 700 MHz - the extent of the cable TV, FM broadcast, and off-air VHF and UHF TV frequencies at the time it was made.

Figure 5 shows the measured "through" loss in a 50 ohm system.  Compared to a plot done at 75 ohms (not shown, using resistive matching) the insertion loss barely changes across the frequency range.  In both 75 and 50 ohm systems, at least at 2 meters, down to 20 meters (14 MHz) seems to be "ok" - but the "dip" in the 3-4 MHz area - and the fact that the attenuation below it drops below 3dB - indicates that it's not likely acting like a splitter at these lower frequencies.

Figure 6:
Port to port isolation at 75 ohms for this splitter.
Click on the image for a larger version.

Figure 6 shows the port-to-port isolation at 75 ohms and we note that in the "low" and "high" VHF band (U.S. channels 2-13 - which more or less includes the 6, 2 and U.S. 222 MHz amateur bands, that the isolation is quite decent - well above 20 dB.

From this plot we can see that the "dip" in the 3-4 MHz area seen on Figure 5 is quite telling as the port-to-port isolation is pretty much gone below this frequency

Figure 7:
Port to port isolation in a 50 ohm system for this splitter.
Click on the image for a larger version.

The plot of Figure 7 shows what happens if the splitter is operated in a 50 ohm system.  The main effect is that the port-to-port isolation is reduced - being on the order of 15 dB or so from the 20 meter band through the 2 meter band (14 MHz - 144 MHz).

From this we can conclude that this splitter is quite usable from the middle of the HF spectrum through at least 2 meters - and is probably usable through 70cm.


A "TV/CATV/Satellite" splitter - preferred for HF use:

Figure 8:
Holland HFS-2P through loss in a 50 ohm system.
Click on the image for a larger version.

I have on hand several splitters that have on their label a frequency range that starts at (typically) 5 MHz with a high end of between 600 MHz and 2450 MHz.  The reason for this extended "low end" is likely due to their being designed for use in systems that have "Cable Internet" where the return (upstream) signal from the user's modem to the cable system are likely to be in the 5-50 MHz (or, possibly, a bit higher) range.  The plots included are those of a Holland Electronics HFS-2P which is a 2-way splitter/combiner that has a stated range of 5-2050 MHz and the results of this device are typical of that type.)

Figure 8 shows the "through" loss in a 50 ohm system showing a reasonable insertion loss (4 dB or below) from below 40 meters (about 5 MHz) through at least 70cm (440 MHz) - but again, the limitations of the measurement set-up make readings higher than this a bit suspect.

Figure 9:
Port-to-port isolation at 50 ohms.
Click on the image for a larger version.

Again knowing that the "isolation" measurement is the way to get the "true" story, port-to-port isolation in a 50 ohm system is depicted in Figure 9.

This verifies - to the extent that the test set-up can - the 5-2050 MHz range showing that the port-to-port isolation from 5 MHz to 1 Ghz is well over 15dB.  A port-to-port isolation measurement at 75 ohms (not shown) is slightly better (by a few dB) over the same range.

The combination of Figure 8 and Figure 9 show that this device may be usable down to the 160 meter band (1.8 MHz) provided that a slight amount of extra insertion loss (about 1dB) and lower isolation (approximately 12dB) can be tolerated.    (The Holland HFS-2D has characteristics similar to the HFS-2P down to 1.8 MHz.)

Figure 10:
The through loss, the other 7 ports being terminated with 75 ohm F-type
connectors.  The insertion loss is reasonable - between 10.5 and
11.5 dB over the range of 1.8 to 450 MHz.
Click on the image for a larger version.

An 8-way splitter:

The final splitter to be tested was the Holland Electronics GHS-8 8-way splitter-combiner.  Often, splitters with an even number of outputs greater than two contain multiple two-way splitters which means that this 8-way splitter might contain seven such devices - but I didn't break it open to check.

Figure 11:
The port-to-port isolation between two adjacent ports with the "in/out"
port and unused ports terminated with 75 ohm "F" loads.  The apparent
isolation is on the order of 35dB from 1.8 through 450 MHz - but this would
likely drop to something closer to 20dB.
Click on the image for a larger version.

Figure 10 shows the typical "through" insertion loss with the seven unused ports being terminated with 75 ohm "F" type terminators:  I don't have enough F-male to BNC-female adapters on-hand to terminate the 7 ports at 50 ohms - but if one were going to use one of these devices, it's probably more convenient to use F-type terminators on the unused ports, anyway.  The typical "through" loss is measured to be about 10.5-11.5 dB - slightly higher than the predicted "ideal" 9dB insertion loss, but typical for these devices.

The port-to-port isolation was also measured and the use of 75 ohm terminations on the other ports and the "common" in/out port likely improved this:  The isolation would likely be significantly worse if all ports were at 50 ohms, for the same reason as the other splitters tested.

Based on these readings, this device is useful down to 1.8 MHz and up through 2 meters - and probably 70cm.

Figure 12:
Coupling coefficient at 50 ohms for this tap
Click on the image for a larger version.

A TV-type signal "tap":

Likely unfamiliar to many, a signal "tap" is a very useful device in multi-drop TV installations found in hotels, hospitals and other larger buildings.  Unlike a splitter - which usually divides a signal equally to its output ports - a "tap" will siphon only a certain amount of signal off the cable and leave the majority of it intact - which is very useful for systems such as those in a hotel or hospital to distribute and split a signal hundreds of times to serve all of the devices.

In some ways it can be considered to be similar to a part of an SWR bridge where only a small amount of signal is sampled - and in only one direction - allowing the majority of the original signal to pass with minimal loss.  Several taps - all from Holland Electronics - were tested as they were what was on-hand and the "DCG-6SB" is represented in the plots. 

Figure 13:
The "reverse isolation" loss of the tap (e.g. turned "backwards") with a
50 ohm termination.
The reverse isolation is described as being the absolute amount of isolation
(e.g. that in the chart above) minus the coupling coefficient which means
that the actual forward coupling loss - which means that using Figure
12, we know that the actual reverse isolation is about 7 dB lower than
indicated by the graph above.
Click on the image for a larger version.

Figure 12 shows the "coupled" energy in a 50 ohm system:  Compared to the coupling in 75 ohm system (now shown) the insertion loss was slightly higher (about 1dB) but the frequency loss/flatness was about the same, being pretty consistent from about 1.8 MHz through 1 GHz.

Figure 13 shows the reverse isolation of the tap:  Rather than 6dB of coupling from the main line for signals going the "other way", the absolute is closer to 20dB - about 13dB lower.  (The actual reverse isolation is the absolute isolation minus the forward loss).  In a 75 ohm system (not shown) the reverse isolation was quite a bit better (closer to 30dB over the 5 MHz-1GHz range) - but this result is completely expected:  The reverse isolation is akin to measuring VSWR, and operating a 75 ohm device at 50 ohms implies a VSWR of 1.5:1 - a "return loss" of 14dB - very close to the values depicted in Figure 13 over much of the frequency range when the "forward" loss is taken into account.

On a tap there is yet another measurement to be taken - the loss between the in and out port.  Because we are measuring a 6dB tap - a device which siphons off about 25% of the signal - we would expect at least that amount (theoretically 1.25dB for 6dB) to be lost as it is coupled to the "tap" port. Figure 14 we can see that the measured loss is slightly higher than this between 1.8 and 200 MHz- a bit over 2dB.  Some of this "extra" loss is due to the intrinsic losses of the device, but a smaller amount is a result of the use of a 75 ohm device on a 50 ohm system.

Figure 14:
Through loss of the 6dB tap in a 50 ohm system.
Click on the image for a larger version.

This device - which is rated down to 5 MHz - may be useful through at 160 meters (1.8 MHz) - but the insertion loss goes up rather quickly at lower frequencies.

This device is NOT suitable for passing DC (e.g. for amplifiers, control signals) as it has a DC short across it - but that is not true of all taps.  For example, the Holland Electronics "HDCS" series does allow low frequency RF down to DC to flow through it - but like the DCG-6SB, its coupling coefficient deteriorates quickly below about 1.8 MHz.

* * *

General conclusions:

If you are going to use TV-type splitters for HF, make sure that you get devices that are explicitly rated down to 5 MHz.  Based on the (limited!) sample of devices that were tested, these devices can be expected to work into the 160 meter amateur band (down to 1.8 MHz).  While these devices may be usable thoughout the entire AM broadcast band (down to 540 kHz) expect performance to drop quickly in terms of added "through" attenuation and worse port-to-port isolation.

A "TV" type device - one that may indicate a start frequency of 5 MHz, or just any device that is claimed to work at TV (VHF/UHF) and FM broadcast frequencies will likely work from 6 meters through 70cm (50 MHz - 450 MHz).

Again, for general signal splitting and combining, these 75 ohm devices, used at 50 ohms, are quite usable for non-critical applications - provided that they be used at low power levels (a few 10s of milliwatts at most) and where one need not have precise 50 ohm matching and high port-to-port isolation.  Remember that most 50 ohm devices (receivers, amplifiers, filters) have only "approximately" 50 ohm source/load impedances - and filters in particular will, out of their design frequency range (outside the band-pass, on a notch frequency, above the low-poss cut-off, below the high-pass cut-off) will likely have anything but a 50 ohm characteristic impedance, so even a "proper" 50 ohm splitter/tap device would not necessarily yield any better performance in those situations.

For information about the design and use of splitters/combiners in general, a good reference is Mini-Circuits AN10-006, "Understanding Power Splitters" - link.

* * *

Far more data was gathered than was presented here, but that depicted above is representative of the devices that were on hand.

* * *

This page stolen from ka7oei.blogspot.com

[End]

When Band-Pass/Band-Reject (Bp/Br) duplexers really aren't band-pass

Addendum: 

Since this article was originally published, the availability of low-cost test equipment like the NanoVNA has allowed more-thorough testing of their RF signal paths.

With the advent of inexpensive and "good" test equipment like a NanoVNA, there is little excuse these days for not knowing if one's duplexer has the needed band-pass characteristics described in this article.

* * * * * *

In the repeater world there is a misconception that just because the duplexer may say "Band Pass, Band Reject" on its label - or even in its 'spec sheet - that it really does offer a proper band-pass response over a wide range of frequencies - but this is usually NOT the case.

A close-in look at a typical Band-Pass/Band-Reject duplexer:

Take Figure 1, below, as an example.

Figure 1:
The magenta trace is that of a proper band-pass cavity, the yellow trace is that of a one side (3 cavities) of a 6-cavity Phelps-Dodge "Band-Pass/Band-Reject" duplexer while the cyan trace is the combination of the two.  The top of the yellow peak (with the "1" marker) represents the center frequency of the duplexer with a bit over 1dB loss.

In the analyzer trace above, the YELLOW is the response of one of half of a typical amateur "Band-Pass/Band-Reject" 6-can Phelps-Dodge duplexer tuned in the 2-meter amateur band and from this trace we can see several things happening:

• As there should be, there is a peak at the pass frequency corresponding to the "band-pass" of the duplexer - in this case, a bit over 1dB loss.
• Just above the peak - 600 kHz, to be precise - is a very deep notch - corresponding to the frequency to "band-reject" part of the name.  In reality, the depth of the notch depicted in Figure 1 is about 100dB, but the true depth is not apparent from the trace.
• Once one moves about 1 division (1.5 MHz) either side of the peak/notch frequency, the attenuation isn't that great - only about 20-30dB, and the trace above the center seems to be on a asymptotic trajectory upwards (lower attenuation) as frequency increases.

From the above we can see that while this duplexer offers a "Band-Pass/Band-Reject" response, this occurs only at frequencies very near the input/output frequencies of our hypothetical repeater.  Once you get "farther away", this "band-pass" response diminishes.

On the other hand the MAGENTA trace shows a single band-pass cavity filter.  While its attenuation is not very high at the notch frequency - on the order of 10-15dB - it is apparent that by 2 MHz above the center frequency it is offering greater attenuation than the so-called "Band-Pass/Band-Reject" filter and that below the center, the trend indicates that they might cross over at a point just to the left of the trace.

Comment:  This "Bp/Br" nomenclature is widely applied amongst many manufacturers to duplexers that have the same response as the Phelps-Dodge duplexer above, including Motorola and Wacom - to name but a few.  It is the rare exception to find a "Band-pass/Band-Reject" duplexer that does NOT exhibit the properties described on this page!

Unless you have already installed some band-pass cavities on each leg of your duplexer and/or have done proper sweep responses at frequencies far removed from the designed frequency, you should not assume that your "Bp/Br" duplexer is truly Band-Pass/Band-Reject over a very wide frequency range!

Taking a wider view:

Figure 1 only spans about 7.5 MHz on either side of 2 meters, so let us widen it a bit as shown in Figure 2, below:

Figure 2:
Spanning from 30 MHz to 1 GHz, the same cavities/filters as noted above.  Again, the yellow trace is one half of a 6-cavity "Band-Pass/Band-Reject" duplexer, the magenta trace is the pass cavity alone and the cyan trace is the result of the bandpass cavity and the Bp/Br duplexer cascaded.  It should be noted that at odd-numbered harmonics the pass cavity
will present a narrow bandpass response that can be eliminated with the addition of a simple low-pass filter.

When looking over a much wider frequency range - 30 MHz to 1 GHz - the picture is quite different.  Based on this sweep we can see that our typical "6 can" duplexer - of which 3 "cans" of the transmit or receive side - are represented above in YELLOW and that for the majority of the frequency range there is relatively little attenuation offered overall!  Paying particular attention we see that the attenuation in much of the VHF-low TV band (channels 2-6) and the FM broadcast band is quite poor - on the order of 3-10dB - as is the case over much of the VHF-high (channels 7-13) and large sections of the UHF TV band.

What we can see from this picture is that if we rely on only our so-called "band-pass/band-reject" duplexer on a site with other services such as FM or TV broadcast, or even land-mobile, those frequencies just above the amateur band, such a duplexer offers relatively little protection against those signals getting into the transmitter or receiver.

Why it matters:

One might wonder why it would matter whether or not a duplexer offered good "far-off-frequency" rejection.

In many cases, particularly in mountainous areas, amateur repeaters are co-located at sites with other transmitters and if adequate filtering is not implemented those "other" signals can get into the repeater's receiver and/or transmitter.

The effects of these other signals' ingress into the receiver is easier to envision:  Many of us have observed that, while driving about, our mobile radios have occasionally been overloaded with other signals - the effect being that we are hearing signals on frequencies where they are not.  This phenomenon is the inevitable result of the receiver's mixer - a device that is designed specifically to make new signals out of multiple signals in the first place - synthesizing entirely new ones out of the several that get in via its antenna.

Several decades ago it was common for land-mobile VHF and UHF radios to have receivers that had very tight filtering as there were typically only a few, closely-spaced channels that were used.  By virtue of this extensive filtering it was unlikely that other signals' somewhat-removed frequencies could even get in and cause undesired signals to be generated.  These days most radios have very broad filtering in their receiver inputs - this, to allow a wide range of frequencies to be accommodated.  While convenient, this also has a down side:  Those formerly widely-spaced frequencies from other services now have little impediment and it is more likely that they will get into the receiver and produce undesired, spurious signals.

Many years ago it was also the case that many repeaters used modified land-mobile radios with their extensive filtering, but nowadays many "store bought" repeaters (such as the Icom D-Star and Yaesu Fusion lines) are simply beefed-up mobile radios with "broad as the proverbial barn door" filtering on their receivers.  While this is convenient for the repeater owner to not have to dig up some test equipment and tune up these receivers' narrow filters, this also means is that there are many instances where a club has replaced their old, crystal-controlled analog repeater with a new one - only to find out that it did not work well at all when these off-frequency signals - formerly blocked by the old receiver's narrow front-end filter - clobbered the new receiver.  Worse still, some of these repeaters (namely the Icom D-Star) provided no analog test points where the receiver performance could be directly analyzed to determine if there was a problem, much less its extent!

What's worse is it is often the case that at many sites this sort of interference may be intermittent in nature - occurring only when a certain combination of transmitters happened to be online at once:  With most repeaters using subaudible tones for access, this degradation is often masked since the repeater may stay silent when it is being impacted, the only clue being that some users may suddenly find it difficult to get in to the repeater with a good signal at random times.  In other words, unless one uses the proper test equipment to take and record repeatable measurements at or away from the site, gradual or occasional degradation of the receiver's performance may not be so apparent.

An insidious problem:

While the overloading of a receiver is a familiar problem to many of us, it may not be as obvious that a similar thing can happen in a transmitter.  Like a receiver, a transmitter has the ability to take two signals and produce others via mixing.  For this to happen it usually requires that the "other" signals are very strong - but this is something that can happen at a busy radio site!

As a demonstration of what can happen, it was noted that via a VHF antenna atop Farnsworth Peak near Salt Lake City, Utah - a very busy broadcast site - one could read 100-150 milliwatts of RF on the coaxial cable at the input to the duplexer.  When this energy was analyzed it was found to be a combination of FM broadcast and UHF TV signals - the same transmitters that produce several megawatts of effective radiated power, combined.  If the same 6-cavity duplexer depicted in Figure 1 and Figure 2 was inserted in the line, this power would reduced - but only to the 20-50 milliwatt level!

This power was measured on the feedline of what would be a D-Star repeater, but prior to the installation of that repeater an analog Kenwood TK-740 repeater had been used for several months to assess coverage and performance prior to the installation of the Icom D-Star repeater.

On the day that the D-Star repeater was installed it was discovered that no-one could get into it, despite their running 50 watts.  Upon analysis it was discovered that the 20-50 milliwatts coming back into the coax was causing the Icom repeater's receiver to be deafened (desensed) by about 40dB - a factor of 10,000-fold!  Upon reconnecting the TK-740, no problems were noted and it was realized that the Kenwood repeater had a more traditional, narrow-band helical resonator filter assembly in its front end and compared to the more modern "broad-band" front end of the Icom repeater - which used parts of modified mobile radios - that the power in from the antenna was completely demolishing its receiver!

Figure 3:
A typical Motorola  4-can duplexer for UHF.  Just like its VHF counterparts
it easily passes energy at frequencies above and below its tuned frequency.
Click on the image for a larger version.

The installation of two bandpass cavities on the receive side allowed the Icom repeater to work as well as the old Kenwood analog repeater with its superior filtering - but this brings up the question about what might happen on transmit?

The transmitter can also act as a mixer:  Multiple signals - one of which might be the repeater's output frequency - can combine within the circuitry and instead of only the transmit frequency being emitted, some conglomeration of signals can appear!

In the example of a VHF transmitter we know that while the low-pass filter may remove the frequencies above the 2-meter band - say, UHF land-mobile and UHF TV - it will do nothing to remove energy from FM broadcast stations.  Similarly, if this were a UHF transmitter, its low-pass filter might remove some of the UHF land-mobile and UHF TV energy, but it would have little effect on signals from FM broadcast and VHF high and low band TV.

It might be suggested at this point that the use of an isolator - a device that, while allowing the transmitter's energy to go to the antenna, it directs any power coming back down the coax into a dummy load so that it cannot even get to the transmitter, might be appropriate here - and this would be correct...  Mostly.  While these devices are invaluable - and even required equipment at many radio sites - to both prevent RF from nearby-frequency transmitters from getting into your transmitter - and then re-radiated again and also to insulate your transmitter from a bad VSWR - it is far less-effective when the frequencies that are coming back down the coaxial cable are away from its design frequency.  In other words, while your VHF isolator may work okay from 140 to 160 MHz, it will probably do comparatively little at the FM broadcast band and in the UHF range.

Adding a pass cavity:

It is, therefore, a very good idea to equip any repeater with at least two pass cavities:  One on the receiver, tuned to the input frequency and another on the transmitter, after the isolator, tuned to the output frequency.

If one examines both Figures 1 and 2 you can see the Magenta trace showing the response of a single pass cavity.  When compared to the response of a typical Bp/Br duplexer (the YELLOW trace) the general trend is that the farther away one gets from the pass frequency, the more attenuation it offers.  One quirk of band-pass cavities is that they also have a response at odd multiples of their pass frequency, which means that a 2-meter pass cavity will also pass energy around the low end of 70cm, around 700 MHz, and so-on.  In the case of 2 meters, this spurious response could be eliminated by the addition of a low-pass filter.

Both figures 1 and 2 also show something else:  What happens if you cascade a Bp/Br duplexer with a single pass cavity (the CYAN trace)?  For the most part the overall attenuation of the two sets of filters is complementary - that is, the "best of both worlds."  As can be seen the simple addition of a pass cavity knocks out almost everything that is off-frequency from that which is desired.

Figure 4:
A typical "4 can" (2 on transmit, 2 on receive)
2-meter duplexer.  Even though it is labeled
as a "band-pass/band reject" unit, this refers only
to the two frequencies of interest - the transmit
and receive - and not to the RF spectrum overall!
The plots in Figures 1 and 2 are from a similar,
"6-can" (3 on tx, 3 on rx) duplexer, but the
rejection of frequencies "far removed" from
where they are tuned is comparable.
Click on the image for a larger version.

Bandpass cavities have another important property as well:  Lightning protection.  Because lightning is a broad-band energy spike, it would make sense that if you reduce the passband of the signal path from the antenna, less RF energy, overall, will get in - and the use of a passband cavity also guarantees that there is NO DC path from the center pin of the coax from the antenna to the center pin of the coax going to the radio.  One radio club - the Utah Amateur Radio Club - has several mountain top repeaters and there have been a number of instances where the repeater antenna has taken a direct lightning hit, sometimes destroying the antenna, but never has the attached receiver or transmitter ever been damaged.

Summation:

If you are installing a repeater or other radio at a site with any other transmitters you should not assume that just because the label or specifications of the duplexer say that it is "Band-Pass/Band-Reject" that it will actually do so over a wide range of frequencies.  Again, most brands of duplexers will simply pass, with relatively little attenuation, those frequencies that are far removed from the operating frequencies and the "band-pass/band-reject" nature is limited to the specific frequencies of interest - such as the transmit side of the duplexer passing the transmit signal but rejecting energy at the receive frequency.

Such a duplexer should always be supplemented with at least one bandpass cavity for the transmit frequency and another for the receive frequency to provide additional off-frequency rejection - and adding a simple low-pass filter on each leg won't hurt, either.  While these added elements result in higher signal loss, this need only be 1dB or less in most cases.  Adding this extra cavity will increase the effectiveness of an isolator on the transmitter - which works only well near its design frequency anyway - but it will also prevent excess, off-frequency energy from getting into the repeater's receiver which, these days, is more typically a "mobile" unit with a very broad front end that has been converted.  Finally, the humble band-pass cavity provides good lightning protection, just by its very nature!

[End]

This page stolen from ka7oei.blogspot.com



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