As noted in the previous post(s), the problem is two-fold when it comes to broad-band SDRs that are intended to cover the entire HF spectrum all at once:
- HF noise power and signal level is (generally) inversely proportional to frequency. At lower frequencies - say, 2-8 MHz - the noise power is far higher than it typically is at around 20-30 MHz.
- A direct-sampling SDR - or any receiver, for that matter - can tolerate only so much RF power on its front end. Traditionally, this is a mitigated with the use of narrow-band RF band-pass filters, but this can't be done if one intends to be able to cover the amateur radio bands 160 through 10 meters (1.8-30 MHz).
- To accommodate the very strong signals and high noise levels at lower HF frequencies, the RF signal gain in front of the A/D converter must be carefully set to prevent overload.
- In order to "hear" the noise floor at 10 meters, the system gain must be set fairly high.
What these two factors, together, imply is that if we have enough gain to comfortably detect the noise floor at 10 meters, our receiver will be badly overloaded during strong-signal conditions on the lower bands. Conversely, if we scale (e.g. attenuate) the input to accommodate the very large signal excursions, the receiver will simply be unable to detect signals at/near the "quiet" 10 meter noise floor.
There will (hopefully) be the day that the upper HF propagation conditions improve greatly with the arrival of solar cycle 25 and at that time, strong signals will appear on the bands >=15 MHz. When this happens, we will likely be faced with a problem similar to that which we are trying to solve here (e.g. very strong signals overloading the A/D converter). At this time, the only recourse will likely be a means of using an external device to adjust the gain/attenuation in front of the receiver, probably using the existing I/O lines under receiver control.
A revised circuit:
Why talk about this issue a THIRD time? I decided to make one that provided a better 50 ohm match across all frequencies than the previous versions. This revised circuit may be seen in the figure below:
Generic pre-emphasis network set for about 50 ohms.
Click on the image for a slightly larger version.
Some readers will recognize the topology of the circuit in Figure 1 as the classic pre-emphasis network found in the signal path of FM video transmitters. Whereas those circuits are typically designed for 75 ohms, this one is intended for a 50 ohm system - but careful observers will notice that 47 ohm resistors are used, instead: For receive-only purposes, I have chosen the components in this article to be standard values at the expense of a slight increase in mismatch - but the VSWR of these circuits, when terminated at 50 ohms - is likely to be no more than about 1.1:1.
This circuit - compared with the previous versions - has the advantage that it presents a consistent source and load impedance across the frequency range, making it a bit more "friendly" in systems that may be impedance sensitive (e.g. following a band-pass filter, long coaxial cable runs, following/preceding conditionally-stable RF amplifiers.) The obvious trade-off is that as compared to the previous version (which was based on a high-pass filter and some resistive bypassing) this circuit has definite limitations on how sharp and deep the "knee" may be at any given frequency as only a single inductor and capacitor are used.
By tweaking the values of R1, R4, C1 and L1 we can adjust both the amount of low-frequency attenuation and the frequency of the "knee" where the attenuation takes place - but for our purposes, we will be placing the center of that "knee" around 10 MHz to provide both the minimal loss at 30 MHz and adequate attenuation at and below 7 MHz.
Here are a few examples of values of R1, R4, C1 and L1 using standard-value components and approximate attenuation values at various frequencies:
|R1 = 68 ohms R4 = 39 ohms
C1 = 390pF L1 = 1uH
DC attenuation: 7.3dB
@ 2 MHz: 7.0dB @4 MHz: 6dB
@ 7 MHz: 4.6dB @10 MHz: 3.4dB
@ 14 MHz: 2.3dB @28 MHz: 0.8dB
|R1 = 120 ohms R4 = 20 ohms
C1 = 330pF L1 = 0.82uH
DC attenuation: 10.8dB
@ 2 MHz: 9.8dB @4 MHz: 8.1dB
@ 7 MHz: 5.6dB @10 MHz: 3.9dB
@ 14 MHz: 2.5dB @28 MHz: 0.8dB
|R1 = 120 ohms R4 = 20 ohms
C1 = 270pF L1 = 0.68uH
DC attenuation: 10.8dB
@ 2 MHz: 10.1dB @4 MHz: 8.7dB
@ 7 MHz: 6.5dB @10 MHz: 4.8dB
@ 14 MHz: 3.3dB @28 MHz: 1.2dB
|R1 = 100 ohms R4 = 27 ohms
C1 = 270pF L1 = 0.68uH
DC attenuation: 9.4dB
@ 2 MHz: 8.9dB @4 MHz: 8dB
@ 7 MHz: 6.3dB @10 MHz: 4.8dB
@ 14 MHz: 3.6dB @28 MHz:1.3dB
Table showing some possible values for the circuit of Figure 1 and the example attenuation values.
In practice, several of these sections will likely need to be cascaded to achieve the desired amount of attenuation at the lower HF frequencies which brings up the question: Could you not choose components to do this for a single section? The answer is theoretically, yes - but the fact is that practical inductors - particularly the molded type - are quite lossy, departing from the intended attenuation curve, and achieving the predicted, higher amount of lower-frequency attenuation with a single stage can become problematic - so it's probably better to cascade several of these networks together, instead.
A practical example:
The exterior of the four channel filter network.
Click on the image for a larger version
A practical example of such a network is one that is
to be currently installed in the KFS (Half Moon Bay, CA)
KiwiSDR/WSPRDaemon system. There, four wideband antennas are available
to feed the KiwiSDRs on site, so a box was constructed with four,
identical pre-emphasis networks, each to feed its own receiver stack.
As is the case at the Northern Utah WebSDR, noise and signals at the lower end of the HF spectrum is often very much stronger than at the high end: If amplification is added to allow the detection of the noise floor at 10 meters, there is the very high probability that the receiver will badly overload on HF signals from the lower end of the spectrum.
Each "channel" of the device depicted in Figure 3 is identical, consisting of two cascaded sections. The first section is that from the upper-left quadrant of the table (R1=68 ohms, C1 = 390 pf) and the upper-right quadrant (R1=120 ohms, C1 = 330pF). Rather than the use of molded chokes, the individual inductors were wound using 30 AWG wire on T25-2 toroids: 17 and 15 turns for the 1 uH and 0.82 uH inductors, respectively.
The interior of the four-channel network.
The circuit is simple enough to be wired "Manhattan"
style on glass-epoxy PC board material between the
two center pins of the BNC connectors.
Click on the image for a larger version
As can be seen in Figure 4, the construction is very simple, requiring no circuit board at all when using standard, through-hole components. The circuit was built into a die-cast aluminum box with the BNC connectors holding the piece of PCB material in place.
To secure the components - particularly the small, toroidal inductors - RTV sealant (white) was used to hold components in place and to prevent adjacent wires of C1/R1 and R2/R3 from coming into contact with each other.
This method of construction is very simple and effective, offering good performance into the VHF range when reasonable care is taken. With the 20mm high dividers between the sections installed as shown, the channel-to-channel isolation exceeded 85dB (the limit of convenient measurement) at 30 MHz.
Figure 5, below, shows the typical response of the sections:
The response of one of the sections as measured on a DG8SAQ VNA.
Click on the image for a larger version.
Because it can be a bit difficult to read, the values of attenuation and VSWR in the upper-left corner are reproduced below:
|Frequency (MHz)||Insertion Loss (db)||VSWR|
Attenuation and VSWR of the network at amateur band frequencies.
For specifics relating to a wideband direct-sampling SDR like the KiwiSDR or Red Pitaya, refer to the earlier article linked above - "A Limited Attenuation High Pass Filter".
- A good test is to see if, on 10 meters when it is "dead", you are hearing your local noise floor. Note the S-meter with the antenna connected and disconnected - preferably, with the input to the receive system terminated with a 50 ohm load when disconnected. If you do not see an increase in the S-meter reading and on the waterfall by 3-5 dB, the overall system gain is too low to allow the receiver to see the noise floor at your antenna system.
- If you do not see an increase in noise when the receiver is connected to an antenna, a bit of extra gain is recommended. Given an ideal isotropic antenna at a very quiet receive site, it will probably take about 12 dB of gain to comfortably "see" the antenna's noise floor - assuming no other losses (coax, splitter, etc.)
- The preferred location of an amplifier is after the filter described above as it, too, will be protected against the very strong lower-frequency HF signals - even though a device like the above will increase the loss (and noise figure) by about 1.4dB.
- In cases where there are splitting losses (e.g. feeding multiple receivers) it may be beneficial to split the gain. A modest-gain amplifier (10-14dB) might precede the splitters - the modest gain being enough to overcome splitting losses and to maintain system noise figure.
- In the case of a low noise level receive site, the splitting losses may put the 10 meter noise floor below the detection threshold of the receiver and, if necessary, another amplifier may be placed just after the filter described above to make up for it.
- It's worth noting that if you can detect a 3-5dB increase in noise floor with the antenna connected (versus disconnected) than even more gain will NOT further-improve system performance: On the contrary, more gain than necessary will increase the probability of receiver overload - particularly on a direct-sampled SDR that has no AGC in its signal path like the KiwiSDR. If one has more than 3-5dB of noise floor increase with the antenna connected on 10 meters when it is quiet, it's suggested that several dB of attenuation be added. The preferred place to add this attenuation is in front of the amplifier to maximize its strong-signal handling - but only if one can still detect the noise floor on the antenna after doing so. If one has a very high gain amplifier (say 20-25dB) and the gain is excessive, judicious addition of attenuation on both the input and output of the amplifier may be required.
- When an amplifier is to be considered for HF use, it should have clearly-defined ratings - one of the most important of these is the output power capability (often "P1dB" which is the output power at 1dB compression) which, for a modestly good amplifier capable of handling strong, off-air signals, should be in excess of +20dBm. Second to this would be the 3rd order intercept point, which should be stated as being in excess of +30dBm - and the higher the better. Both of these parameters are indicative of how well an amplifier might deal with multiple, strong signals that may be present at the antenna without adding significant distortion of its own.
- If you wish to pick your own frequency and impedance, the following will get you "close enough". At the point where a single section of this circuit (as depicted in Figure 1) has an attenuation of about 4.1dB, the reactance of the "L1" and "C1" components will be equal to the desired characteristic impedance of the circuit - which will also be the same as "R2" and "R3". Unfortunately, other parameters (e.g. the amount of attenuation at a specific frequency) are not predicted by this formula, although it's worth noting that both C1 and L1 will disappear at extremes in frequency and the circuit effectively turns into a resistive attenuator. For example, C1 disappears (goes to infinity ohms) and L1 goes to zero ohms at DC while L1 disappears and and C1 goes to zero at infinity and one can pretend that these particular components no longer exist (e.g. either a short or open as appropriate).