A "sharp" 160 meter receive-only band-pass filter

Why the filter?

Several years ago I needed a "sharp" 160 Meter band-pass filter - one that would pass 160 Meters with little attenuation, but effectively block top-end signals on the AM broadcast band.

Nowadays, the AM (mediumwave) broadcast band goes up to 1700 kHz - but there are relatively few signals in the "new" portion (1610-1700) - yet Murphy's law would dictate that one of those stations would be located near your QTH - but that wasn't the case here.  The actual application was that this filter was to be used at the Northern Utah WebSDR where we have very strong (-20dBm, or about "50 over S9") signals coming across salt water (the Great Salt Lake) - one of which was from a transmitter on 1600 kHz.

If you happen to live near an AM broadcast transmitter and operate on 160 meters, you may well have faced challenges, yourself:

• A full-sized 160 meter antenna (dipole, vertical) will likely do a decent job of intercepting RF from any AM broadcast transmitter - particularly one near-ish the top end of the band.  This might mean that your receiver is subject to very strong signals from that transmitter even if the antenna wasn't designed specifically for that frequency.
  

• The filtering in most HF transceivers and receivers isn't particularly "sharp" and will likely do little to prevent a significant amount of RF energy from getting into the sensitive circuits from AM broadcast signals that are even hundreds of kHz away.
  

• Both of these can cause overload of the RF amplifiers, mixers and/or non-linear responses in the PIN diodes typically used to select filtering.
  

• In a software-defined radio, a strong signal can also overload the same sorts of stages, but there's the additional problem of possibly overloading the A/D (analog-to-digital) converter - or at least, causing the receiver's gain to be reduced (to prevent A/D converter overload) to a point where it starts to become "deaf" to other, weaker signals.  This "gain reduction" also has the effect of reducing the number of A/D converter bits that are used to represent weaker signals - something that can also increase distortion products.

The effects of all of these can a general degradation of the receiver, the most obvious being the generation of IMD (InterMoDulation) products, often manifesting themselves as spurious signal being produced in the receiver that may be harmonics of an AM transmitter and/or "mixes" of two or more AM broadcast signals. ¹  To a degree, these may be reduced by attenuating the input signal (e.g. the attenuator on the receiver) - and if these spurious signals' levels reduce in level more than the "real" signals on the band, that is a sure sign of overload within the receiver itself.

If the above are happening to you - and you care about 160 meters - a filter might help. ²

The challenge of such a filter

Achieving the combination of low loss at 1800-2000 kHz while providing a reasonable amount of attenuation at 1700 kHz - or even 1600 kHz - is a bit of a challenge as the percentage difference in frequency where you want signals to pass and and the frequency that you want to block is small,  but this made the project even more attractive.

As I wasn't going to be transmitting through this filter, a bit of loss was acceptable and this filter ended up losing about 3dB (half of the power - or about 1/2 of an S-unit) through it.  This may sound terrible, but if you have even a modest antenna for 160 meters, you will have way more signal+noise than is needed to deflect your S-meter significantly - even if you are lucky enough to be in an area with no man-made noise as the chart below indicates:

Figure 1:
"Typical" noise floor for various radio environments.  Because the above chart is based on a 500 Hz bandwidth, one would subtract 27dB from the power level to scale to a 1 Hz bandwidth.

As you can see, compared to 40 meters, the noise floor - due to natural sources (the "quiet rural" graph) - is 5-10dB higher on 160 meters than on 40 meters, so it's likely - in most cases - that losing 3dB through a filter will go unnoticed:  The ultimate test would be that of noting whether or not the S-meter read higher with the antenna connected (despite the loss) than without:  If the former, you are probably hearing everything that there is to hear! ³

A practical receive-only filter

The schematic of the filter is here:

Figure 2:
Schematic of the receive-only band-pass filter for 160 meters.
Both "exact" and "standard" values for the capacitors and inductors are shown.
Click on the image for a larger version.

The filter isn't very complicated - using only ten components.  As can be seen from the above diagram the "exact" component values were computed - but I also recomputed using "standard" values of capacitors and inductors and simulated both - first, the "ideal" values using the "Elsie" program configured to take into account the limited "Q" of real-world inductors and capacitors:

Figure 3:
Plot using ideal values, from 1.5-2.5 MHz.
Click on the image for a larger version.

Now, using "standard" values for the capacitors - and some of the inductors:

Figure 4:
Plot using "standard" values over the 1.5-2.5 MHz range.
Click on the image for a larger version.

The result is that the response is somewhat less flat - but only by about dB or so, most notably at the upper end.  In both cases, the filter is attenuating signals at 1700 kHz by 15dB and at 1600 kHz by about 35dB.  15dB might not seem like much, but it represents a 32-fold decrease in signal level and this may well reduce IMD products to inaudibility. ⁴

Building the filter

In analyzing the schematic, you may note L1, L3 and L5 are paired with rather low-value capacitors (220, 150 and 220pF, for C1, C3 and C5, respectively for the "standard" value version) implying higher impedance.  Conversely, inductors L2 and L4 are paired with high-value capacitors (14700pF for C2 and C4) implying low impedance and higher current.

Figure 5:
As-built 160 meter band-pass filter on a piece of glass-epoxy
board using "ME" squares.
Click on the image for a larger version.

On the workbench, I first built the filter using toroid-wound inductors throughout and measured the loss - but I then replaced L1, L3 and L5 with small, molded inductors (which are lower "Q" and higher loss) and noticed only a fraction of a dB difference and no obvious difference in the filter response.  For L2 and L4, I stayed with wound toroids using as large a wire as I could fit on them to minimize the loss:  I briefly tried some 0.47uH molded inductors, but the filter's response was poorer and the loss was several dB higher, so do not be tempted to use molded chokes for L2 and L4.

The as-built filter is shown below:

The filter was built onto a piece of glass-epoxy copper-clad circuit board, the landings using "Me-Squares" from the QRP.me web store (link) glued to the substrate using cyanoacrylate ("super") glue.  The in/out connections are in the lower left/lower right corners and the wound toroids are visible, glued to the board (to keep them from moving around - and to immobilize the windings to prevent mechanical de-tuning) using RTV ("silicone") adhesive.  The three molded inductors are clearly visible as well.  Precise alignment of this filter - mostly squeezing/spreading the turns on the toroids - is easily done with even the most inexpensive NanoVNA.

Figure 6:
The filter installed in the chassis of one of the
filter modules at the Northern Utah WebSDR.
This filter was added on the port feeding the
160 meter receiver.
Click on the image for a larger version.

It's worth noting that the "small" value capacitors (C1, C3 and C5) are NP0 (a.k.a. "C0G") temperature-stable ceramic - but silver-mica capacitors would be an excellent choice as well.  For the larger-value capacitors (C2, C4) which are 14700pF total each consist of a parallel 0.01uF (e.g. 10000pF) and 0.0047 (4700pF) plastic dielectric capacitors:  Silver Mica capacitors would be "better" - but very expensive, but decent-quality plastic capacitors work fine at 160 meters, their internal inductance having minimal effect.

If you build this, DO NOT use disk ceramic capacitors for C2/C4 as 0.01uF and 0.0047uF are not likely to be temperature stable, low-loss NP0/C0G types - but rather they will probably be very temperature-unstable and lossy "Z5U", "Y5P", "X7R" or similar - and these types are completely unsuitable in this application.

As can be seen in Figure 6, the completed filter was installed in the filter module that feeds the 160 meter receiver at the Northern Utah WebSDR, but I could have easily put it in its own, shielded box.

Can a transmit-capable filter be constructed?

In theory, it should be possible to build a filter like this that would allow (survive!) being transmitted-through - but several things would need to be taken into account:

• Low-loss inductors.  The loss through the inductors account for the majority of the losses here, mainly because - compared to capacitors - real-world inductors are terrible.  For L1, L3 and L5, it may be that larger-gauge wire (16 AWG or so) on fairly large toroidal cores would suffice, but for L2 and L4 - which will be carrying quite a bit of current, - very heavy wire (perhaps 10 AWG) would be needed.
• Low-loss capacitors.  Silver-mica capacitors would be used throughout.  For C1, C3 and C5 some high-voltage units (1kV) should suffice, but for C2 and C4, paralleling a half-dozen or so 1kV silver-mica units to attain the desired capacitance - and to divide the current and losses - would be recommended.

I've not been motivated to build a "transmit-capable" filter to test it out, but I suspect that the losses of a filter built as noted above could likely be kept down to about 1dB or so using practical, "real-world" components.

Using the filter

As this filter has a rather high loss - which isn't really important for reception - that is not the case for transmitting:  Running more than a watt or so through it would likely cause heating of the molded chokes - and even if lower-loss inductors were used, it would still have several dB loss and cause heating.  If the molded chokes were replaced with toroidal inductors, it would have slightly lower loss, but likely be capable of handling QRP power levels (5 watts maximum).

What this means is that some means would be required to assure that this filter was used only used in receive:  Some transceivers have connectors to allow the insertion of a filter in the receive signal path, but it may well be that RF relays will be required to switch the filter in/out.

Footnotes

1. IMD products resulting from more than signal source are common.  For example, if there are two local radios stations - one on 850 kHz and another on 1010 kHz they can mix together at the sum of the two frequencies - namely (850 + 1010 = ) 1860 kHz.  This "mixing" is very apparent in the resulting frequency because the signal at 1860 kHz would consist of the audio of both the 850 and 1010 kHz signals.  A harmonic of a transmitter - say one operating on 960 kHz - which might appear at twice the frequency (1920 kHz) would have only the audio of that transmitter - and it would probably be somewhat distorted.  Most of the time, these mixing products are created within the receiver itself.
   
2. A filter will help ONLY if the spurious signals are being generated within the receiver itself.  Low-level IMD (intermodulation) products can also be generated in strong RF fields by non-linear junctions in the vicinity - which can include metal fencing, rain gutters, rusty wires, corrosion, or even the "re-radiation" of IMD from another device that may have high-level RF energy on it.
3. It is usually the case that modern receivers have way more gain than is necessary to receive signals at the noise floor - particularly at lower frequency where the natural noise floor is usually quite high.  What this means is that you can "throw away" signal on the input of your receiver and not actually "miss" any signals at all.  For example, if you have an attenuator on your receiver and without the attenuator the noise floor is S-6, but with the attenuator is still an S-3 - and the noise floor of your receiver without an antenna connected is S-0 - signals in the noise at S-3 with the attenuator switched in are not being lost:  It is only how far above the signals are above the noise that affects their readability.  As noted above, in the presence of very strong signals (as in the case of a very nearby AM broadcast transmitter) or even in the case of an event like Field Day where other transmitters are nearby - if you can still hear the "band noise" with the attenuator switched in, you may well be better off overall by using the attenuator - which can minimize the generation of IMD products within the receiver.  For more information about this, see the earlier blog entry "Revisiting the 'Limited Attenuation High Pass' filter - again" (link).
   
4. As a general rule, IMD products resulting from nonlinearities within an amplifier (or receiver) will diminish by about 3dB for every 1dB of total signal reduction.  In our example where, at 1700 kHz, the band-pass filter reduced the undesired signal by 15dB, this could, in theory, reduce the IMD by 45dB or so - about 5-8 S-units, depending on the radio.  If this IMD was "only" about S-9 to begin with, reducing it by 40+dB may make it inaudible - even though the filter knocked it down by only 15dB in the first place.
   

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This page stolen from ka7oei.blogspot.com

[END]

DXing distant SolarEdge PV optimizer modules (or long-distance propagation of PV system QRM)

From how far away can you hear the spurious emissions from a known-noisy PV system?

Quite a racket!

Figure 1:
The spectrum of a SolarEdge PV system from several meters
away across the 6-8 MHz range showing "spurs" (clumps of
low-level carriers) at 200 kHz intervals and other places.
In the above plot the true nature of the individual peak -
the fact that each contain many carriers - is not apparent.
Click on the image for a larger version.

In a previous post (linked HERE) I described the interference produced by a SolarEdge PV (photovoltaic) system to an amateur from installations on neighboring houses.

The "take-away" from this analysis is that the current version of SolarEdge systems produce rather strong signals at 200 kHz intervals - each module on the back side of a solar panel producing its own carrier at its own frequency as depicted.  The peaks in Figure 1 show these groupings of carriers every 200 kHz (plus some additional frequencies) while the image in Figure 2 shows, in extremely high spectral resolution, many individual, narrow carriers that comprise each of these peaks.

In driving around with an HF mobile station in my vehicle I can hear these 200 kHz-spaced carrier groups almost everywhere around town during daylight hours - the roar getting much stronger in/near residential areas as you would expect.  If driving through a residential neighborhood, it is very easy to tell when you drive past a house equipped with a SolarEdge PV system - and it is easily audible from a block or two away.  Knowing the "fingerprint" of this PV system allows it to be identified uniquely - even at some distance.

Figure 2:
A "zoomed in" view of the spectrum of local SolarEdge
carriers recorded just below 7.4 MHz from my home.
See Footnote #3, below for detailed information.
Click on the image for a larger version.

Are they DX? ¹

A question arose in my mind:  Does this "grunge" produced by the SolarEdge PV systems propagate long distances?

To answer this question I checked a KiwiSDR at the Northern Utah WebSDR (link) - a site with which I am very familiar ².  This receive system is located about 3 miles (5km) from any residential area, bounded on three sides with mosquito-laden bird refuges (wetlands) and on the fourth side - the same as the closest houses - by a mountain.  Additionally, the antenna used for the reception in Figures 3 and 4 below was the TCI-530 omnidirectional log-periodic (with circular polarization) - which does not have good gain at very low radiation angles, further precluding the reception of "nearby" PV systems via "ground wave".

The quick answer to the above question is YES - the roar of SolarEdge systems is propagated when conditions are "reasonable" ⁴ as shown in the screen capture below:

Figure 3:
Propagated noise from myriad SolarEdge PV systems from the remote Northern Utah WebSDR's
remote HF receive site.  The "hump" in the middle is the combined energy of likely thousands of
SolarEdge PV systems that are being ionospherically propagated.  Amateur signals are
visible at 14.200 MHz and above.
Click on the image for a larger version.

The signals represented by the "hump" in the highlighted portion in the center of the analyzer plot in the top part of the image - and the "band" of noise on the waterfall display - between 14.199 and 14.200 MHz are the sum of the propagated low-level PV system carriers from... who knows where?  To be clear, this energy is not likely to be from just one SolarEdge PV system and its individual optimizers (one for each panel) but more likely from the many thousands of such devices that are each, individually, radiating energy.  What we are seeing is the total energy of the propagated systems, the frequency spread being centered around 14.1993 MHz.

It's worth noting that the fact that these signals do not land on exactly the same frequency ⁵ - hence the Gaussian-like distribution of energy - and this has interesting implications.  Even though the signal from each, individual optimizer is (more or less) a CW (unmodulated) carrier, the fact that there are so many of them clustered together means that, for statistical purposes, they might as well be a distribution of noise energy:  Unlike with a single coherent CW signal, the DSP filtering on modern radios will be able to do little/nothing to reduce their effects if they were to cause interference due to its similarity to white noise.

A quick power and spectral analysis of the signal above showed that if the signals above were a single, coherent CW signal, the total amount of energy contained in the "hump" in Figure 3 would have easily been at least 15-20dB above the noise in a 50 Hz detection bandwidth:  A CW signal of this strength would certainly be cause for complaints!

I also looked at other 200 kHz multiples around 14.000 and 14.400 and the same, exact types of signals were present on those frequencies - and similar bunches of energy fitting this profile were noted at least as low as 10.200 and as high as around 18.200 MHz as well (probably higher) and every (otherwise) clear frequency in between - this range being related to current ionospheric propagation at the moment that I checked (e.g. around 1845 on September 10 UTC, 2025).  ⁶

To verify that these signals were propagated and were likely from SolarEdge systems, several things were done:

• The presence at many 200 kHz multiples/intervals across the HF spectrum is telling!  Their being slightly below exact 200 kHz multiples as noted in Footnote 5 adds to their "uniqueness".
  
• On days with poor propagation overall, these signals were absent - or limited to frequencies commensurate with the MUF (Maximum Useable Frequency).

• These signals disappear at night.  (This test is somewhat complicated by the fact that propagation on these bands also changes at night - but sunlight is still illuminating the ionosphere well after sunset on the ground.) 
  

• An "S-meter" plot was run over the period of several minutes:  A propagated signal(s) would show variations in signal strength - but this can be foiled to a degree by the fact that many, many individual point sources would each be propagated differently and unlike a single source, would not experience as deep a fading as the plot below shows:

Figure 4:
Propagated signal strength variations caused by ionospheric variations.  This would seem to indicate
that the signals are propagated - but the magnitude of the fading would be mitigated by the large
number of point sources, each being affected individually along the signal path.
The top/bottom of this chart represents 10dB.
Click on the image for a larger verion.

As noted in the original article analyzing a system close-up (linked above) the SolarEdge optimizers produce other signals 6-10 dB weaker at various points above each 200 kHz interval - these are visible in Figure 1.  When the above plots were made these signals weren't readily apparent - but I suspect that they will be visible during "excellent" propagation conditions rather than the "mediocre-to-average" conditions that were present when Figures 3 and 4 were produced.

Conclusion:  They do get propagated!

So yes, you can DX SolarEdge PV systems - it's just that there are so many of them each doing their own radiating that you probably won't know from where those signals originate, so it's hard to know from how far away you might actually be hearing them!  To be clear, it's difficult to determine if a the radiated RF from a single optimizer would be audible via ionospheric propagation, and with many thousands of them out there this may be impossible to determine - but it is clear that the summation of many thousands of them does produce an audible signal.

Do these signals actually cause QRM ⁷ ?  As noted in the earlier post (liked above) they most certainly do if you live within a city block or two of one of the SolarEdge PV systems and operate on or near any of the frequencies occupied by the spurious radiation represented in Figure 1.  If your receive system is located well away from a SolarEdge installation, the above shows that you may still experience interference from these systems - even from a significant distance.

Figure 3 also shows that the emissions do propagate over long distances:  The 20 meter band's optimal "single skip" distance would likely place the majority of these signals in a 700-1500 mile (1100-2400 km) radius of Northern Utah - and this includes quite a few populated areas in parts of the U.S. where the number of solar installations is quite high. 

You, too, can check for QRM at your station

If you have an HF station with a receiver with a waterfall display you might want to check the various amateur bands just below the 200 kHz multiples ⁸ during daylight hours:  If there is a SolarEdge PV system within a couple city blocks of you ⁹ you will most likely see and hear it - but don't blame me if, after finding that you can see those signals, you can't "un-see" them!

* * *

Links to related pages (about solar power) on this blog:

• Analysis of a SolarEdge system (link) - This is the article linked at the top of the page where careful measurement was done to characterize the interference created by a SolarEdge system neighboring a local amateur.
  

• It *is* possible to have an RF quiet home PV system (link) -  On this page I describe the SunnyBoy based system at my home QTH - and the fact that it does NOT produce QRM.
  

• Reducing QRM from a Renogy 200 watt (or any other) portable solar panel system (link) - Here I describe a portable solar system that I take "car camping" and how I reduce its interference to the point of inaudibility.

• The Solar Saga Part 1: Avoiding Interference - (Why I did not choose Microinverters) - link

• The Solar Saga Part 2:  Getting the system online - link
  

• Does the Tesla Powerwall 2 produce Radio Frequency Interference? - link
  

• Reducing RFI from the Tesla Powerwall 2 - link 
  

• Tesla Powerwall susceptibility to RF Transmissions - and how to deal with it - link 

Footnotes:

1. The term "DX" means distance.  Generally speaking, if a signal is "DX" it is understood that it must be being propagated over much more than a line-of-sight distance - in this case, via ionospheric propagation at distances of hundreds or thousands of miles/km.
2. The author of this post is one of the original founders and current maintainers of the Northern Utah WebSDR which has a remote HF receive site about 80 miles (94km) north of Salt Lake City.
3. Figure 2 shows a "close-up" spectral view of the signals emitted by several SolarEdge PV systems within a mile/kilometer or two of my house - the closest system being about a block away.  The center frequency of this cluster of signals was approximately 7.39965 MHz and a 256k-point FFT with a bin width of 183 mHz (milliHertz) - along with some averaging - was used to create this plot.  Clearly visible are a large number of individual carriers along with a background "roar" of many more weaker carriers that are not individually distinguishable in this plot.  This plot was purposely done on a frequency above the 40 meter amateur band during daylight hours (the local time is visible in the image) and during this time there is no strong, long distance propagation (a fact verified by the absence of a similar set of signals on the remote Northern Utah WebSDR site) indicating that this energy is, in fact, originating from systems proximate to my own receive site.  At sunset, these carriers will gradually disappear - often "blinking" out - as the solar panels lose their light and will reappear the next morning:  This "blinking" can be heard as individual tones flicker on/off during the day<>night transition by listening on an ordinary SSB-capable receiver at one of the frequencies noted above.
4. The frequencies mentioned have also been checked when ionospheric propagation is poor (comparatively few strong signals) and the characteristic SolarEdge carriers were absent at the remote receive site.  This further illustrates the fact that the signals described above are not local to the remote receive site and reinforces the likelihood that they are, in fact, being propagated. 
5. Observation of a SolarEdge PV system at very close distance (less than 50 feet/15 meters) indicates that each, individual optimizer - a device attached to the back of every individual solar panel - will radiate the signals at 200 kHz intervals.  Due to the slight variations in oscillator frequencies (e.g. quartz crystals or MEMs devices) the precise frequencies of these signals - and their harmonics - will vary, but the mean frequency separation appears to be around 199.9901 kHz which puts them slightly below a precise 200 kHz multiple which is why the peak of the distribution shows up around 14.1993 MHz on 20 meters, 7.19965 MHz on 40 meters and so on.  As noted in the text, the actual frequency spread of the individual modules is such that it has a Gaussian-like distribution above and below the mean frequency.
6. I also checked several remote receive systems around the world during their local daylight hours and could see the same "humps" of energy at frequencies just below the aforementioned 200 kHz multiples on some of them.  One such system was that located at the University of Twente in the Netherlands:  It is not known to what degree the signals that were radiated (likely) from PV systems were propagated and which might be within a few kilometers of this receive site, but they are certainly "there".
   
7. "QRM" is a "Q" signal referring to "Man Made Interference" and the magnitude of this interference in comparison to the desired signals determines if this is harmful interference.  If QRM makes it difficult/impossible to receive a signal on frequency, that would fit the definition of harmful interference.
8. The frequencies on which the radiated signals from a SolarEdge PV system (every 199.9901 kHz) will likely land within an HF amateur band are clustered around the following:  3.5998, 3.7998, 3.9998,  7.1996, 14.1993, 21.1990, 21.3990, 28.1986, 28.3986, 28.5986, 28.7986, 28.9986, 29.1986, 29.3986 and 29.5986 MHz plus similar frequencies in the 6 meter band:  They can also be heard on non-amateur frequencies at the same 199.9901 kHz intervals as well.  As the above frequencies are the actual frequencies, you will need to tune above or below the frequencies (using LSB or USB, respectively) by 1.5 kHz or so to hear the "roar".  Of course, you will only hear these signals during daylight hours when the PV systems are active.  Note that the combination of naturally-higher noise levels on the lower bands (80, 40 meters) and the likely lower efficiency of the PV system's component ability to radiate RF there - plus the tendency for nighttime propagation on those bands (when the PV systems are inactive) - means that observing this phenomenon on those frequencies via the ionosphere is much less likely.
9. If you do some remote operation like POTA or SOTA at a significant distance from any likely PV system, you might want to take a look at one of the 200kHz-interval frequencies mentioned above during daylight hours and good propagation:  You'll probably see the propagated PV signals there, too.
   

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This page stolen from ka7oei.blogspot.com

[END]

Exploring the Ameco PCL-P Nuvistor cascode preamp/preselector

"And now, for something completely different!"

This past January - at Quartzfest - there was a table covered with "junque" and taped to it was sign with the word "FREE" on it.  That's how I ended up with this box.

Figure 1:
The front panel of the Ameco PCL-P preamp.  The left-hand
control tunes the front end of the preamp while the right-hand
control selects the "band".  The in/out switch is on the right.
Click on the image for a larger version.

The Ameco PCL-P

The PCL-P - which went on sale around 1965 - seems to have originally cost around $32.95 according to the RadioMuseum web page (link) - equivalent to around $300 today!  ^{Footnote 1}. The specifications say that it has about 20dB of gain and can be tuned for any frequency from 160 through 6 meters.

But what's it for?

Back in 1965 many amateurs still used separate receivers and transmitters - and it was often the case that this gear would, itself, be at least a few years old - likely WW2 surplus and/or gear from the 1950s.  Similarly, shortwave listening was still in its heyday and it's likely that many of the receivers used by SWLs (ShortWave Listeners) were also likely to be "vintage".

In those days, tube (e.g. "valve") based gear was still the rule and this - particularly for older gear (from the mid-late 1950s and earlier) - often meant several things were likely true about the receivers:

• Insensitivity on higher bands.  On the higher bands - namely 15-6 meters - it was often a struggle to attain good sensitivity at these higher frequencies.  This is particularly true on "simple" (e.g. inexpensive) where sensitivity would be fine on lower bands, but drop off precipitously with increasing frequency where signals were generally lower, anyway  Remedying this is surely the main purpose of this device.
  

• Image rejection may be marginal.  Most receivers of this vintage were single conversion - that is, they converted from the receive frequency to a lower-frequency IF (Intermediate Frequency) - typically around 455 kHz.  Some "fancier" receivers converted to something in the lower MHz range (often between 1.6 and 2 MHz) and then down-converted to something even lower - often in the 40-100 kHz range - where the final band-pass filtering was done. 
  

A device like the PCL-P might be touted as an aid to mitigate both of the above:  Its gain and low-noise amplification should help a "deaf" receiver and the fact that this device is somewhat selective may help the image problem as well - although that last point is debatable.

Whether or not a device like this was really helpful or not isn't strictly relevant to our discussion - rather, this article mostly is about the device itself.

Inside the PCL-P

Let's first take a look at the schematic diagram of PCL-P:

Figure 2:
Schematic of the Ameco PCL-P preamplifier.
Additional component annotations added to aid clarity of the description below.
Click on the image for a larger version.

First, notice S3a and S3b on the input/output terminals:  This allows the user to bypass the amplifier entirely - most useful when the unit is turned off - but note that this switch does not power down the unit when set to "out" (bypass) mode.  Immediately following S3a is S2a which is a rotary switch used to select the frequency range:  As can be seen from Figure 1, above, this switch has four overlapping frequency ranges:  1.8-4, 4-10, 10-23 and 23-54 Megacycles ^{Footnote 2}.

L1 is a coil (actually an autotransformer)  - tapped at 50 ohms - that covers the lowest frequency range (1.8-4 Mc) and is the large coil visible in Figure 3, below, but the higher bands' couplers - in the form of T1-T3 - are transformers (actually axially-wound coils with another winding over the top) clinging to the rotary switch, the turns ratios of the primary to secondary providing the impedance transformation from the 50 ohm input to the tuned grid circuit:  All of these, switched by S2b, connect to C1, an air variable tuning capacitor across the grid of the first of two vacuum tubes (valves), V1.

It's worth noting that the fact that this preamplifier is tunable is more of an artifact of the necessity of the technology used:  While it would, in theory, be possible to construct a "no tune" broadband amplifier to make its use slightly more convenient, but doing so - and maintaining equivalent performance over this wide frequency range - would have been a challenge.  The obvious advantage of making it tunable is that rather than amplifying the entire HF spectrum at once, its amplifying is limited to the vicinity of the frequency at which the input network is resonant meaning that by rejecting frequencies elsewhere, it's less likely to be overloaded by RF energy that is well away from the frequency of interest (e.g. strong shortwave broadcast stations on other bands).

There are two identical tubes here - 6DS4s in the case of my preamp  (other units may have been equipped with the similar 6CW4) and these are Nuvistor tubes:  About the size of a very large pencil eraser, these were some of the smallest vacuum tubes that were mass-produced - most Nuvistors being triodes like V1 and V2, above.  Being very small, they were well-suited for high frequency operation, finding their way into the UHF tuners of many contemporary televisions:  It was at about the same time as this unit was made that U.S. Federal law mandated the inclusion of UHF tuners on all new TVs so Nuvistors were widely available and comparatively inexpensive owing to the economy of mass production.  (Wikipedia article about Nuvistors - link).

Figure 3:
Top view of the Ameco PCL-P chassis, the variable capacitor
visible near the upper-right, L1 the big coil in the center and
the two Nuvistors visible just below/left of center.
While this was originally equipped with RCA (phono)
 "Motorola" type connectors,
it has since been retrofitted with BNCs.
adsfadsf
Click on the image for a larger version.

To some, the connection between V1 and V2 may look a bit odd, but the description on the front panel (seen in Figure 1) gives a clue:  They are connected in cascode configuration - possibly a portmanteu for "cascaded triode/pentode" or similar.  In this configuration the "bottom" tube (V1 in this case) gets its plate voltage via the cathode of the "upper" tube (V2) - but you might notice something else:  The grid of V2 is at RF ground via C3 - being somewhat neutrally biased at DC by R2 which allowed current from V2's plate to get to V1's plate via V2's cathode.

This cascode circuit has a distinct advantage for higher frequencies:  As the current through V2 (effectively running in "grounded grid" configuration) is somewhat proportional to its grid-cathode voltage, when V1 conducts more - trying to pull the cathode of V2 lower - V2 conducts harder in response.  As V2's grid is "grounded" at RF via C3, pulling its cathode lower effectively increases the grid-to-cathode voltage:  V2 also tries to counter this by conducting more, trying to pull the cathode back up.  Because of this arrangement, the voltage on V2's cathode (and, of course, V1's plate) changes relatively little compared to the change in current through it.

What this means it that the effect of Miller capacitance is minimized ^{Footnote 3}.  Here we are concerned with the capacitance between the grid and plate of the tube - V1 in this case - and this capacitance couples the two together lightly, but this has the bad side effect of somewhat cancelling out the tube's amplification action:  As the grid voltage tries to go up with the input signal, the plate voltage would - in a typical single-tube circuit - go down by a comparatively large amount as the tube conducts more in response - and the capacitance between the two will cancel out the signal on the grid to a degree:   This is one of the reasons why it can be difficult to get a single-tube RF amplifier to work well at high frequencies.  If we prevent the plate voltage from changing as much and convey the signal more as current instead - as we are doing with the action of V2 in this cascode circuit- we can significantly reduce the Miller effect. 

Figure 4:
The underside of the PCL-P chassis prior to repair - the 2-
section yellow capacitor and the diode on the left.
Click on the image for a larger version.

With the cascode configuration, the swing of the plate voltage of V1 is minimized - and so is the Miller effect, resulting in better gain, flatter frequency response and potentially, lower amplifier noise overall.  As such, we get varying current on the plate of V2 which, via transformer T4 (visible on the far right in Figure 4 as several turns of enameled wire on what appears to be a threaded, ferrite transformer core) is coupled to the output.  Resistor R3 was likely added to help ensure stability of the amplifier both when it is being bypassed (the input and output having nothing connected to either) and also in the event that the input impedance of the receiver connected to the (un-tuned) amplifier output is a poor match at some frequencies.

The rest of the circuit is a pretty straightforward power supply:  The PCL-P used a silicon diode (D1) to half-wave rectify the plate supply, filtered first by C8 - the neon power-on indicator (V3) is connected to this point via R5 - and then decoupled by 1k resistor R4 and filtered again by C9:  The ultimate result is a nice, clean source of about 145-155 volts for tubes when this is operated from a modern 123 volt U.S. mains source ^{Footnote 4}.

Construction quality

I'd say that the Ameco PCL-P is constructed "well enough":  It looks as though a bit of thought and refinement occurred to assure stable operation at 6 meters - a frequency range that was above what the average amateur of the mid 1960's had for equipment - while maintaining low cost and simplicity.  A nice touch is the use of a feedthrough capacitor (C4) as a component mounting point/stand-off (not actually "feeding through" the chassis, though) and bypass for the plate supply feeding the bottom of the output transformer, T4:  This is surely the one place where the use of a somewhat expensive component was absolutely necessary as a lowly disc ceramic would probably not have sufficed owing to the comparatively high ESR and self-resonant properties that type of capacitor.

From what I can tell, the PCL-P was originally fitted with "RCA" (phono) "Motorola" type connectors (like those found on car radios) on the input/output - a somewhat common practice on HF, VHF and even UHF amateur and commercial radios - but they have clearly been replaced with the more-common BNC types by a previous owner.

Refurbishing

Figure 5:
This time, with a new diode and capacitors on the left.
Output transformer T4 is visible near the right edge,
supported by feedthrough capacitor C4.
Click on the image for a larger version.

Although I don't really have any intention to put this device into regular service, I did want to get it into operational condition.

Carefully powering it up on a current-limited mains supply, I noted that the dual-section power supply capacitor (C8/C9 - in the same yellow tube visible in Figure 4) was bad with about 10 volts ripple on the plate supply - but I was able to verify that the unit had good gain, indicating that both of the Nuvistor tubes were working properly despite receive signals being overlaid with 60 Hz "hum".

As the line cord was in very good shape the only thing I had to replace was the yellow dual-section capacitor (C8/C9) with individual 22uF, 200 volt units (partly to accommodate the somewhat higher mains voltage these days) - but I also replaced the diode (D1) with a more modern 1N4007 with a 1kV rating.  Ultimately, the ripple on the plate supply was well under a volt - as it should be!  (Sharp-eyed readers may have noticed that the PCL-P is sitting atop the defunct filter capacitor in Figure 1.)

Not surprisingly, I noted that the transformer in this amplifier "buzzed" quite a bit - but with a half-wave, capacitor-input rectifier conducting on the peak of every half-cycle, this isn't unexpected:  The addition of a resistor (say, 100-470 ohms) in series with the diode (D1) would probably reduce this by limiting the peak current on the top of the AC waveform.

Performance

It's worth noting that any amateur receiver made by a major manufacturer since the 1980s - when it is working correctly - will very likely have more than adequate sensitivity on all bands to hear the local receive noise floor, so the PCL-P amplifier probably has little place in the modern ham shack - but for a "deaf" radio from the 1950s and 1960s, of which there were many - particularly if they were in need of alignment - it would have likely been useful.

The one place where this unit might be useful in the modern ham station - if only for nostalgic purposes - might be for a low-gain wire antenna (e.g. Beverage-On-Ground, Loop-On-Ground or Loop-Under-Ground) for the 160 and 80 meter band.  Nevertheless, I decided to check the gain and selectivity of this device in the (non-WARC) amateur bands 160 through 6 meters:  I have included these plots and comments below the conclusion of this article.

According to the official specifications of this amplifier, its gain is about 20dB - and my measurements - with 50 ohms in/out - corroborate this, more or less:  At 10 and 6 meters it fell slightly short of this figure, but not dramatically so and this variance can be forgiven given the vagaries of manufacturing differences and age.  It's worth noting that the 6DS4 triodes used in this copy have a very slightly lower rated gain than the nearly-identical 6CW4 triodes (an amplification factor 63 versus 65) that the schematic notes as an alternate, but the difference would likely be negligible in the real world as the in-circuit gains would surely be much lower - or in the case of this amplifier, it's around 20dB (e.g. voltage amplification factor of 10 and a power gain of 100).

Unfortunately, I don't have a means of accurately measuring the noise figure, but testing with a "modern" radio (an FT-817) across HF and 6 meters indicates that this amplifier is NOT noisier than the FT-817 implying that its noise figure is at least as good as it needs to be to be able to hear above the atmospheric noise level - even in an RF-quiet environment.  These Nuvistor tubes are capable of a noise figure of as low as 3dB on 6 meters, but mismatch and losses in the input (and, to a lesser extent, the output) networks would surely degrade this - but a noise figure of only about 9 dB  ^{Footnote 5} is likely to be sufficient in 6 meter work for anything other than, perhaps, EME (Earth-Moon-Earth).

Above, I touched briefly on the idea of IF image rejection being slightly improved by a device like this that offers a bit of band-pass filtering:  With a single-stage L/C filter, any improvements afforded by it are likely significant only at the lowest frequencies where the width of the peak is at its narrowest - but negligible on the  higher bands as noted in the comments below the response plots.

Conclusion

As noted earlier, the PCL-P Nuvistor preamplifier is probably not a useful addition to a modern-day ham shack with radios made since at least the 1980s:  The issue that it solves - notably that of addressing the lack of sensitivity of some older radios on the higher bands - is simply a "non problem" these days.  If you have some old "boat anchor" radios - particularly of the less-expensive variety - this sort of device may help pick up weak signals - particularly on a mostly "dead" band.

The noise floor of this preamplifier appears to rival that of a modern radio - but this doesn't mean that it would improve the sensitivity of a such a radio, but only that it would simply make the S-meter read higher without improving the signal-to-noise ratio:  If a radio in question can already hear the noise floor on a given band when connected to your antenna, further amplification will not improve absolute sensitivity and may simply degrade receiver performance by feeding it with too much signal!

As it is, this unit will sit on a shelf with some other "vintage" gear, always ready for some possible future use.

* * *

Footnotes:

1. If you think about this for just a second, you can buy some really nice accessories for $300 these days such as an automatic antenna tuner, a low-end laptop, or even one of several very nice QRP radios - some of which are software-defined radios.  How times have changed!
   
2. Until somewhere around 1970 or so, it was common - at least in the U.S. - to use "cycles" (e.g. Cycles per second) rather than Hz (Hertz) which is why older equipment may show "kc" (kilocycles) and "Mc" (Megacycles) rather than the modern "kHz" (kiloHertz) and "MHz" (MegaHertz), respectively.  And no, you don't need a special "Mc to MHz" converter to use your old receivers!
3. As noted, the Miller capacitance is often a limitation on the performance of high-frequency/high speed electronic components which is why the cascode configuration is used - and a similar reason why transimpedance amplifiers are the norm for interfacing with photodiodes in high-speed optical detectors  The Wikipedia article on the Miller effect is here:  link.   
4. When this unit was made the nominal residential mains voltage in the U.S. was closer to 110-115 volts and now it is more typically in the 120-125 volt range.  It's unclear when this (gradual) change occurred - and it didn't seem to happen everywhere in the U.S. all at once - but the shift from "about 115" to "around 125" likely happened over the period of the mid 1960s into the 1980s.  "Vintage" gear - that being from the 1960s or earlier - likely was designed to operate closer to 110 volts (especially devices from the 1940s and earlier) than 120 volts meaning that the supply voltages (filaments, B+, etc.) are going to be higher as will the magnetization current/losses in the transformers - something to consider if you routinely operate such gear:  The use of a Variac ^TM or a "buck" transformer in series (e.g. an out-of-phase 9-12 volt filament transformer wired to reduce the 120 volt mains) is suggested to prevent overvoltage of filaments, capacitors, transformers, etc. to maximize the lifetime of those components.
   
5. The article "Measurements on a Multiband R2Pro Low-Noise Amplifier System, Part 2" by Gary Johnson, WB9JPS, discusses the effects on noise figure on real-world performance and concludes that a receive system noise figure of 9dB is likely to be adequate for typical 6 meter operation:  Link (from the Web Archive)
   

 * * * * *

Frequency response plots of the Ameco PCL-P preamplifier/preselector

The following plots were taken using a DG8SAQ VNA with 20 dB of attenuation on its "Output" port (connected to the input of the PCL-P) and 6 dB of attenuation on its "input" port (connected to the PCL-P's output) to prevent overload of both the preamplifier and the VNA as well as present a nice, resistive 50 ohm source and load impedance.  (Ignore the S11 and Smith plots as I forgot to turn them off).  These plots cover the range from 1 through 80 MHz, overlapping all of the HF bands (plus 6 meters).  I did note that all of these bands overlap slightly, leaving no "gaps" in coverage and as expected, the gain and the "sharpness" of the filtering in these overlap areas (e.g. top end of the lower band with the tuning capacitor near minimum and the bottom end of the next higher band with the capacitor near maximum) were slightly different:  None of the amateur bands tested below fell  entirely within an "overlap" area.

For the response plots there is a marker (#2) indicating the center (peak) frequency while other markers indicate the -10dB and -20dB responses (relative to the peak) - the numbers in the upper-left corner indicating the forward gains at those frequencies.

The final plot shows the insertion loss of the unit when the "in/out" switch is set to "out" (bypass).

Click on any of the plots below for larger version.

Tuned to 1.9 MHz (160 meters) in the 1.8-4.0 MHz position, the peak gain being about 23dB.  The preselector does a decent job of rejecting a possible IF image (910 kHz above the center frequency for a 455 kHz IF).  Note also that the input preselector does a decent job of attenuating much of the AM broadcast band - although it might still be overloaded by a local transmitter operating near the top end of that band.

Tuned to 3.7 MHz (80 meters) in the 1.8-4.0 MHz position, the peak gain being a bit short of 28dB.  On 80 meters and higher there is only minimal image rejection for 455 kHz IF radios.

Tuned to 7.2 MHz (40 meters) in the 4-10 MHz position, the peak gain being just under 24dB.

Tuned to 14.2 MHz (20 meters) in the 10-23 MHz position, the peak gain being just under 23dB.

Tuned to 21.2 MHz (15 meters) in the 10-23 MHz position, the peak gain being just under 23dB.  At these higher bands the limitation of the simple, single-stage L/C filter starts to show up as an asymmetrical response - the filtering above the center frequency being less effective that below it.  Note also that at the marked 20dB point above the center frequency (marker #5) the gain of the amplifier is still about 2dB!

Tuned to 28.5 MHz (10 meters) in the 23-54 MHz position, the peak gain being just under 19dB.

Tuned to 52 MHz (6 meters) in the 23-54 MHz position, the peak gain being just a bit more than 19dB.  Its worth noting that the input network does appear to attenuate signals in the FM broadcast band by more than 20dB - something that may have been useful for receivers that suffered from ingress from a strong, local transmitter.

The "through" loss when switched to bypass ("out") mode.  Loss is measured at 0.53dB at 53.5 MHz and 0.16dB at 28.1 MHz as indicated by the markers.

This page stolen from ka7oei.blogspot.com

 [END]

Reducing RF susceptibility for the HamGadgets "Ultra Pico Keyer" - and mimizing RF issues on portable HF stations in general

Note:

While this article describes a modification of the Pico Keyer to reduce RF susceptibility, it also talks about methods to minimize/reduce RFI-related issues in general for both portable and "base" stations:  This specific topic is covered near the end of this blog entry.

POTA operation 

Over the past several years I've done a bit of POTA (Parks On The Air) operating, racking up "about" 1000 contacts as an activator in a number of parks - usually as an "activator", and mostly on CW.  Typically, I have operated from a campsite using a portable antenna - usually the JPC-7 loaded dipole (discussed in this blog entry) or the JPC-12 loaded vertical (discussed here) - but I have also used an end-fed half-wave and a simple dipole on occasion - and even the Yaesu ATAS-100 on my vehicle.

Figure 1:
Operating CW POTA from US-0004,
Arches National Park in Utah
Click on the image for a larger version.

In the recent past there has been a revolution in portable power sources in that a LiFePO4 battery - which can supply 20-ish amps - is both light enough to be practical and fairly inexpensive.  For those instances where I may be staying at one location for several days the advent of inexpensive solar to maintain the power budget - and the solar controllers can be made to be RF quiet to make it compatible with HF operation (see this article).  With this in mind it's practical to operate the transmitter at 100 watts much of the time, something that makes it as easy as possible for those who wish to work me.  Despite the ability to run 100 watts, I have occasionally operated QRP (5 watts or less) - again, usually on CW.

A memory keyer

Having used a number of different radios for POTA operation (Yaesu FT-100 and FT-817, Icom IC-706MK2G and even a RockMite) - none of them with a memory keyer - I decided that an "Upgrade" was in order so I got the Ham Gadgets "Ultra Pico Keyer" (Link here).  This device is small, powered by a single CR2032 lithium coin cell and costs about US$40 as a kit (not including shipping) including a (partially) 3-D printed case.  For portable use, I couple it with the "Outdoor Pocket Double Paddle" (with magnets!) from CW Morse (link).

This is a nice, little device in that it provides a consistent interface to the user - no matter which radio you might use - and it has a number of message memories (up to eight), perfect for an activity like POTA where a message (e.g. "CQ POTA") may be repeated many, many times during the course of the operation.

Getting "stuck" 

Figure 2:
The Ham Gadgets "Pico Keyer" (left) along with the
CW Morse Outdoor Paddle.
Click on the image for a larger version.

While the Ultra Pico Keyer works as advertised, I did notice a problem on the first trip out while using a portable antenna:  It would get "stuck".

Clearly, this was an RF susceptibility issue - verified by reducing transmit power and observing that it no longer happened.  In short, at 5 watts there was usually no issue, but at 100 watts  the radio would stay keyed continuously after the first Morse element whether it was sent from a stored message or via the paddle:  While it was "stuck", I could still hear the sidetone - via the keyer's internal speaker - sending the message or what was keyed via the paddle indicating that it was not the microcontroller that had crashed but the circuitry that keyed the radio that was the problem.

Further testing showed that when the unit got "stuck" due to RF and simply unplugging the paddle from the back of the keyer would cause it to release (get "un-stuck").  The fact that this happened using a portable antenna provided further evidence of potential RF sensitivity.

Analyzing the problem

As I'm wont to do, I decided to take a look at the Pico Keyer's schematic to see if there was something about its design and construction that might make it more susceptible to RF interference - and I was surprised at what I found.  Here's the diagram found in the manual that is freely available online on the web site (link):

Figure 3:
Annotated diagram of the Pico Keyer with RF current paths shown.
The components in question are Q1 and Q2, in the upper-right corner.  The lines highlighted in yellow are those through which RF currents will flow (between the radio chassis and the paddle/cable) if no bypass capacitor is installed. 
The added capacitor is shown below the "OUTPUT" jack near the upper-right with the RF current path around Q1/Q2 shown in magenta.
Click to get a larger image.

While there are protection capacitors on the paddle input (C1, C2) my eye was immediately drawn to the output keying (upper-right) where I was, at first, confused as to the arrangement with an N-channel MOSFET in both the keying line and the "common" (ring) of the "OUTPUT" connector (e.g. Q1 and Q2) - but then I remembered that the manual stated that this device would key both positive and negative voltages, explaining the "unusual" arrangement.

While admittedly clever, I could immediately see a susceptibility issue here - the problematic RF current path highlighted in yellow in Figure 3, above:  The "OUTPUT" jack more or less will "float" compared to the "ground" of the keyer itself, which is also connected to the "ground" lead of the cable to the paddle as well as the external paddle itself.  This configuration almost guarantees that there will be at least some RF current flowing from the radio and through the keyer's output circuit for several reasons:

• If you are using this in a portable situation, the radio will surely have some RF on its chassis.  As noted in the final section of this blog entry, it's almost impossible to prevent all RF current from getting onto the feedline - even if you do use a common-mode RF choke and a very nearby antenna is likely to immerse the radio and its interconnecting gear in a rather strong radio-frequency field.

• The paddle and the cable that connects it to the keyer should be considered as part of an antenna - and this situation is made worse if one is sitting at, say, a metal table and also if you, the operator, place your hand at/near the paddle/cable, further encouraging a "through" path for RF.
  

What this means is that there will be at least some RF current flowing from the radio chassis, through the keyer and then, as indicated by the yellow-highlighted lines - via transistor Q2 (and Q1) and then through the cable to the paddle.  I didn't really investigate the exact mechanism by which RF current through this path was causing the keying line to get "stuck" - but here are a couple of possibilities.

• RF may be coupling from the drain of Q2 into its gate - and subsequently into Q1's gate as well, which is tied in parallel with it with the peaks of the RF voltage turning on the FET.  Even if RF through the FET was causing it to conduct only on half of the RF cycle, this would surely be enough to key the radio.  It's also possible that the transistor was turned, on average, only "partially" on by the RF energy - not enough to completely shunt out the RF, but enough to key the radio.

• The RF could also be getting into the output pin of the microcontroller via the FET, causing its totem pole output to get "stuck" on while it was present.

Figure 4:
The added capacitor(s) can be seen soldered between the
"sleeve" pins of the "OUTPUT" and "PADDLE" jacks,
on the bottom of the board.  As you can see, I've made this
modification to both of my Pico Keyers!
Click on the image for a larger version.

Regardless of the cause, the fix was clear:  Add a capacitor to bypass RF current around Q1 and Q2 and the output pin of the microcontroller.  In Figure 3, above, the magenta highlight shows how the added capacitor conducts RF currents around the sensitive components.

When this occurred, I happened to be on a POTA activation, but I had my "electronic toolbox" in the car which included a number of useful items such as a soldering iron and a smattering of useful electronic components (a some common resistors, capacitors, etc.).  Grabbing a 1000pF capacitor, I connected one end to the "sleeve" (ground) pin of the "PADDLE" jack and the other end to the "sleeve" of the "OUTPUT" jack - effectively providing a bypass to RF energy on Q2's drain to the circuit "ground" to eliminate any RF voltage potential between the cable connecting the radio and that going to the paddle.  

This modification completely solved the problem:  It is my opinion that this capacitor should be supplied with the kit.  Additionally, Q2 could be eliminated completely and its source/drain leads jumpered if negative keying is not needed.  ^{See Footnote 1}

Since the topic of "RF on the rig" was already broached, the rest of this article will describe how to reduce it.  It's worth noting that the susceptibility of the memory keyer was such that even with the measures described below, it was affected at 100 watts.

* * * * *  

Suppressing RF on the gear and connecting cables

Some readers of this may immediately say "You are obviously doing something wrong with your set-up if there's enough RF on your gear to cause a problem".  

The problem of RF going somewhere other than out the antenna has been known for many decades and is sometimes referred to as "Hot Mic", a situation where there is enough RF on the radio - and the microphone - that the operator can even get an RF burn from touching the gear.  When this happens RF can get into the radio itself and cause undesired operation (malfunctions, distorted audio, etc.) but accessories connected to the radio - most notably sound interfaces, computers and even keyers - can be adversely affected.

While in the case above there was apparently some RF present on the gear to cause a problem, there isn't anywhere near enough to cause issues with the radio itself, and the radio+microphone (when running SSB) seemed immune.  Some types of antennas - typically ground-plane verticals, random-wires and end-fed half-wave antennas can, by their nature, put RF on the feedline - and thus the radios - unless extra steps are taken to minimize this problem in addition to properly installing/configuring the antenna, namely:

• Common-mode choke on the feedline.  Typically placed near the antenna, this usually consists of coaxial cable wound on a ferrite toroid - typically 6-12 turns on an FT240 or FT140 core with either Mix 31 or Mix 43 as the material - the latter being generally more useful/preferred for portable operations where the higher bands (40 meters and up) are most likely to be used.  Sometimes operators wish to have the feedline itself act as part of the counterpoise/ground - something that can risk a "hot mic" situation and in this case placing the common-mode choke farther along the coax - often near the radio - is the better choice.  (Some operators will put a choke at the antenna and near the radio.)
  

• Use of a "balanced" antenna.  A balanced antenna like a dipole is generally more likely to induce less RF current on its feedline than a purely end-fed antenna (a vertical is included) as it contains its own counterpoise - but having a perfectly-balanced antenna is not really possible and the feedline itself will usually participate in conducting/radiating RF along with the antenna to some degree.  A high-impedance antenna like an end-fed half-wave can sometimes reduce the probability of RF currents on the gear, but note that current can peak at every odd-numbered quarter-wave interval along the feedline and if the radio happens to be at one of these current nodes, issues are more likely to arise:  Placing a common-mode choke at a current node can help.
  

• Counterpoise/ground plane at the radio.  If you are operating in a metal vehicle it's less likely that RFI will be a problem as one is likely to be surrounded (e.g. shielded) - plus the fact that the shield of the coaxial cable feeding the antenna can be electrically bonded to its chassis.  Barring being in a Faraday cage like a vehicle, having a counterpoise connected at the radio (particularly if it's 1/4 wave long at the operating frequency - and if there is more than one of them) this can siphon off some of the RF that might be present owing to its lower impedance.  The use of a common-mode choke prior to the counterpoise at the radio will help to raise the impedance of the conducted RF and will usually improve the efficacy of a counterpoise/ground plane.

• Ferrites only go so far.  At HF, a simple "snap on" choke will probably do very little for the simple fact that there does not exist a common ferrite material that will offer a reasonable degree of choking impedance at, 14 MHz with just one turn (e.g. wire passed through it).  What is required is that multiple turns of a conductor be passed through the device (snap-on choke, toroid, etc.) as the impedance/inductance is proportional to the square of the number of turns.  Even so, there's a practical limit as to the choking impedance of a piece of wire around a ferrite (probably in the hundreds of Ohms for a "casually-wound" device).  As in the case of the keyer, I chose to use a capacitor, instead:  It is a tiny, inexpensive device able to fit inside the keyer rather than a large lump in a cable and it directly addresses the issue at hand by making the circuit intrinsically RF-tolerant.  In other words, it's the correct component for the job!
  

• Place the antenna far away from the radio.  As noted, this isn't always practical - or even desirable during portable operation.  In my opinion, equipment used with a radio transceiver should already have a modicum of resistance to stray RF energy so that even small/moderate amounts of RF on the gear will not cause any problems.
  

If you are operating portable, there's one thing that you probably aren't going to get very faraway from:  The antenna itself.  Almost by definition, portable operating implies being near the antenna owing to the need to have a feedline of manageable length and also due to practicalities of not wanting to lug a long feedline along or taking up more real estate than necessary.  What this means is that it's likely that you and your radio will be immersed in a rather strong RF field - and this also means that anything made out of anything that is conductive (the radio, power cables, microphones, interconnect cables to your paddle and keyer - and even you) are likely to intercept RF energy this will get into everything.

* * * * *

Footnote

1. A 1000pF capacitor has theoretical impedance of about 23 ohms at 7 MHz and it did the job here, but a 10000pF (e.g. 0.01uF or 10nF - ideally about 2.3 ohms at 7 MHz) capacitor would to just fine as well.  For positive keying (which is what likely what any modern radio uses) values as large as 0.1uF (100nF) would work as well - but this large of a value may cause issues with radios that use negative keying (e.g. high-impedance lines on some vintage radios).

 If you never plan to use a radio with negative keying, you could simply short together the source and drain leads of Q2 together to reduce RF susceptibility.

This kit is actually supplied with an "extra" capacitor:  The user can select between a 0.01uF (10nF) and a 0.047uF (47nF) capacitor (C3) on the "headphone" jack to set the loudness.  As I installed the 0.047uF capacitor, I had the 0.01uF left over.  Unfortunately, the specific capacitor supplied was thick enough that it prevent the board from sitting in the bottom of the case, raising it up and preventing the lid from fitting properly.  I could have probably connected this capacitor to the same circuit points on the top side of the board, but as I was home when I made this modification to my second keyer I simply found a lower-profile capacitor that didn't interfere with the board clearance.

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This page stolen from ka7oei.blogspot.com

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