Exploring the Ameco PCL-P Nuvistor cascode preamp/preselector

"And now, for something completely different!"

This past January - at Quartzfest - there was a table covered with "junque" and taped to it was sign with the word "FREE" on it.  That's how I ended up with this box.

Figure 1:
The front panel of the Ameco PCL-P preamp.  The left-hand
control tunes the front end of the preamp while the right-hand
control selects the "band".  The in/out switch is on the right.
Click on the image for a larger version.

The Ameco PCL-P

The PCL-P - which went on sale around 1965 - seems to have originally cost around $32.95 according to the RadioMuseum web page (link) - equivalent to around $300 today!  ^{Footnote 1}. The specifications say that it has about 20dB of gain and can be tuned for any frequency from 160 through 6 meters.

But what's it for?

Back in 1965 many amateurs still used separate receivers and transmitters - and it was often the case that this gear would, itself, be at least a few years old - likely WW2 surplus and/or gear from the 1950s.  Similarly, shortwave listening was still in its heyday and it's likely that many of the receivers used by SWLs (ShortWave Listeners) were also likely to be "vintage".

In those days, tube (e.g. "valve") based gear was still the rule and this - particularly for older gear (from the mid-late 1950s and earlier) - often meant several things were likely true about the receivers:

• Insensitivity on higher bands.  On the higher bands - namely 15-6 meters - it was often a struggle to attain good sensitivity at these higher frequencies.  This is particularly true on "simple" (e.g. inexpensive) where sensitivity would be fine on lower bands, but drop off precipitously with increasing frequency where signals were generally lower, anyway  Remedying this is surely the main purpose of this device.
  

• Image rejection may be marginal.  Most receivers of this vintage were single conversion - that is, they converted from the receive frequency to a lower-frequency IF (Intermediate Frequency) - typically around 455 kHz.  Some "fancier" receivers converted to something in the lower MHz range (often between 1.6 and 2 MHz) and then down-converted to something even lower - often in the 40-100 kHz range - where the final band-pass filtering was done. 
  

A device like the PCL-P might be touted as an aid to mitigate both of the above:  Its gain and low-noise amplification should help a "deaf" receiver and the fact that this device is somewhat selective may help the image problem as well - although that last point is debatable.

Whether or not a device like this was really helpful or not is strictly relevant to our discussion - rather, this article mostly is about the device itself.

Inside the PCL-P

Let's first take a look at the schematic diagram of PCL-P:

Figure 2:
Schematic of the Ameco PCL-P preamplifier.
Additional component annotations added to aid clarity of the description below.
Click on the image for a larger version.

First, notice S3a and S3b on the input/output terminals:  This allows the user to bypass the amplifier entirely - most useful when the unit is turned off - but note that this switch does not power down the unit when set to "out" (bypass) mode.  Immediately following S3a is S2a which is a rotary switch used to select the frequency range:  As can be seen from Figure 1, above, this switch has four overlapping frequency ranges:  1.8-4, 4-10, 10-23 and 23-54 Megacycles ^{Footnote 2}.

L1 is a coil (actually an autotransformer)  - tapped at 50 ohms - that covers the lowest frequency range (1.8-4 Mc) and is the large coil visible in Figure 3, below, but the higher bands' couplers - in the form of T1-T3 - are transformers (actually axially-wound coils with another winding over the top) clinging to the rotary switch, the turns ratios of the primary to secondary providing the impedance transformation from the 50 ohm input to the tuned grid circuit:  All of these, switched by S2b, connect to C1, an air variable tuning capacitor across the grid of the first of two vacuum tubes (valves), V1.

It's worth noting that the fact that this preamplifier is tunable is more of an artifact of the necessity of the technology used:  While it would, in theory, be possible to construct a "no tune" broadband amplifier to make its use slightly more convenient, but doing so - and maintaining equivalent performance over this wide frequency range - would have been a challenge.  The obvious advantage of making it tunable is that rather than amplifying the entire HF spectrum at once, its amplifying is limited to the vicinity of the frequency at which the input network is resonant meaning that by rejecting frequencies elsewhere, it's less likely to be overloaded by RF energy that is well away from the frequency of interest (e.g. strong shortwave broadcast stations on other bands).

There are two identical tubes here - 6DS4s in the case of my preamp  (other units may have been equipped with the similar 6CW4) and these are Nuvistor tubes:  About the size of a very large pencil eraser, these were some of the smallest vacuum tubes that were mass-produced - most Nuvistors being triodes like V1 and V2, above.  Being very small, they were well-suited for high frequency operation, finding their way into the UHF tuners of many contemporary televisions:  It was at about the same time as this unit was made that U.S. Federal law mandated the inclusion of UHF tuners on all new TVs so Nuvistors were widely available and comparatively inexpensive owing to the economy of mass production.  (Wikipedia article about Nuvistors - link).

Figure 3:
Top view of the Ameco PCL-P chassis, the variable capacitor
visible near the upper-right, L1 the big coil in the center and
the two Nuvistors visible just below/left of center.
While this was originally equipped with RCA (phono)
 "Motorola" type connectors,
it has since been retrofitted with BNCs.
adsfadsf
Click on the image for a larger version.

To some, the connection between V1 and V2 may look a bit odd, but the description on the front panel (seen in Figure 1) gives a clue:  They are connected in cascode configuration - possibly a portmanteu for "cascaded triode/pentode" or similar.  In this configuration the "bottom" tube (V1 in this case) gets its plate voltage via the cathode of the "upper" tube (V2) - but you might notice something else:  The grid of V2 is at RF ground via C3 - being somewhat neutrally biased at DC by R2 which allowed current from V2's plate to get to V1's plate via V2's cathode.

This cascode circuit has a distinct advantage for higher frequencies:  As the current through V2 (effectively running in "grounded grid" configuration) is somewhat proportional to its grid-cathode voltage, when V1 conducts more - trying to pull the cathode of V2 lower - V2 conducts harder in response.  As V2's grid is "grounded" at RF via C3, pulling its cathode lower effectively increases the grid-to-cathode voltage:  V2 also tries to counter this by conducting more, trying to pull the cathode back up.  Because of this arrangement, the voltage on V2's cathode (and, of course, V1's plate) changes relatively little compared to the change in current through it.

What this means it that the effect of Miller capacitance is minimized ^{Footnote 3}.  Here we are concerned with the capacitance between the grid and plate of the tube - V1 in this case - and this capacitance couples the two together lightly, but this has the bad side effect of somewhat cancelling out the tube's amplification action:  As the grid voltage tries to go up with the input signal, the plate voltage would - in a typical single-tube circuit - go down by a comparatively large amount as the tube conducts more in response - and the capacitance between the two will cancel out the signal on the grid to a degree:   This is one of the reasons why it can be difficult to get a single-device vacuum tube RF amplifier to work well at high frequencies.  If we prevent the plate voltage from changing as much and convey the signal more as current instead - as we are doing with the action of V2 in this cascode circuit- we can significantly reduce the Miller effect. 

Figure 4:
The underside of the PCL-P chassis prior to repair - the 2-
section yellow capacitor and the diode on the left.
Click on the image for a larger version.

With the cascode configuration, the swing of the plate voltage of V1 is minimized - and so is the Miller effect resulting in better gain, flatter frequency response and potentially, lower amplifier noise overall.  As such, we get varying current on the plate of V2 which, via transformer T4 (visible on the far right in Figure 4 as several turns of enameled wire on what appears to be a threaded, ferrite transformer core) is coupled to the output.  Resistor R3 was likely added to help ensure stability of the amplifier both when it is being bypassed (the input and output having nothing connected to either) and also in the event that the input impedance of the receiver connected to the (un-tuned) amplifier output is a poor match at some frequencies.

The rest of the circuit is a pretty straightforward power supply:  The PCL-P used a silicon diode (D1) to half-wave rectify the plate supply, filtered first by C8 - the neon power-on indicator (V3) is connected to this point via R5 - and then decoupled by 1k resistor R4 and filtered again by C9:  The ultimate result is a nice, clean source of about 145-155 volts for tubes when this is operated from a modern 123 volt U.S. mains source ^{Footnote 4}.

Construction quality

I'd say that the Ameco PCL-P is constructed "well enough":  It looks as though a bit of thought and refinement occurred to assure stable operation at 6 meters - a frequency range that was above what the average amateur of the mid 1960's had for equipment - while maintaining low cost and simplicity.  A nice touch is the use of a feedthrough capacitor (C4) as a component mounting point/stand-off (not actually "feeding through" the chassis, though) and bypass for the plate supply feeding the bottom of the output transformer, T4:  This is surely the one place where the use of a somewhat expensive component was absolutely necessary as a lowly disc ceramic would probably not have sufficed owing to the comparatively high ESR and self-resonant properties that type of capacitor.

From what I can tell, the PCL-P was originally fitted with "RCA" (phono) "Motorola" type connectors (like those found on car radios) on the input/output - a somewhat common practice on HF, VHF and even UHF amateur and commercial radios - but they have clearly been replaced with the more-common BNC types by a previous owner.

Refurbishing

Figure 5:
This time, with a new diode and capacitors on the left.
Output transformer T4 is visible near the right edge,
supported by feedthrough capacitor C4.
Click on the image for a larger version.

Although I don't really have any intention to put this device into regular service, I did want to get it into operational condition.

Carefully powering it up on a current-limited mains supply, I noted that the dual-section power supply capacitor (C8/C9 - in the same yellow tube visible in Figure 4) was bad with about 10 volts ripple on the plate supply - but I was able to verify that the unit had good gain, indicating that both of the Nuvistor tubes were working properly despite receive signals being overlaid with 60 Hz "hum".

As the line cord was in very good shape the only thing I had to replace was the yellow dual-section capacitor (C8/C9) with individual 22uF, 200 volt units (partly to accommodate the somewhat higher mains voltage these days) - but I also replaced the diode (D1) with a more modern 1N4007 with a 1kV rating.  Ultimately, the ripple on the plate supply was well under a volt - as it should be!  (Sharp-eyed readers may have noticed that the PCL-P is sitting atop the defunct filter capacitor in Figure 1.)

Not surprisingly, I noted that the transformer in this amplifier "buzzed" quite a bit - but with a half-wave, capacitor-input rectifier conducting on the peak of every half-cycle, this isn't unexpected:  The addition of a resistor (say, 100-470 ohms) in series with the diode (D1) would probably reduce this by limiting the peak current on the top of the AC waveform.

Performance

It's worth noting that any amateur receiver made by a major manufacturer since the 1980s - when it is working correctly - will very likely have more than adequate sensitivity on all bands to hear the local receive noise floor, so the PCL-P amplifier probably has little place in the modern ham shack - but for a "deaf" radio from the 1950s and 1960s, of which there were many - particularly if they were in need of alignment - it would have likely been useful.

The one place where this unit might be useful in the modern ham station - if only for nostalgic purposes - might be for a low-gain wire antenna (e.g. Beverage-On-Ground, Loop-On-Ground or Loop-Under-Ground) for the 160 and 80 meter band.  Nevertheless, I decided to check the gain and selectivity of this device in the (non-WARC) amateur bands 160 through 6 meters:  I have included these plots and comments below the conclusion of this article.

According to the official specifications of this amplifier, its gain is about 20dB - and my measurements - with 50 ohms in/out - corroborate this, more or less:  At 10 and 6 meters it fell slightly short of this figure, but not dramatically so and this variance can be forgiven given the vagaries of manufacturing differences and age.  It's worth noting that the 6DS4 triodes used in this copy have a very slightly lower rated gain than the nearly-identical 6CW4 triodes (an amplification factor 63 versus 65) that the schematic notes as an alternate, but the difference would likely be negligible in the real world as the in-circuit gains would surely be much lower - or in the case of this amplifier, it's around 20dB (e.g. voltage amplification factor of 10 and a power gain of 100).

Unfortunately, I don't have a means of accurately measuring the noise figure, but testing with a "modern" radio (an FT-817) across HF and 6 meters indicates that this amplifier is NOT noisier than the FT-817 implying that its noise figure is at least as good as it needs to be to be able to hear above the atmospheric noise level - even in an RF-quiet environment.  These Nuvistor tubes are capable of a noise figure of as low as 3dB on 6 meters, but mismatch and losses in the input (and, to a lesser extent, the output) networks would surely degrade this - but a noise figure of only about 9 dB  ^{Footnote 5} is likely to be sufficient in 6 meter work for anything other than, perhaps, EME (Earth-Moon-Earth).

Above, I touched briefly on the idea of IF image rejection being slightly improved by a device like this that offers a bit of band-pass filtering:  With a single-stage L/C filter, any improvements afforded by it are likely significant only at the lowest frequencies where the width of the peak is at its narrowest - but negligible on the  higher bands as noted in the comments below the response plots.

Conclusion

As noted earlier, the PCL-P Nuvistor preamplifier is probably not a useful addition to a modern-day ham shack with radios made since at least the 1980s:  The issue that it solves - notably that of addressing the lack of sensitivity of some older radios on the higher bands - is simply a "non problem" these days.  If you have some old "boat anchor" radios - particularly of the less-expensive variety - this sort of device may help pick up weak signals - particularly on a mostly "dead" band.

The noise floor of this preamplifier appears to rival that of a modern radio - but this doesn't mean that it would improve the sensitivity of a such a radio, but only that it would simply make the S-meter read higher without improving the signal-to-noise ratio:  If a radio in question can already hear the noise floor on a given band when connected to your antenna, further amplification will not improve absolute sensitivity and may simply degrade receiver performance by feeding it with too much signal!

As it is, this unit will sit on a shelf with some other "vintage" gear, always ready for some possible future use.

* * *

Footnotes:

1. If you think about this for just a second, you can buy some really nice accessories for $300 these days such as an automatic antenna tuner, a low-end laptop, or even one of several very nice QRP radios - some of which are software-defined radios.  How times have changed!
   
2. Until somewhere around 1970 or so, it was common - at least in the U.S. - to use "cycles" (e.g. Cycles per second) rather than Hz (Hertz) which is why older equipment may show "kc" (kilocycles) and "Mc" (Megacycles) rather than the modern "kHz" (kiloHertz) and "MHz" (MegaHertz), respectively.  And no, you don't need a special "Mc to MHz" converter to use your old receivers!
3. As noted, the Miller capacitance is often a limitation on the performance of high-frequency/high speed electronic components which is why the cascode configuration is used - and a similar reason why transimpedance amplifiers are the norm for interfacing with photodiodes in high-speed optical detectors  The Wikipedia article on the Miller effect is here:  link.   
4. When this unit was made the nominal residential mains voltage in the U.S. was closer to 110-115 volts and now it is more typically in the 120-125 volt range.  It's unclear when this (gradual) change occurred - and it didn't seem to happen everywhere in the U.S. all at once - but the shift from "about 115" to "around 125" likely happened over the period of the mid 1960s into the 1980s.  "Vintage" gear - that being from the 1960s or earlier - likely was designed to operate closer to 110 volts (especially devices from the 1940s and earlier) than 120 volts meaning that the supply voltages (filaments, B+, etc.) are going to be higher as will the magnetization current/losses in the transformers - something to consider if you routinely operate such gear:  The use of a Variac ^TM or a "buck" transformer in series (e.g. an out-of-phase 9-12 volt filament transformer wired to reduce the 120 volt mains) is suggested to prevent overvoltage of filaments, capacitors, transformers, etc. to maximize the lifetime of those components.
   
5. The article "Measurements on a Multiband R2Pro Low-Noise Amplifier System, Part 2" by Gary Johnson, WB9JPS, discusses the effects on noise figure on real-world performance and concludes that a receive system noise figure of 9dB is likely to be adequate for typical 6 meter operation:  Link (from the Web Archive)
   

 * * * * *

Frequency response plots of the Ameco PCL-P preamplifier/preselector

The following plots were taken using a DG8SAQ VNA with 20 dB of attenuation on its "Output" port (connected to the input of the PCL-P) and 6 dB of attenuation on its "input" port (connected to the PCL-P's output) to prevent overload of both the preamplifier and the VNA as well as present a nice, resistive 50 ohm source and load impedance.  (Ignore the S11 and Smith plots as I forgot to turn them off).  These plots cover the range from 1 through 80 MHz, overlapping all of the HF bands (plus 6 meters).  I did note that all of these bands overlap slightly, leaving no "gaps" in coverage and as expected, the gain and the "sharpness" of the filtering in these overlap areas (e.g. top end of the lower band with the tuning capacitor near minimum and the bottom end of the next higher band with the capacitor near maximum) were slightly different:  None of the amateur bands tested below fell  entirely within an "overlap" area.

For the response plots there is a marker (#2) indicating the center (peak) frequency while other markers indicate the -10dB and -20dB responses (relative to the peak) - the numbers in the upper-left corner indicating the forward gains at those frequencies.

The final plot shows the insertion loss of the unit when the "in/out" switch is set to "out" (bypass).

Click on any of the plots below for larger version.

Tuned to 1.9 MHz (160 meters) in the 1.8-4.0 MHz position, the peak gain being about 23dB.  The preselector does a decent job of rejecting a possible IF image (910 kHz above the center frequency for a 455 kHz IF).  Note also that the input preselector does a decent job of attenuating much of the AM broadcast band - although it might still be overloaded by a local transmitter operating near the top end of that band.

Tuned to 3.7 MHz (80 meters) in the 1.8-4.0 MHz position, the peak gain being a bit short of 28dB.  On 80 meters and higher there is only minimal image rejection for 455 kHz IF radios.

Tuned to 7.2 MHz (40 meters) in the 4-10 MHz position, the peak gain being just under 24dB.

Tuned to 14.2 MHz (20 meters) in the 10-23 MHz position, the peak gain being just under 23dB.

Tuned to 21.2 MHz (15 meters) in the 10-23 MHz position, the peak gain being just under 23dB.  At these higher bands the limitation of the simple, single-stage L/C filter starts to show up as an asymmetrical response - the filtering above the center frequency being less effective that below it.  Note also that at the marked 20dB point above the center frequency (marker #5) the gain of the amplifier is still about 2dB!

Tuned to 28.5 MHz (10 meters) in the 23-54 MHz position, the peak gain being just under 19dB.

Tuned to 52 MHz (6 meters) in the 23-54 MHz position, the peak gain being just a bit more than 19dB.  Its worth noting that the input network does appear to attenuate signals in the FM broadcast band by more than 20dB - something that may have been useful for receivers that suffered from ingress from a strong, local transmitter.

The "through" loss when switched to bypass ("out") mode.  Loss is measured at 0.53dB at 53.5 MHz and 0.16dB at 28.1 MHz as indicated by the markers.

This page stolen from ka7oei.blogspot.com

 [END]

Reducing RF susceptibility for the HamGadgets "Ultra Pico Keyer" - and mimizing RF issues on portable HF stations in general

Note:

While this article describes a modification of the Pico Keyer to reduce RF susceptibility, it also talks about methods to minimize/reduce RFI-related issues in general for both portable and "base" stations:  This specific topic is covered near the end of this blog entry.

POTA operation 

Over the past several years I've done a bit of POTA (Parks On The Air) operating, racking up "about" 1000 contacts as an activator in a number of parks - usually as an "activator", and mostly on CW.  Typically, I have operated from a campsite using a portable antenna - usually the JPC-7 loaded dipole (discussed in this blog entry) or the JPC-12 loaded vertical (discussed here) - but I have also used an end-fed half-wave and a simple dipole on occasion - and even the Yaesu ATAS-100 on my vehicle.

Figure 1:
Operating CW POTA from US-0004,
Arches National Park in Utah
Click on the image for a larger version.

In the recent past there has been a revolution in portable power sources in that a LiFePO4 battery - which can supply 20-ish amps - is both light enough to be practical and fairly inexpensive.  For those instances where I may be staying at one location for several days the advent of inexpensive solar to maintain the power budget - and the solar controllers can be made to be RF quiet to make it compatible with HF operation (see this article).  With this in mind it's practical to operate the transmitter at 100 watts much of the time, something that makes it as easy as possible for those who wish to work me.  Despite the ability to run 100 watts, I have occasionally operated QRP (5 watts or less) - again, usually on CW.

A memory keyer

Having used a number of different radios for POTA operation (Yaesu FT-100 and FT-817, Icom IC-706MK2G and even a RockMite) - none of them with a memory keyer - I decided that an "Upgrade" was in order so I got the Ham Gadgets "Ultra Pico Keyer" (Link here).  This device is small, powered by a single CR2032 lithium coin cell and costs about US$40 as a kit (not including shipping) including a 3-D printed case.  For portable use, I couple it with the "Outdoor Pocket Double Paddle" (with magnets!) from CW Morse (link).

This is a nice, little device in that it provides a consistent interface to the user - no matter which radio you might use - and it has a number of message memories (up to eight), perfect for an activity like POTA where a message (e.g. "CQ POTA") may be repeated many, many times during the course of the operation.

Getting "stuck" 

Figure 2:
The Ham Gadgets "Pico Keyer" (left) along with the
CW Morse Outdoor Paddle.
Click on the image for a larger version.

While the Ultra Pico Keyer works as advertised, I did notice a problem on the first trip out while using a portable antenna:  It would get "stuck".

Clearly, this was an RF susceptibility issue - verified by reducing transmit power and observing that it no longer happened.  In short, at 5 watts there was usually no issue, but at 100 watts  the radio would stay keyed continuously after the first Morse element whether it was sent from a stored message or via the paddle:  While it was "stuck", I could still hear the sidetone - via the keyer's internal speaker - sending the message or what was keyed via the paddle indicating that it was not the microcontroller that had crashed but the circuitry that keyed the radio that was the problem.

Further testing showed that when the unit got "stuck" due to RF and simply unplugging the paddle from the back of the keyer would cause it to release (get "un-stuck").  The fact that this happened using a portable antenna provided further evidence of potential RF sensitivity.

Analyzing the problem

As I'm wont to do, I decided to take a look at the Pico Keyer's schematic to see if there was something about its design and construction that might make it more susceptible to RF interference - and I was surprised at what I found.  Here's the diagram found in the manual that is freely available online on the web site (link):

Figure 3:
Annotated diagram of the Pico Keyer with RF current paths shown.
The components in question are Q1 and Q2, in the upper-right corner.  The lines highlighted in yellow are those through which RF currents will flow (between the radio chassis and the paddle/cable) if no bypass capacitor is installed. 
The added capacitor is shown below the "OUTPUT" jack near the upper-right with the RF current path around Q1/Q2 shown in magenta.
Click to get a larger image.

While there are protection capacitors on the paddle input (C1, C2) my eye was immediately drawn to the output keying (upper-right) where I was, at first, confused as to the arrangement with an N-channel MOSFET in both the keying line and the "common" (ring) of the "OUTPUT" connector (e.g. Q1 and Q2) - but then I remembered that the manual stated that this device would key both positive and negative voltages, explaining the "unusual" arrangement.

While admittedly clever, I could immediately see a susceptibility issue here - the problematic RF current path highlighted in yellow in Figure 3, above:  The "OUTPUT" jack more or less will "float" compared to the "ground" of the keyer itself, which is also connected to the "ground" lead of the cable to the paddle as well as the external paddle itself.  This configuration almost guarantees that there will be at least some RF current flowing from the radio and through the keyer's output circuit for several reasons:

• If you are using this in a portable situation, the radio will surely have some RF on its chassis.  As noted in the final section of this blog entry, it's almost impossible to prevent all RF current from getting onto the feedline - even if you do use a common-mode RF choke and a very nearby antenna is likely to immerse the radio and its interconnecting gear in a rather strong radio-frequency field.

• The paddle and the cable that connects it to the keyer should be considered as part of an antenna - and this situation is made worse if one is sitting at, say, a metal table and also if you, the operator, place your hand at/near the paddle/cable, further encouraging a "through" path for RF.
  

What this means is that there will be at least some RF current flowing from the radio chassis, through the keyer and then, as indicated by the yellow-highlighted lines - via transistor Q2 (and Q1) and then through the cable to the paddle.  I didn't really investigate the exact mechanism by which RF current through this path was causing the keying line to get "stuck" - but here are a couple of possibilities.

• RF may be coupling from the drain of Q2 into its gate - and subsequently into Q1's gate as well, which is tied in parallel with it with the peaks of the RF voltage turning on the FET.  Even if RF through the FET was causing it to conduct only on half of the RF cycle, this would surely be enough to key the radio.  It's also possible that the transistor was turned, on average, only "partially" on by the RF energy - not enough to completely shunt out the RF, but enough to key the radio.

• The RF could also be getting into the output pin of the microcontroller via the FET, causing its totem pole output to get "stuck" on while it was present.

Figure 4:
The added capacitor(s) can be seen soldered between the
"sleeve" pins of the "OUTPUT" and "PADDLE" jacks,
on the bottom of the board.  As you can see, I've made this
modification to both of my Pico Keyers!
Click on the image for a larger version.

Regardless of the cause, the fix was clear:  Add a capacitor to bypass RF current around Q1 and Q2 and the output pin of the microcontroller.  In Figure 3, above, the magenta highlight shows how the added capacitor conducts RF currents around the sensitive components.

When this occurred, I happened to be on a POTA activation, but I had my "electronic toolbox" in the car which included a number of useful items such as a soldering iron and a smattering of useful electronic components (a some common resistors, capacitors, etc.).  Grabbing a 1000pF capacitor, I connected one end to the "sleeve" (ground) pin of the "PADDLE" jack and the other end to the "sleeve" of the "OUTPUT" jack - effectively providing a bypass to RF energy on Q2's drain to the circuit "ground" to eliminate any RF voltage potential between the cable connecting the radio and that going to the paddle.  

This modification completely solved the problem:  It is my opinion that this capacitor should be supplied with the kit.  Additionally, Q2 could be eliminated completely and the source/drain leads jumpered if negative keying is not needed.  ^{See Footnote 1}

Since the topic of "RF on the rig" was already broached, the rest of this article will describe how to reduce it.  It's worth noting that the susceptibility of the memory keyer was such that even with the measures described below, it was affected at 100 watts.

* * * * *  

Suppressing RF on the gear and connecting cables

Some readers of this may immediately say "You are obviously doing something wrong with your set-up if there's enough RF on your gear to cause a problem".  

The problem of RF going somewhere other than out the antenna has been known for many decades and is sometimes referred to as "Hot Mic", a situation where there is enough RF on the radio - and the microphone - that the operator can even get an RF burn from touching the gear.  When this happens RF can get into the radio itself and cause undesired operation (malfunctions, distorted audio, etc.) but accessories connected to the radio - most notably sound interfaces, computers and even keyers - can be adversely affected.

While in the case above there was apparently some RF present on the gear to cause a problem, there isn't anywhere near enough to cause issues with the radio itself, and the radio+microphone (when running SSB) seemed immune.  Some types of antennas - typically ground-plane verticals, random-wires and end-fed half-wave antennas can, by their nature, put RF on the feedline - and thus the radios - unless extra steps are taken to minimize this problem in addition to properly installing/configuring the antenna, namely:

• Common-mode choke on the feedline.  Typically placed near the antenna, this usually consists of coaxial cable wound on a ferrite toroid - typically 6-12 turns on an FT240 or FT140 core with either Mix 31 or Mix 43 as the material - the latter being generally more useful/preferred for portable operations where the higher bands (40 meters and up) are most likely to be used.  Sometimes operators wish to have the feedline itself act as part of the counterpoise/ground - something that can risk a "hot mic" situation and in this case placing the common-mode choke farther along the coax - often near the radio - is the better choice.  (Some operators will put a choke at the antenna and near the radio.)
  

• Use of a "balanced" antenna.  A balanced antenna like a dipole is generally more likely to induce less RF current on its feedline than a purely end-fed antenna (a vertical is included) as it contains its own counterpoise - but having a perfectly-balanced antenna is not really possible and the feedline itself will usually participate in conducting/radiating RF along with the antenna to some degree.  A high-impedance antenna like an end-fed half-wave can sometimes reduce the probability of RF currents on the gear, but note that current can peak at every odd-numbered quarter-wave interval along the feedline and if the radio happens to be at one of these current nodes, issues are more likely to arise:  Placing a common-mode choke at a current node can help.
  

• Counterpoise/ground plane at the radio.  If you are operating in a metal vehicle it's less likely that RFI will be a problem as one is likely to be surrounded (e.g. shielded) - plus the fact that the shield of the coaxial cable feeding the antenna can be electrically bonded to its chassis.  Barring being in a Faraday cage like a vehicle, having a counterpoise connected at the radio (particularly if it's 1/4 wave long at the operating frequency - and if there is more than one of them) this can siphon off some of the RF that might be present owing to its lower impedance.  The use of a common-mode choke prior to the counterpoise at the radio will help to raise the impedance of the conducted RF and will usually improve the efficacy of a counterpoise/ground plane.

• Ferrites only go so far.  At HF, a simple "snap on" choke will probably do very little for the simple fact that there does not exist a common ferrite material that will offer a reasonable degree of choking impedance at, 14 MHz with just one turn (e.g. wire passed through it).  What is required is that multiple turns of a conductor be passed through the device (snap-on choke, toroid, etc.) as the impedance/inductance is proportional to the square of the number of turns.  Even so, there's a practical limit as to the choking impedance of a piece of wire around a ferrite (probably in the hundreds of Ohms for a "casually-wound" device).  As in the case of the keyer, I chose to use a capacitor, instead:  It is a tiny, inexpensive device able to fit inside the keyer rather than a large lump in a cable and it directly addresses the issue at hand by making the circuit intrinsically RF-tolerant.  In other words, it's the correct component for the job!
  

• Place the antenna far away from the radio.  As noted, this isn't always practical - or even desirable during portable operation.  In my opinion, equipment used with a radio transceiver should already have a modicum of resistance to stray RF energy so that even small/moderate amounts of RF on the gear will not cause any problems.
  

If you are operating portable, there's one thing that you probably aren't going to get very faraway from:  The antenna itself.  Almost by definition, portable operating implies being near the antenna owing to the need to have a feedline of manageable length and also due to practicalities of not wanting to lug a long feedline along or taking up more real estate than necessary.  What this means is that it's likely that you and your radio will be immersed in a rather strong RF field - and this also means that anything made out of anything that is conductive (the radio, power cables, microphones, interconnect cables to your paddle and keyer - and even you) are likely to intercept RF energy this will get into everything.

* * * * *

Footnote

1. A 1000pF capacitor has theoretical impedance of about 23 ohms at 7 MHz and it did the job here, but a 10000pF (e.g. 0.01uF or 10nF - ideally about 2.3 ohms at 7 MHz) capacitor would to just fine as well.  For positive keying (which is what likely what any modern radio uses) values as large as 0.1uF (100nF) would work as well - but this large of a value may cause issues with radios that use negative keying (e.g. high-impedance lines on some vintage radios).

 If you never plan to use a radio with negative keying, you could simply short together the source and drain leads of Q2 together to reduce RF susceptibility.

This kit is actually supplied with an "extra" capacitor:  The user can select between a 0.01uF (10nF) and a 0.047uF (47nF) capacitor (C3) on the "headphone" jack to set the loudness.  As I installed the 0.047uF capacitor, I had the 0.01uF left over.  Unfortunately, the specific capacitor supplied was thick enough that it prevent the board from sitting in the bottom of the case, raising it up and preventing the lid from fitting properly.  I could have probably connected this capacitor to the same circuit points on the top side of the board, but as I was home when I made this modification to my second keyer I simply found a lower-profile capacitor that didn't interfere with the board clearance.

* * * * *

This page stolen from ka7oei.blogspot.com

[END]

A 15 (and 10) meter high-pass filter for Field Day

QRM from a transmitter to receivers on lower bands

[Go to the end of this article for an "After-Field-Day follow-up" about this filter] 

A friend of mine belongs to a club in a town north of me and he was describing an issue that they've been having for the past several years during ARRL Field Day:  A station on an upper band (e.g. 15 or 10 meters) degrading reception on 20 or even 40 meters when transmitting.  What was needed was something that could be used on both 15 and 10 meters and protect the lower bands (e.g. 20, 40 and 80) meters - and this protection would go the other way, preventing the 15/10 meter station's receiver from being overloaded by transmissions on the lower bands.

Figure 1:
Exterior of the 15 Meter high-pass filter,
built into a die-cast aluminum box with single-hole
UHF connectors on the sides.
Click on the image for a larger version.

First, a bit of background.

The ARRL Field Day event is held on the fourth (but not last) full weekend every June.  During this event thousands of clubs and individuals go forth into the wilds to set up and operate an event where they attempt to contact as many stations as they can in a 24 (or 27) hour period.  In the case of club stations - or where multiple individuals are involved - it's very common to have more than one transmitter at a given site.

As the Field Day rules stipulate that all antennas/radios be within a 1000 foot (305 meter) circle it isn't possible to provide much geographical separation between different transmitters.  This separation is important because a transmitter produces a very strong signal and the received signals are very weak by comparison:  Receivers can be easily overloaded by these nearby strong signal sources and transmitters can produce low-level signals on frequencies other than those on which they are operation - ones that are too weak to cause problems under normal situations but when placed in close proximity to a receiver these weak emissions can block out/interfere with other receivers - even on different frequency bands.

Other-band signals can cause problems

The degree to which a transmitter radiates these low-level spurious signals - and that to which a receiver is able to tolerate a very strong signal - depends considerably on the transmitter/receiver itself.  Some high end makes of radios (e.g. Elecraft, Flex ^{Footnote 1}) can be very clean in terms of transmitted spectra and higher-end receivers of all makes may be capable of tolerating a very strong signals - perhaps even in the same band - and this strategy works as long as the potentially-interfering transmitter itself is clean:  If that "other" transmitter is producing noise at any frequency of reception, there's nothing that can be done at that receiver to fix the problem other than to quiet or clean up the errant transmitter.  Meanwhile, even a "good" radio - such as an Icom IC-706MK2G or IC-7300 ^{Footnote 2} or a Yaesu FT-757 - which works well by itself - may not "play nice with others" when immersed in an environment with multiple transmitters and receivers in very close quarters for reasons largely related to their design architecture.

What this means is that if one uses directional antennas (e.g. Yagis or beams) they are often placed north-south of each other and pointed parallel  ^{Footnote 3} so that there is some isolation off to the sides of these antennas - and it goes without saying that antennas of any sort are separated as far as the rules - or the operating space (e.g. park, yard, forest clearing) allows.

Sometimes, this isn't enough:  Interference can result despite the precautions (e.g. transmit/receiver separation) so additional filtering may be necessary.

The use of Band-Pass filters

Barring the ability to separate antennas or place them in each others' nulls, there are other options:  From a number of manufacturers ^{Footnote 4} there are available band-pass filters that - as the name implies - are designed to pass one specific amateur HF band with low attenuation (loss) while offering significant rejection of other bands above and below.  By placing one of these filters inline with the radio and the antenna, it not only will reduce the probability that a very strong signal from another band might overload the receiver (a particular problem with a radio like the Icom IC-7300 and certain models of other radios from other manufacturers) but it also attenuates the broad-band noise ^{Footnote 5} that almost all HF radios produce that can encompass frequencies other than the band on which they are operating.

This low-level interference - often in the form of a white noise (or hiss) is produced by the amplifier stages in the transmitter itself.  Most modern HF transceivers - while equipped with low-pass filters that attenuate harmonics at multiples of the transmitted signal and generally prevent this noise from being emitted on the next-higher non-WARC band - do NOT have an equivalent high-pass filter in them that prevents low-level spurious signals or broadband noise from being output to the antenna on frequencies below that on which it is operating.  What this means is that a transmitter operating on, say, 15 meters, can produce a "hiss" that may degrade reception on 20, 40 or even 80 meters whenever it is keyed up - in this example, depending on how well that 15 meter antenna can radiate such signals and how close the two antennas are to each other.  (I discussed this very problem in an early article of this blog:  Getting the rigs ready for Field Day - Link).

For this reason it is often preferable to use a band-pass filter on every transmitter that is used, for the specific band on which it will be operated:  This will not only protect that receiver from the other bands' signals but also prevent the low-level energy from being emitted on bands other than that on which it is being used.

Note, however:  If you are experiencing interference from another transmitter that is actually producing noise on your receive frequency (this, given the presumption that your receiver isn't being overloaded) the addition of a filter on the station receiving interference will do no good at all since it cannot possibly filter out interference that is already there, on-frequency.  In this case, the only appropriate remedy would be to filter the offending transmitter.

As an aside:  In some cases simply enabling the radio's built-in antenna tuner - or using an external tuner - may significantly reduce the amount of out-of-band energy that the transmitter emits as well as adding to the attenuation from "other-band" signals during receive. ^{Footnote 6} 

A high-pass filter

While band-specific filters are preferred, my friend presented a case where a high-pass filter (one that blocks signals below a certain frequency) may be appropriate.  In his Field Day environment there has always been a station operating on 20 meters and usually another operating on 40 meters as well - but a third station was available to operate on 15 or 10 meters - depending on propagation conditions.  The problem was that when this third station transmitted, 20 and 40 meters were often degraded - likely by the broadband noise mentioned earlier.

While it would be possible to obtain separate 15 and 10 meter band-pass filters at some expense, I decided on a different approach:  A 15 meter high-pass filter.  This filter - which could be made to strongly attenuate frequencies on the non-WARC amateur bands below 15 meters (e.g. 20, 40 and 80 meters) - it would have the advantage of also being usable on both 15 and 10 meters.  Since it was unlikely that they would have stations on both 15 and 10 meters this strategy seemed sound for their application.

Using the ELSIE program (from Tonne software - link), I first calculated an "N=5" pro-forma high-pass filter using the "Elliptical" (e.g. "Cauer") circuit topology observing that I could get low attenuation at 15 meters and above while achieving more than 40dB on 20 meters and below.  Using the ability of the ELSIE program to do Monte-Carlo type optimizations, I then tweaked the filter topology from a pure Elliptical filter to a hybrid one and this resulted in even better attenuation at 40 meters than the original:  The schematic diagram of this filter is shown below:

Figure 2:
The 15 meter high-pass filter as iterated by the ELSIE program.  The circuit topology - originally
Cauer (Ecliptic) was modified by using simple inductors in sections 1 and 5 and a capacitor at
position 6 and then re-iterated to optimize performance.
Click on the image for a larger version.

As can be seen from the diagram, there are three capacitors in series with the signal path with two inductors directly to ground:  The center inductor is in series with another capacitor, forming one of the "notches" typical of the Elliptical filter topology - and it so-happens that it's possible to tweak the filter so that this notch just happens to land in the middle of the 20 meter band to maximize attenuation there.

Rummaging around in my junk box I found several 500 volt silver-mica capacitors:  For some reason I have a lot of 160pF units, so that was placed at section #2 and three of them were put in parallel for the series capacitor in section #3.  I found a 200pF capacitor for section #6 and I paralleled a 120pF silver mica and an NP0 disc ceramic for that in section #4.

Many people doing homebrew construction seem to intensely dislike toroidal inductors - but while they would be more compact, there is no need to use them here, so large-ish air-core inductors were used.  As the inductors are all "about" the same value (in the 225-300nH range) I wound 7 turns of 17 AWG (but anything 14-18 AWG would do) wire on a 13/32" drill bit for each of them, the precise value being unimportant as their turns would be stretched/compressed while using a VNA to "dial in" the filter response.  As mentioned earlier, I'd added one more inductor from the initial design because the inductors were the cheapest of all of the components (they are just wire!) and they are very adjustable - simply by compressing/spreading the turns which meant that by picking capacitor values that were just "pretty close" to those called out by ELSIE, the coils could be used to tweak the filter's response.  Note in Figure 3 that the inductors that are close-ish to each other are placed at right-angles, or in parallel with each other:  Avoid placing two adjacent coils "end to end" with each other to minimize coupling between them.

Figure 3:
Inside the 15 meter high-pass filter.  Copper-clad PC board
material is used as the backplane (ground) with small pieces
used as "islands" for connection and support points.  The
added capacitor and inductor are those on the right-hand side.
Click on the image for a larger version.

The filter was built on a piece of copper-clad PC board material as a back-plane and ground and small pieces of that circuit board material were cut out to form "islands" - the so-called "Manhattan" construction:  These islands would allow the junctions of the various circuit components to be connected together and with these islands glued to the back-plane and mechanically support the components soldered to them.

The piece of circuit board used as the backplane was sized to fit in the bottom of a die-case aluminum box that I had handy (about 6" x 3.25" x 2" or approx. 15 x 8.3 x 5 cm - but it could have been a bit smaller) onto which I'd installed two chassis-mount UHF connectors:  These connectors were placed rather close to the bottom of the box so that their ground lugs could be soldered to the backplane, providing both the "ground" connection to the copper clad and for mechanical support

Using a VNA, I first adjusted the inductor in section 3 to provide a notch at about 14.24 MHz and then iteratively tweaked the inductors in sections 1 and 5 to provide the lowest insertion loss and lowest VSWR at 15 and 10 meters.  When I was done, the insertion loss was just fine - less than 0.5dB - but the VSWR was about 1.45:1 at 15 and 10 meter so I added two more components (the 30pF capacitor in section 7 and the inductor in section 8 of the diagram) to act as a bit of a "tuner" to improve the match:  In the figure above you can see an inductor (in section 8) that goes to the right-hand UHF connector (6 turns of the same wire as the other coils on a 13/32" drill bit) and a 30pF disk-ceramic capacitor (section 7) between the PC board "island" to which it connects and ground:  With a bit more adjustment of all four inductors this brought the VSWR at 15 and (most of) 10 meters down to about 1.25:1 or better - plenty good enough!  The response of this filter is shown below:

Figure 4:
Insertion loss and VSWR plot of the 15 MHz high-pass filter as plotted by a VNA.
As can be seen, attenuation at 40, 30 and 20 meters is well over 45dB with less than 0.5dB
at 15, 12 and 10 meters:  The VSWR is also acceptably low on these band as well.
Click on the image for a larger version.

Not shown in Figure 3, I later used RTV (silicone) adhesive to stabilize the coils and add support - after tuning, of course:  This reduces the probability of the coils being detuned by the filter being jarred or dropped.  RTV is fairly low loss (at least at HF) and far superior to "hot melt glue" in this case (it's lighter - and it won't melt!) and unlike hot glue or cyanoacrylate (e.g. "Super") glue, it can withstand mechanical shock without breaking loose - even when cold.

This filter should easily handle 100 watts - and the low loss is largely due to the use of silver-mica capacitors:  After all, 500 volt silver mica capacitors - such as those used here - may be found in wide-range antenna tuners made by LDG and the like where they would be exposed to more stress than in the filter.  If you are wondering about the use of the small, disc-ceramic capacitors, they are used in "low stress" parts of the circuit - to "trim" the capacitance to the needed value (e.g. a NP0 ceramic in parallel with a 120pF silver mica to get about 130pF) or used to "tune" the filter as in the case of the 30pF capacitor on the output.  While it might seem risky to use these tiny ceramic capacitors at 100 watts, a quick look at almost any Japanese-made amateur HF transceiver - particularly those made up until fairly recently - you'll find them sprinkled with these capacitors in the low-pass filters and even for matching in the final amplifiers - both at HF and VHF/UHF - for matching:  If it works for them, I'll not worry about using them here in the right places.

Don't forget to change the filter when you change bands!

One hazard with outboard filters of any type:  Be sure that the filter is removed if you attempt to transmit on a frequency for which it is not designed!  After nearly every Field Day I hear/read reports where someone - say, originally on 20 meters - then tries to QSY to another band with the 20 meter filter still inline using the radio's built in antenna tuner or an outboard tuner:  The result is is often that the filter is damaged!  ^{Footnote 7} 

Final comments

As can be seen from the response plot of Figure 4 this filter will attenuate signals on the bands 20 meters and below by more than 45dB and this should be enough to quash to inaudibility any low-level noise produced by the transceiver at these lower frequencies that might degrade reception on these bands.  Similarly, energy from transmissions on 20 meters and lower from other stations will be at a much lower level prior to reaching the front end of the radio using this filter, further reducing the probability that they could overload/cause noise.

One thing that has not been discussed thus far is the fact that harmonically-related frequencies (e.g. a transmitter on 7.05 MHz would have harmonics at 14.10 and 21.15 MHz) are likely to be audible on other receivers, despite heroic attempts to fully-filter them.  The reason for this is that these harmonically-related signals will be fairly strong compared to the noise floor of the amateur bands and, unlike the low-level noise discussed earlier, would have their energy concentrated into a small bandwidth.  

Such signals are also likely to be radiated not only from the antenna ports, but from other cables connected to the radios themselves - namely the power cables, audio/microphone connections, data and PTT lines which means that a filter on the output won't suppress those other leakage sources.  Other than wide-spaced separation (e.g. not placing radios in the same location and moving them as far apart as possible) there's no way to completely prevent harmonically-related QRM other than to coordinate efforts and simply avoid operations that could result in harmonically-related interference.

As it is not yet Field Day, I don't know if this filter will "fix" the problem that my friend was reporting, but  should help, and it was quick, cheap and easy to throw together. ^{Footnote 8}

* * * * *

After Field Day follow-up

A few days after 2025 Field Day I spoke again with the friend for whom I constructed the filter described above. While 10 meters was mostly dead, 15 meters was reasonably productive at times and operation on that band was successfully carried out - mostly using digital modes.

There was time to do a bit of A/B testing and it was noted that without the high-pass filter, the 15 meter transmitter did produce a very audible "hiss" on lower bands - most notably 40 meters - when it keyed up, but this was totally absent with the filter in place.  Additionally, a slight amount of QRM from the lower-band stations (on 80, 40 and 20 meters) transmitting was noted in 15 meter reception without the filter - likely due to strong signals impinging on the radio's internal switching diodes - but this was also absent with the high-pass filter installed.

It's also worth noting that several other tactics were employed to minimize the possibility of QRM (interference) between stations, including:

• Separating antennas as much as practical.  Depending on the site - not to mention the "1000 foot" rule - you can do only so much, but it helps to carefully consider the layout of the site to maximize distance between antennas.

• Using band pass filters.  These go a long way toward preventing interference to and from radios on other bands. 

• Use of mains filters.  L/C filtering (e.g. filters using inductors/capacitors) were installed on the long extension cords that fed the individual stations to prevent RF from being conducted on the electrical power leads and for this, "Isobar" plug strips were used.  Alternatively, putting a half-dozen or so turns of the extension cord through an FT-240 or "Monster" toroid (using 31 or 43 material for either one) would work as well - but it's recommended that the toroid be protected from damage by putting it in a box (e.g. a plastic electrical box with notches to allow the cord to pass).

• The use of "balanced" 4:1 baluns.  As it turns out, most "4:1" baluns are NOT very balanced - and as such they will put significant RF current onto the radio and feedline, not only reducing antenna efficacy but also potentially improving susceptibility of RFI (Radio Frequency Interference) both TO the radio and devices connected to it (e.g. computers, other radios) but also allow RF interference (from generators, switching supplies, lighting, computers, etc.) to find their way onto the feedline.  This "un-balanced" nature of most baluns can be proven by noting signal strength of a consistent off-air signal and disconnecting only one side of the feed to the balun:  With a truly balanced balun you should see signals reduce by 20dB (3-5 S-units, depending on the radio) or more, but with most baluns you will see only a small (6-10 dB - 1-2 S-units) at most.  Of all of the commercial offerings, one of the very few truly "balanced" baluns is the Balun Designs "Hybrid" balun (Model 4116 - link) - but most 4:1 baluns can be made to be balanced by immediately preceding it with a common-mode current choke (e,g, 8-14 turns of coax on an FT240-31 or FT-240-43 toroid).
  

 * * * * *

This page stolen from ka7oei.com

[END]

Footnotes:

1. Unlike most radios, Flex radios do include filtering to prevent low-level noise from being emitted on bands lower than the one on which it's being operated:  Specific models of other manufacturers may also include this - although most do not.
   
2. Direct-sampling receivers such as that of the IC-7300 have "different" problems in the presence of very strong signals compared to more conventional superheterodyne receivers:  Any signal that hits the analog-to-digital converter can cause overload, no matter the frequency.  While a conventional receiver can have a very "strong" mixer and some "roofing" filters in its IF (Intermediate Frequency) stages, this is not possible on a direct-sampling receiver.  Instead, it must rely on a rather large number of individual, overlapping band-pass filters to cover its intended frequency range and the ultimate attenuation of these filters may not be "strong" enough to prevent a nearby transmitter on another band from adding to the already-strong melee of signals on the crowded bands during Field Day and causing overload - or, at least, significant de-sensing (e.g. reduction in sensitivity).  This property is also what almost certainly makes them very poor candidates for being able to tolerate another local transmitter on the same band (e.g. a 20 phone and a 20 CW/digital station at the same Field Day site).  There are strategies that can improve the probability of two stations co-habitating on the same band - mostly having to do with picking the "right" radios (e.g. Elecraft K3S or the K4HD are known to work in this environment as are a few others) - as can the use of parallel-pointed Yagi antennas (see the next section, below) - or very "sharp" band-pass and notch filters can be constructed as described in two articles on this blog, namely:  A 100 watt "Helical" resonator bandpass/notch filters to increase isolation of 20 meter stations during Field Day (link) and Revisiting the 20 meter "helical resonator" band-pass/notch filters (link).
   
3. Being able to point beams parallel to each other is at least partly a matter of geography.  A station on the east coast is likely pointing their antennas west while the situation would be reversed on the west cost:  A station in the middle of the country - with signals coming from potentially all directions - would be less-likely to be able to use this tactic, at least not without a degree of coordination among the individual transmitters/stations.
4. A number of different manufacturers make band-specific filters for HF.  Depending on the design, these can offer modest (>=30dB) adjacent-band suppression - which is usually enough to solve most interference problems - or much higher degrees of filtering, even more than 50dB.  In addition, individual-band "Notch" filters are available from some suppliers that reject a specific band of frequencies which can be used several ways - on a transmitter to suppress any low-level noise that it might be generated on a specific band, or on another station to reduce the levels from a transmitter on that other band to prevent overload - and it can also be used to further-improve performance of a band-pass filter and increase attenuation on that specific band.  One of the companies that supplies such filters is Morgan Manufacturing (link):  Full disclosure - I know the person that runs this company and am quite familiar with the products.  Other manufacturers also make similar, excellent products as well.
   
5. This "hiss" can usually be detected without any sort of special equipment.  To do this, one would set up two transceivers in a relatively RF-quiet location (perhaps NOT a suburban home), each on its own antenna spaced within a few hundred feet/meters of each other.  On the radio doing the transmitting turn down the microphone gain all of the way after verifying that the RF power output would otherwise yield 100 watts peak when talking.  On the receiver, tune in the next band lower than the transmitter and note the noise floor with and without the transmitter keyed up.  In many cases, a "hiss" that can mask weak signals can be observed - particularly if using a resonant antenna on the transmitter without an antenna tuner.  If you couple carefully into the transmitter (using attenuators or directional couplers) this noise floor can be measured directly with a spectrum analyzer - even the $50-ish "TinySA" is up to the task!  It also goes without saying that a transmitter that IS outputting full power will also be prone to producing such hiss as well - not only above and below its actual transmit frequency, but on the "lower" bands as well.
   
6. Testing to determine the efficacy of the built-in tuner as a band-pass filters was done using a Kenwood TS-450SAT, a radio from the 1990s.  When the tuner was switched in and "tuned" - even if the load was already matched - it functioned as a low-Q band-pass filter that reduced the broadband noise and adjacent band signals by at least 8dB - and typically 20dB or so.  Whether or not this strategy is likely to work on specific radios (e.g. some may switch out the tuner if there is already a good match) would require testing as described above.  (These measurements were discussed in a very early entry of this blog - link).
   
7. The most likely components to be damaged when trying to "force feed" RF on the "wrong" band are the capacitors, followed by toroidal inductors being somewhat less-likely - and this will often happen when transmitting is attempted at full power (100 or more watts) rather than at the low power level used for tuning.  Usually, the operator realizes the mistake after the tuner fails to find a match, or it does find a match but signals are weak or absent.  For this reason, if you are using a filter with a radio and an external tuner it's strongly recommended that you place the filter between the radio and the tuner:  This will prevent damage to the filter as the radio will protect itself if it's used on the wrong band, presumably alerting the operator to the problem!
8. It took far longer to put together this article than it did to design, gather parts, assemble and tune the filter!

 

The construction of a hybrid ring combiner: Using the same duplexer, feedline and antenna for two repeaters/links/transmitters

Two transmitters, one antenna

There are occasions when it is desirable to combine two transmitters - and receivers - onto a single antenna - perhaps at a busy repeater site where more than one link or repeater is required.

The most obvious answer to this would be to have a separate antenna - and duplexer - for each of the links - but there are potential problems with this:

• Antennas are expensive
• Feedline is expensive
  
• Duplexers are even more expensive
• If you are on a shared site, you may have difficulty installing - or even getting permission to install - another antenna - and if you are renting tower space, this "other" antenna will be an ongoing expense.
  

If you have the luxury of planning ahead and you have a bit if extra link margin ^{Footnote 1} (e.g. you have "excess" signal and can afford a bit of loss), by placing two repeaters and/or link radios on adjacent - or nearly adjacent - frequencies, you can probably use a single antenna, feedline and duplexer - potentially saving money and hassle in the long run.  Let's take as an example as a hypothetical pair of full-duplex links at a single site in the U.S. on the 70cm amateur band:

• Transmit frequencies:  421.000 and 421.100 MHz
• Receiver frequencies:   434.500 and 434.600 MHz

As these pairs of frequencies are only 100 kHz apart from each other, they will handily fall into each others' notches in their duplexers meaning that neither transmitter is likely to bother either receiver.  What this means is that you could use a single duplexer along with a bit of extra gear (more on that later) to put both radios on the same antenna.  This same technique could also be applied if, for some reason, you had two repeaters at the same site (say an analog and a digital) as will be mentioned a bit later.

Note:  While this article describes usage on the 70cm band, there's no reason why it could not be applied to other bands - higher and lower - as well.

First, let's briefly cover using a single antenna to feed more than one receiver.

Receive:  Splitting the receive signal path

Receiving is pretty easy:  Just use a splitter.  If you are using a 2-way splitter this will result in 3.0-3.5 dB loss on receive, and there is - in most cases - likely to be enough link margin so that this won't be a problem, but if not, placing a preamplifier of modest gain (say, 10dB) in front of the splitter will overcome these losses:  Avoid the temptation to use a higher-gain amplifier than this to minimize the probability that it might make the problem even worse due to overloading (desensing) and producing intermodulation products that will cause other interference issues.

In a pinch, a 2-way "Cable TV" type of splitter - with appropriate adapters - will work fine for 2 meter and 70cm even though they are designed for 75 ohm system:  They will have reasonably low loss (less than 4dB for a good-quality unit) and a port-to-port isolation of 20dB or better.  For more information see the article on this blog:  Using TV (F-connector) 75 ohm splitters and taps in 50 ohm systems on the amateur HF, VHF and UHF bands (link).

Methods of combining closely-spaced verses more-distantly spaced transmitters   When combining transmitters, there are two general situations that arise:  Those that are very closely-spaced in terms of frequency, and those that are spaced farther away from each other. For two transmitters whose frequencies are separated by a reasonable distance (say, 5-10% of the frequency) it is usually practical to use a combination of notch cavities and phasing lines to separate them from each other - not unlike the way a duplexer allows both transmit and receive at the same time on a given antenna:  There's no reason why one couldn't have two transmitters on a duplexer.  If done carefully, such combination can incur additional losses of only a couple of dB. The problem changes if the two transmitters are quite close in frequency:  Eventually, the separation is so small that it is not possible to use resonant cavities to separate them from each other:  For 2 meters, this spacing is about 400kHz or narrower while on 70cm a spacing of less than 1.5-2 MHz starts to become more difficult.  As the spacing gets narrower and narrower, losses go up and in the case of combining just two transmitters, if the losses exceed about 3 dB, it may be better to simply use a hybrid combiner like those described here rather than potentially large, expensive notch cavities.  If there are more than just two transmitters to be combined, everything gets even more complicated and careful system design and frequency planning are of paramount importance to minimize losses.

You can also find on the surplus market (e.g. EvilBay) splitters - typically made by Mini-Circuits - that are native 50 ohm.  It's worth noting that Mini-Circuits tends to rate their products very conservatively and even a splitter/combiner rated for "only" 200 or 250 MHz at the top end will work just fine at 450 MHz.  One device frequently found is the Mini-Circuits ZFSC-2-1(+) which is rated for 5-500 MHz and is suitable for all repeater bands 10 meters through 70cm.

Transmit:  Combining multiple transmitters onto a single feedline

This is a bit trickier.  

If your frequencies are very near each other (within 1-2 MHz at 70cm, closer than 500 kHz at 2 meters) it's not likely to be practical to use complicated band-pass/notch schemes to isolate the two transmitters with less than about 3 dB of loss.  Again, we are presuming that your link budget will tolerate an additional 3dB of loss - and in most cases in the "real" world, this is likely to be true.

The "easiest" way to combine two transmitters - whether the frequencies are close to each other or not - is with a hybrid combiner.  If the frequencies are quite close together - as in our example above - then the same duplexer, feedline and antenna can be used.

Note:  On 2 meters, frequencies closer than 100 kHz can likely be combined and mutually filtered with a single duplexer/band-pass cavity while on 70cm you may be able to get away with up to 500 kHz of separation - and both may require a bit of careful tuning of the duplexers/band-pass cavities and - at the wider spacings noted above, the possible need to tolerate a bit of extra loss or reduction of isolation.  Clearly, the closer the better!

Wilkinson Power Combiner

There are two common types of combiners that would be suitable for this, both of which having the property of being useful only "near-ish" their design frequency - say, less than +/- 10% or so.  The simplest of these is the Wilkinson type, depicted in Figure 1.  This circuit is frequently used in power amplifiers when there are multiple gain stages in parallel that are phased identically, being fed from the same source and combined to get more output.^{See Footnote 6}  However, if you are combining different transmitters at different frequencies you will theoretically lose half the power - but there's really no way around this.  Again, it's quite likely that a loss of half of the transmit power (3 dB) at the antenna will have little actual effect on the coverage of the transmitter/link in a real-world situation.

Figure 1:
  Wilkinson combiner diagram (from Wikipedia)
In a 50 ohm system, this uses 72 ohm line and a
100 ohm resistor between P2 and P3.

How it works

Ports P2 and P3 represent the connections to the two transmitters and P1 is the combined output.  As can be seen, there is a total of 1/2λ (half  wavelength) between P2 and P3:  Any signal from P2 will arrive at P3 180 degrees out of phase and vice-versa, the ultimate result being that the signal from P3 is cancelled out at P2 and the signal from P2 is cancelled out at P3, effectively isolating the two transmitters from each other - which is important, as we'll later see.

Note that there a resistor between P2 and P3 which has a resistance of twice the system impedance - or in the case of a 50 ohm system, it should be 100 ohms.  Additionally, half of the TOTAL transmit power will be dissipated by this resistor if the signal sources are dissimilar (e.g. NOT being used to combine two parallel stages in an RF amplifier).  Finding a 100 ohm resistor suitable for 70cm that is also capable of handling 10s of watts of RF with a good return loss is tricky - and then placing it across two terminals that are "hot" with RF - makes the Wilkinson less attractive in this application.

Note that this device requires transmission line that is √2 * Zo - or in the case of a 50 ohm system, it's 50 * 1.414 = 70.7 (e.g. 71 ohms).  For our purposes, "75 ohm" cable would be fine if we were to construct one.

Note that the Wilkinson may also be used to divide (split) power equally to two loads should that be required.  This is most commonly done in a power amplifier where two, separate amplifier stages need to be fed with the same signal after which another combiner would be used to sum the outputs again. 

Hybrid ring combiner

Another type combiner that will fit the bill is the so-called "Hybrid Ring" (a.k.a. "Rat Race") combiner, depicted in Figure 2.

Figure 2: 
Hybrid Ring Combiner (from Wikipedia)
a.k.a. "Rat Race" combiner.  Like the Wilkinson,
this uses 72 ohm feedline in a 50 ohm system, but
the terminator (usually at P4) is 50 ohms.

Referring to Figure 2, our two transmitters are connected to ports P1 and P3 while output P2 contains the sum of the P1 and P3 signals and P4 contains the difference of the P1 and P3 signals.  All ports must be sourced/terminated at 50 ohms and we would typically connect 50 ohm transmitters at P1 and P3 and 50 ohm loads (antennas or dummy loads) at P2 and P4 as appropriate.  

If we look carefully, we can see that between P1 and P3 (via P2) there is 1/2λ of feedline between the two if we go clockwise around the circle - but if we go the other way around the circle we have 1/4λ plus 3/4λ - or a total of 1λ - which which means that there is 1/2λ difference in feedline length between them - thus a 180 degree phase difference - any RF being input to P3 are cancelled out at P1 and vice-versa.  With P2 being halfway between P1 and P3 - and going the other way we have 1.5λ of feedline between the same points and no cancellation we get the sum of the two transmitters.  Like the Wilkinson, you could use the Hybrid ring to split power between two loads:  To do this the transmitter (source) would be applied to P2 and the two loads would be connected at P1 and P3.

If we were combining two in-phase RF amplifiers from the same transmitter - as we might if we had two identically-phased amplifiers that we wanted to combine for more power, we would get zero power at P4 and the combination of the two amplifiers would appear at P2:  A dummy load - connected at P4 - would see little or no power if the two two amplifiers were operating with equal power and phase.

In our case, we are combining two different transmitters - on different frequencies - and this means that half of our power will end up each on P2 and P4, so we would typically connect our dummy load at P4 and our output to our antenna (via the duplexer) at P2.

Figure 3:
Andrews FSJ1-75 75 ohm Heliax - unfortnately
no longer being made, but other types of 75 Ohm
cable may be used as well as described in the text.
Click on the image for a larger version.

Like the Wilkinson power combiner, the ring combiner requires transmission line that is √2 * Zo (e.g. 71 ohms) - and again, 75 ohm cable is fine for most purposes in a 50 ohm system.  A distinct advantage of the ring combiner is that unlike the Wilkinson, the dummy load in the ring combiner is ground-referenced rather than having a resistor across two "hot" RF terminals - simplifying design.  Additionally, because the load that we would connect to P4 is the same as our system impedance - 50 ohms - finding a suitable device on the new or surplus market is quite easy.

Either the hybrid-ring or the Wilkinson combiners should yield well over 25dB of isolation between ports when properly constructed over a wide frequency range (8-10% or more) - and values of over 50dB are attainable on the workbench with known-good loads and sources across narrower frequency ranges.  Having good isolation between ports is vital when combining transmitters:  Not only does good isolation imply lower loss  (e.g. closer to the theoretical 3dB) but in the case of two different transmitters, minimizing the amount of energy that transmitter "A" gets from transmitter "B" reduces "wasted" power due to this cross-coupling, but more importantly it reduces the likelihood of the generation of IMD (Intermodulation Distortion) products that could cause receive performance degradation and interference - more on ways to prevent this will be discussed later.

Again, the reader is reminded that both the Wilkinson and Hybrid Ring combiners - unlike their counterparts that use transformers - are not inherently broadband:  They are typically used within less than +/-10% or so of their design frequency for best performance as this is the limit of efficacy of the quarter-wave phasing sections.  Later in this article we'll analyze the constructed hybrid ring combiner and demonstrate its frequency range.

Choice of transmission line

As noted, the impedance of 75 ohm feedline is "good enough" for these two types of combiners on a 50 ohm system in most cases and this sort of coaxial cable is readily available:  RG-59 and RG-11 types are suitable - as are some RG-6 cables such as Belden 1694A that have tinned-copper shields and solid copper center conductors to which we can easily solder - but note that most RG-6 cables use aluminum shields and copper-covered steel center conductors:  In theory these would work, but making a good connection to the shield is a complication.

NOTE:  There are PTFE (Teflon) 75 ohm coaxial cables available - see Footnote 2 at the bottom of this article for more information.

Figure 4: 
A hybrid-ring combiner, constructed as a "ring" using
FSJ1-75 Heliax and "N" type connectors on pigtails with bits
of brass tubing at the joints.
Click on the image for a larger version.

While I had all three of the typical 75 ohm cable types available to me, I found - while rummaging around - some Andrews FSJ1-75 1/4" Heliax ^tm coax.  Unfortunately, this cable hasn't been made since 2011, but it may still be found on the surplus market occasionally - and the data sheet for it is still online (link).  This cable is lower loss than any of the other options mentioned - but this is probably not important as such short lengths of it are used:  Reactance losses due to the construction itself are likely to be greater than the cable losses, anyway - and I would have gotten very similar results with the other choices.  The use of the smaller cable also implies a smaller bending radius - which will be important as we'll soon see.

In theory, one could construct a hybrid ring like this using RG-11-type coaxial cable, N-type connectors and known-good ^{See Footnote 3} N-type "tee" connectors - but you will have to carefully calculate the added lengths of the connectors and adapters when doing so.

Form factor

There are several ways that this ring combiner could be built.  The most obvious is in the form of an actual ring as depicted in Figure 4 - a combiner that I built over 25 years ago, also using the same Andrews FSJ1-75.  This uses short pieces of brass tubing to connect the segments together and holes in the sides of the tubing allow solder connections to be made between the segments and to the short "pigtails" with "N" connectors on them:  It measures better than 35 dB TX port isolation at its intended frequency.   This particular combiner had been used to combine a two UHF repeaters - an FM voice repeater and a 9600 baud packet repeater, on frequencies just 25 kHz apart - onto the same duplexer/antenna for several years with excellent results using the methods described below. 

Figure 5: 
For the ring combiner on this page, it needed to be a bit more
compact and rugged, housed in a Hamnmond 1590D box.
Click on the image for a larger version.

For the combiner that I recently built there was the need for something more "compact":  The "ring" described above (in Figure 4) is about 10" (25cm) in diameter and the connectors are awkwardly spaced, on flexible leads - and the entire thing is a bit fragile.

 

For 70cm, the feedline lengths are short enough that the sections can be put into a large-ish die-cast box - particularly if the cable used has a fairly tight allowed bending radius, which would rule out RG-11.

 

In this case I had on hand a Hammond 1590D which is 7.4" (18.8cm) x 4.7" (11.95cm) x 2.2" (5.6cm).  I also happened (as one does) a number of chassis mount "N" connectors with short lengths of UT-141 (hardline) already connected to them which would allow (literally!) some flexibility as to how the internal phasing sections of feedline could be oriented and connected.  It's worth nothing that if RG-179 were used (see Footnote #2) a smaller-still metal box could be used - even for a 2 meter combiner!

 

Running the numbers

First, calculate the wavelength at the desired center frequency.  In our case, we need 421.0 MHz:

300 / F (MHz) = Wavelength

300 / 421.0 = 0.713 Meters - The wavelength at 421.0 MHz.

Figure 6:
Preparing and measuring the cable pieces.  For the ring
combiner there is one 3/4λ and three 1/4λ pieces.  They are
intentionally cut slightly long to allow stripping and soldering.
Click on the image for a larger version.

Now, we need to find the velocity factor of our particular cable, a number which indicates the tendency for electricity to move slower than light when it is conveyed through conductors in the presence of a dielectric.  For best results, find the manufacturer of the specific cable that you are using as this varies - particularly between solid and foam dielectric cables and the precise type of dielectric.  For our FSJ1-75, the stated velocity factor is 0.78 (78%).  Knowing this, we can calculate the length of one electrical wavelength of our cable:

Vf * Length = Electrical length  So we plug in the numbers:

0.713 * 0.78 = 0.556 Meters

Figure 7: 
The stripped/prepped end of a cable segment as described in
the text.  The calculated lengths are measured between the
ends of the dielectric, where the center conductor protrudes.
Click on the image for a larger version.

As we can see from the diagram of Figure 2, we need four pieces of cable:  One that is 3/4λ (0.75λ) and three more that are 1/4λ (0.25λ), so we multiply the above to get those:

0.556 * 0.75 = 0.417 Meters for 3/4λ section  - approx. 16-3/8"

0.556 * 0.25 = 0.139 Meters for the 1/4λ sections  - approx. 5-1/2"

Prepping the cable

As we need a bit of extra cable to expose the center conductor to which we solder we need to make the sections slightly longer - about 1/2" (13mm) at each end - or about 1" (25mm) longer overall for each piece.  After cutting the pieces of cable, put a mark at the center of each and measure about 1/2" (1cm) less than half the length that we calculated above to the end strip back the outer jacket:  This will leave a section of bare shield to which we can later solder.  Now remove the shield at a point about 1/4" (0.5cm) less than half the length that we calculated above and remove the dielectric, but leaving a couple of mm (about 1/16") out from the end of the shield:  And example may be seen in Figure 7.

Note that when we calculate the length of the cable - 0.139 meters for a 1/4λ section in our example - we are measuring at the points where the center conductor protrudes from dielectric.  As can be seen in Figure 8, the center conductors are then bent over level with the top of the dielectric to be soldered to to the other cable segments and the distances are measured from the points where they are bent over.

Figure 8:
The junction of the three cables, with the center conductors
bent over and soldered - and with the shields firmly soldered
together as well, using 26 AWG bare wire during assembly.
Click on the image for a larger version.

At this point it would be a good idea to consult the specifications of the cable that you are using and determine the minimum bending radius.  For FSJ1-75, this is specified as being 1.25" (31.75mm) and for Belden 1694A this is about 2.75" (70mm):  Try to bend it less than this if you can:  Cut a piece of cardboard or paper with a circle of this radius as a comparison.

To get an idea as to how everything should be routed inside the box, the connectors+cables were first installed and tightened and the four pieces were laid out and moved about to determine what made sense.  As can be seen from Figures 8-11, each of the cables were bent at a gentle right angle so that the center conductors all came together at one point and the shields in parallel with each other - all without bending the cables at too-tight a radius.

Once the orientation of a cable end was determined, the center conductors were bent over - leaving a gap of about 1/16" (2mm) between it and the shield and the connections tacked together.  After this, some tinned 26 AWG wire was wound tightly around the shield and then soldered, making a both a very short-length electrical connection and providing mechanical rigidity as can be seen in Figure 8.

Figure 9: 
A better view of the cables being assembled in the box,
showing how the 26 AWG bare wire first used to tie the
springy cables together and then flooded with solder.
Click on the image for a larger version.

A very hot soldering iron is a must here as it allows connections to be made very quickly and prevent melting of the dielectric.

 

If you have it, use tin-lead solder (e.g. 60/40 or 63/37) as it melts at a lower temperature than "lead free" solder and is less likely to melt and damage the cable's dielectric. Additionally, a few drops of either "no-clean" or rosin flux will help make good, clean joints with minimal heat.

As it turns out, the FSJ1-75 is relatively forgiving in terms of heat if you use a hot iron and work quickly - but if you use something like 1694A or other flexible coaxial cable, you may want to tin the braid prior to assembly - starting with a few bits of "practice" coaxial cable to avoid melting pieces that you have already cut to length and prepared.

Figure 10:
The cable segments were originally fitted in the box and
tacked together as seen in Figure 9, but were carefully removed
so that the connections could be fully soldered on both sides.
Click on the image for a larger version.

Testing

A NanoVNA is a good tool to test the combiner - but you will also need TWO known-good 50 ohm terminators (dummy loads).  For this, you probably won't want to use anything but good-quality units - which are available surplus - and you will want to use an ohmmeter to verify that they are in the range of 50-52 ohms - and don't forget to take into account the resistance of your ohmmeter leads!

Before proceeding, be sure to do an "SOLT" (Short-Open-Load-Through) calibration of your VNA for the frequency range of interest.  In our case, it's 400-500 MHz. 

 As you will be measuring "through" loss, I suggest that for measuring this device that you use the "receive" (second) port of your NanoVNA instead of the termination when doing the SOLT calibration as it will likely not be quite as good as the load that came with the NanoVNA and would otherwise make VSWR/return loss measurements "appear" to be worse than they are:  If you are interested in single-port (S11) measurements only, use your SOLT load for calibration and either it or a known-good load when testing..

Referring again to Figure 2, note which of the four connectors correlate with P1, P2, P3 and P4 (mark them with a pen or label) and connect the dummy loads to P2 and P4.  Then, connect the "In" and "Out" leads of the NanoVNA (or similar) to P1 and P3 - it doesn't matter which way.

Figure 11: 
Initial testing of the combiner, showing the NanoVNA
connected to ports P1 and P3 and the dummy loads on ports
 P2 and P4.
Click on the image for a larger version.

If all goes well, the VSWR should be below 1.25:1 and the "through loss" (which is the isolation between the two transmitters to which P1 and P3 will be connected) should be well above 25 dB.  At some frequency - probably a bit above the intended design frequency you may see the through loss dramatically increase (a "dip" in amplitude) of port-to-port isolation between P1 and P3.  If you do not see significantly more than 25 dB of isolation between P1 and P3, re-check your connections:  If you see way under 20dB you have probably misidentified your connections and should check again - but if you think that you have properly identified everything, re-check your math from when you calculated the cable lengths - and don't forget to include the velocity factor.

 

As we'll see later, a device like this should yield in excess of 30dB transmit-transmit port isolation over a very wide frequency range - even without any "adjustments".  In Figure 11 you can see the initial testing of the ring combiner and in the background on the NanoVNA - on the blue trace - the "dip" showing the frequency of the best port-to-port isolation - which, before tuning, was near 435 MHz.

Having verified that the isolation between P1 and P3 is good, connect one of the NanoVNA leads to P2 and move the dummy load onto the to which the NanoVNA had been connected:  If the unit is working properly, the insertion loss will be about 3dB, which is exactly correct.

Figure 12: 
A "tuning strip" used to make slight adjustments.  The
"grounded" strip is moved closer to the exposed center
conductors to increase capacitance.
Click on the image for a larger version.

Making adjustments

While the "ring" depicted in Figure 4 is elegantly simple, it can't be adjusted:  Scrunching it into the box as shown in Figures 10 and 11 allow a bit of tweaking to optimize both isolation and matching by virtue of the exposed center conductors.

When I built the combiner pictured I noted that the best insertion loss between P1 and P3 appeared as a "dip" around 435 MHz with the lowest VSWR occurring around 440 MHz.  I observed that very lightly touching my finger at the junction of P4 caused this "dip" to shift down in frequency, indicating that a very small amount of capacitance might better things at our 421.0 MHz design frequency.

Rather than connect a small variable capacitor - which would need to be of very low capacitance values (possibly less than 1pF) I did something simpler:  I cut a small strip of brass sheet and soldered one end to the shield of the coax at the junction of the cables as depicted in Figure 12.  This strip was then bent around the exposed junction - but kept at 1/8-1/16" (2-3mm) away from the center conductors to avoid shorting:  This added a bit of capacitance at that point and shifted the "dip" down in frequency, and by adjusting this brass strip closer/farther away from the center conductors of the cables, I was able to "dial" it in at 421 MHz.  If you can't get the strip close enough to the center conductor to bring the frequency down, solder a smaller strip of metal to the center conductor to form a larger capacitor plate, allowing more capacitance with greater distance.

Figure 13:
Inside the combiner - tuning strips installed on P1 and P4
which are used to slightly tweak the performance at the
intended operating frequency.
Click on the image for a larger version.

Similarly, I noted that lightly touching the connection at P1 reduced the VSWR slightly (which had started at around 1.2:1) and a similar brass strip was installed there.  The two interacted slightly with each other and I was able to get both a lower VSWR (under 1.1:1) and very good isolation (more than 50dB) at 421 MHz.  Note that when connected to the "real world" (e.g. actual transmitters and an antenna) which will probably have some reflected power and will NOT likely have precisely 50 ohms of impedance, the isolation will likely be less than what we measure on the workbench.  The lack of a really low VSWR (e.g. higher return loss) anywhere across the frequency range can likely be attributed to the fact that we used nominally 75 ohm feedline for the construction of the combiner rather than "70.7 Ohm" feedline - which is difficult to find in the form of coaxial cable!

With a bit of finessing I was able to get the port-to-port isolation and VSWR shown in the plot below:

Figure 14:
Port-to-port isolation (blue line) and VSWR (green) with a Smith chart in the background
and a legend in the upper-right corner showing the readings at the marked frequencies on the plots.
There is well over 50dB of isolation between P1 and P3 at the intended frequency.
Click on the image for a larger version.

As we can see from Figure 14, the port-to-port isolation at our target frequency is quite good - in excess of 50 dB - while the VSWR is around 1.1:1 as noted in the upper-left corner.  It's worth noting that the port-to-port isolation is better than 30dB between 400 and 445 MHz - a span of about 10% - and it's better than 20 dB from somewhere below 400 MHz to 491 MHz - and no-where on this plot does the VSWR exceed 1.4:1.  What this means is that if your goal is 30dB isolation, the design is quite forgiving.  As can be seen from the Smith chart in the center, the matching is pretty well-behaved.

What about insertion loss?  This plot gives us the answer:

Figure 15: 
TX port to antenna insertion loss (blue) and VSWR (green) - again with a Smith chart in the
background and the legend in the upper-left corner.
This plot shows just 0.05dB above the theoretical at the 421 MHz design frequency and a
good VSWR as well.
Click on the image for a larger version.

This graph shows the measured insertion loss and at our target frequency of 421 MHz, it is around 3.05 dB - very close to the theoretical 3dB loss and getting very near the uncertainty of our measurement.  Over the same "30dB port-to-port isolation" frequency range that we measured above (400-445 MHz) we see that the insertion loss is lower than 3.2 dB and that the VSWR over this range never exceeds 1.2:1.  Even up at 491 MHz where we had only about 20dB port-to-port isolation the insertion loss is a bit over 4 dB with a VSWR of 1.4:1 - not great, but still usable in a pinch.

Putting it into practice

Figure 16:
A two-stage UHF isolator with loads - a Ma-COM 7R011,
owned by the author, designed to work at UHF ^{Footnote 5}.
Although the two loads are only rated for 12 watts, the
 unit itself can handle 125 watts if the load on the left
(nearest the "output") were sized accordingly.
Click on the image for a larger version.

Being able to bash two transmitters on "nearby" frequencies together is a good thing - but we do need more than just the combiner to do so effectively and "cleanly".  (Note:  If the transmitters' frequencies were spaced farther apart, a different scheme would be required - see Footnote #4). 

 

The entire point of having a ring combiner with good port-to-port isolation is to minimize losses - here, we limit them to about 3dB - but it also prevents the RF output power of one transmitter from getting into another where mixing can occur, producing low-level IMD (intermodulation distortion) products that could cause interference to your own gear and - more importantly - other users on site.  As the port-to-port isolation is likely to diminish with "imperfect" sources and loads, we need to do more to prevent these mixing products from occurring than just the combiner.

To minimize the probability of this occurring it is also a very good idea to install a ferrite isolator on the output of each transmitter, prior to the ring combiner as depicted in Figure 18.  These devices - one of which is pictured in Figure 16 - can be through of as a sort of "diode" for RF:  They will let transmit power go through them, toward the antenna, but any power that comes the other way - reflected due to VSWR or RF energy from another transmitter - will end up in its dummy load(s) - effectively preventing RF from getting back into either of the final amplifiers:  Figure 17 shows what a properly-functioning isolator does to block RF coming back from the "load" port - in this case it reduces that energy by around 60dB at the frequency to which it is adjusted.

Figure 17:
The reverse (isolation) plot of the 2-stage isolator of Figure
16.   At the frequency to which it is tuned (450 MHz)
it has over 60 dB of reverse isolation. In the other direction
(not shown in this plot) its loss is only about 1dB.
Note that +/-25 MHz from center, the isolation drops below
30dB which is why a pass cavity is suggested!
Click on the image for a larger version.

These isolators (which are "circulators" with included "dump" loads) generally come in two flavors:  Single stage and double stage - the latter having two devices in the same package like the one in Figure 16.  Typically yielding 20-30 dB of reverse isolation per stage, a double stage device typically has between 40 and 60 db reverse isolation as shown in Figure 17.  This plot also shows something else:  While the reverse isolation is very good at its tuned frequency, it decreases as you move away:  If you have other users on site - and you are not using a pass cavity between the isolator and antenna - these "off-frequency" signals will not be as well-attenuated by the isolator as at the design frequency and its efficacy will be reduced while the probability of IMD being generated in the transmitter's output will increase.

 

While isolators can be quite expensive when new, they are frequently found on the surplus market.  Most isolators are tunable, and this will need to be done to optimize performance on your operating frequency - but this can easily be accomplished with a NanoVNA and instructions as to how to do this may be found on the Repeater Builder web site (link) as well as on videos on YouTube.  Even the extra expense of isolators and a band-pass cavity may well be less expensive than and preferable to the installation of two separate antennas, feedline and duplexers - particularly if another antenna were to incur recurring costs such as maintenance and tower rent!

 

If your combiner had a rather "average" 30dB of port-to-port isolation and you were using a two-stage isolator on each of your transmitters this would mean the RF energy from one transmitter getting into the other would be down between 70 and 90dB - and this nearly guarantees that IMD products will be extremely low and likely undetectable.

To further reduce the probably of harmful IMD product escaping your system, a band-pass cavity should be inserted between the output of the combiner and the "transmit" port of the duplexer as depicted in Figure 18:  As noted in another article on this blog (See the article:  When Band-Pass/Band-Reject (Bp/Br) duplexers really aren't band-pass - link) almost all duplexers used in amateur repeater service will NOT offer much filtering once you move a few MHz away from their tuned frequency. 

Figure 18: 
A typical application using a ring combiner, along with isolators and a band-pass cavity in the TX leg.
The isolator on TX #1 - and the ring combiner - minimize the amount of energy that it "sees" from
TX #2 and vice-versa while the band-pass cavity limits off-frequency energy that can enter the system.
Click on the image for a larger version.

 

By including a band-pass cavity in the "transmit" leg of the duplexer - prior to the duplexer - energy that is away from the operating frequency will be attenuated significantly.  This is important as many isolators have a rather limited frequency range over which they are most effective as seen in the figure above.  Additionally, if there are very low-level IMD products produced by mixing within your two transmitters even with isolators and the port-to-port isolation of the combiner, these will be significantly quashed by the band-pass cavity, practically eliminating even the possibility of self-generated interference.

 

Figure 19:
The completed UHF ring combiner.
The labeling on the sides of the box
identifying the ports is not visible in this photo.
Click on the image for a larger version.

Conclusion:

With a bit of planning and foresight is is possible to combine multiple transmitters - or even full duplex link radios or repeaters - onto a single antenna.  The techniques described here are most useful if you are able to do frequency planning by placing the transmit/receive frequency pairs quite close to each other, permitting the use of a common duplexer and single band-pass cavity in the TX and another in the RX leg.  Even if the frequencies are separated a bit and a common duplexer is not possible, these techniques can still be adapted.

 

In most cases, the extra 3dB of loss on receive and transmit can be ignored as such a reduction in sensitivity radiated power is likely to be unnoticed - and it well may be worth the trade-off in terms of minimizing infrastructure and even cost.

 

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Footnotes:

1. Link margin refers to the amount of signal between a transmitter and receiver - and specifically, the degree to which it could degrade and still produce acceptable results.  Consider the following:  Let us suppose that your goal is to have a link with 20dB SINAD or better and your receiver was capable of producing a signal of this quality with -110dBm at its input terminals.  If the strength of your receive signal from the other end of the link was -90dBm, it could be reduced - via fading or other path degradation - by 20dB before it would be considered to be "faded".  If - in the process of putting two radio systems on the same antenna you were to reduce either the transmitter at the far end or the receiver at the near end by about 3dB, in our example we would still be left with about 17dB of "fade margin".
   
2. RG-179 and RG-302 - both being 75 ohm available with PTFE (Teflon) and similarly-resistant jacket - would be excellent choices in the construction of a ring combiner owing to their flexibility and heat resistance.  RG-179 is available from  both Mouser Electronics - link -and Digi-Key Electronics link at the time of writing and while rather expensive, it doesn't take a lot of cable to construct a ring combiner.  While the PTFE is preferred, it's also available with other types of dielectric/jacket which will work, but care would be required to avoid melting it during soldering.  The PTFE cables' velocity factors are typically around 69% while the polyethylene versions are around 79% - but always check the manufacturer's data sheet!
   
3. If you ever use "Tee" type adapters, be extremely careful what you get - particularly if you are using type "N".  Many "foreign-made" N-type connectors are very poorly built - both mechanically and in terms of RF - and some of them having been found to use springs to make contact (highly inductive - VERY BAD!) and/or compression-type connections that tend to oxidize.  If you can afford it, such "Tee" connectors from Pasternak (link) will likely be good - as are genuine old mil-spec devices from reputable manufacturers (Amphenol, Kings, etc.).  If you don't know their quality, be prepared to measure them carefully using a VNA - and better yet, buy an extra so that you can cut it apart and see for yourself if it resembles anything like a "constant impedance" device with solid, reliable construction.
4. Note that since in our example the two transmitters' frequencies are very close together (only a few hundred kHz at 70cm) there would be little point in putting a band-pass cavity between the output of the isolator and ring combiner, but if the transmitters were several MHz apart - a spacing at which the cavities would offer reasonable rejection - it might make sense to do so.  If one did have several MHz spacing, it's less likely that the band-pass cavity in Figure 18 would be appropriate and that one could get away with using a single duplexer, either. 
5. The two-stage UHF isolator in Figure 16 was bought by the author "as is" - with the pair of 12 watt loads - at a swap meet for about $20 - the low price being due to the fact that it had clearly been submerged in water for a while and was showing some corrosion.  It was very carefully being disassembled and thoroughly cleaned - which included washing the trimmer capacitors with denatured alcohol to remove any contaminants - and resoldering the internal connections as it was reassembled, it once again worked, more than meeting the manufacturer's specifications for both reverse isolation and forward insertion loss.
6. The Wilkinson divider/combiner is often seen in high-power amplifiers - both to split drive power to identical amplifier stages, and to combine them again after amplification - and in this case, the same signal in terms of frequency, phase and power is being combined/split.  In cases like this where the inputs and outputs are operating at identical power levels (e.g. combining two, signals or putting two identical signals together) the 100 ohm load across P2 and P3 in Figure 1 sees NO (or very little) voltage difference and dissipates little/no power.  If, however, the two signal paths become different (e.g. one of the two amplifiers fails) this resistor will then dissipate a significant percentage of the remaining signal power.  Anyone who has had to repair a power amplifier that uses such a combiner/splitter arrangement knows that in addition to needing to repair the amplifier stages - and make sure that their outputs are phased equally (there's often a capacitor to do this) that equalizing resistor will have been "smoked" (destroyed) almost instantly when one of the amplifier sections failed as they typically use a resistor that is rated for a fraction of the power output:  If everything is fine, the resistor survives - but if not...
   

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This page stolen from ka7oei.blogspot.com

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