An ESP8266-based Temperature, Humidity and Line Voltage monitor

Figure 1:
The completed Temperature/Humidity/Line Voltage web
server/telemetering device.  The remote temperature/humidity
sensor is the unit to the left.  The two AC-DC wall adapters used for
powering the unit and monitoring mains voltage are not visible.
Click on the image for a larger version.

As anyone who reads this blog probably knows, I have a bit to do with the operation and maintenance of the Northern Utah WebSDR - a remote receiver system that allows anyone with Internet access and a web browser to listen to the LF, MF, HF and some of the VHF bands as heard from a rural site in Northern Utah.  The equipment for this receiver system is located a small building in the middle of mosquito and deer-fly infested range land near brackish marshes - no-where that anyone in their right mind would like to be during most of the year.  With the normal weather in the summer and many clear days, this building gets hot at times:  It's been observed to exceed 130F (55C) on the hottest days inside - a temperature that causes the fans on the computers scream!

Even though electronic equipment is best kept at much lower temperatures, this isn't practical in this building as it would be prohibitively expensive to run the on-site air conditioner full time - but all we really need to do is to keep the building closer to the outside temperature and even though it may be uncomfortable for humans, it is enough to keep the electronics happy.  To that end, vents have recently been installed to allow convection to pull away most of the heat and the exterior will soon been painted with white "RV" paint to (hopefully) reduce the heating effects of direct sun.

It would make sense, then, that we had a way to remotely monitor the building's internal temperature as a means of monitoring the situation.  Additionally, temperature information can also be used to make minor adjustments to the frequencies of some of the receivers' local oscillators to help counter thermal drift.

Figure 2:
The "business end" of the small board that contains the
ESP8266 module - the device with the metal shield.
This board also includes a USB plug, a CH340-like
USB to serial converter that allows for programming
and debugging and a voltage regulator that allows direct
operation of this board from a 5 volt supply.
As can be seen here and in Figure 4, the ESP8266 board was,
itself, mounted to a larger prototyping board for construction
of the ancillary circuitry.
Click on the image for a larger version.

On site we do have an Ambient Weather ^(tm) station, but anyone who has used this (or similar) hardware knows that some vendors of this type of gear make it difficult to obtain your own data without jumping through hoops:  Although this data is visually available on the local display or even on a web site, it is a bit awkward to pull this data from their system and (at least with the newer versions of the hardware) one cannot get this data locally from the weather station itself.

Fortunately, the most-needed data - temperature inside the building - is easily measured using inexpensive sensors, so it made sense to throw together a device that could make these measurements and present them in an easy-to-use web interface.

The ESP8266 "Arduino" board:

As is often the case with projects like this, the Internet has the answer.  The ESP8266 is an inexpensive embedded computer module that has a reasonable amount of program memory and RAM and it also sports hardware such as a WiFi module, several digital  I/O pins and a 10 bit A/D converter.  What this means is that for less than U.S.$12 you can get two of these delivered to your doorstep that contain an already-mounted ESP8266 module on a carrier board with a USB port in a format that strongly resembles that of the ubiquitous Arduino development board.  More importantly, the Arduino IDE supports this board meaning that it is pretty easy to use this hardware in your own projects.

Because the '8266 board has been available for quite a while, there is a large library of software for it - including a small web server and code to interface with many types of devices, including the well-known (and relatively inexpensive) DHT-22 temperature and humidity sensor.

Comment:  The ESP8266 variant used here appears to be the "12E" version which has 32 Mbit (4 Mbytes) of Flash memory and "around 50k" of RAM.

The "DHT Humidity and Temperature web server":

It took only a few minutes to find online several implementations of a web server coupled with the DHT-22 sensor - and I chose what seemed to be a popular version on a web site by Rui Santos - to look at it yourself, go here:

randomnerdtutorials.com/esp8266-dht11dht22-temperature-and-humidity-web-server-with-arduino-ide/

Presented in good detail, it was only about 20 minutes from the start to tack a few flying leads to my $6 ESP8266 "Arduino" board to connect the DHT-22 sensor before I had a wireless web server on my workbench that was happily reading the temperature and humidity.

Of course, getting something working can be miles from a finished project and that was certainly the case here as the project was about to be subject to self-inflicted feature creep and code bloat as I'd already decided that I wanted it to do two other things as well:

• Monitor the AC line voltage.  The WebSDR receive site - being rural - suffers from very dirty AC mains power.  We have seen the nominal 120 volt mains exceed 140 volts for brief periods in addition to the frequent outages - and it would be nice to have a device that would allow us to record such excursions.

• Telemeter the gathered information via RF.  Because the ESP8266 is a small computer - and it has data that we want - and we are at a radio receive site - it would be a simple matter to have this unit tap out the information using Morse code on a low-power, unlicensed (part 15) transmitter that was capable of being received by the on-site receivers.

The final result is this, in schematic form:

Figure 3:
The schematic of the support circuitry of the ESP8266 unit described, including a pictorial representation of the processor board itself.
Click on the image for a larger version.

Circuit description:

The ESP8266 is treated as a single component - the support circuit being connected to the pins as noted with the '8266 itself being mounted on a larger board as can be seen in Figure 4, below.  It's worth noting that this is a 3.3 volt device which means that the "high" output voltage is around 3 volts:  If I'd needed a digital input, I would have had to make sure that the logic high input level was appropriately limited in voltage.

Power supply and monitoring:

There are two uneregulated AC-DC transformer "wall warts", both being a low-voltage transformer (9-12 volts AC) with full-wave rectification and capacitive filtering  - one to power the unit and the other to monitor the line voltage.  The separation of these two function is necessary for obvious reasons:  We'd want the unit to continue to function when the AC mains was out, but continue to run from the UPS which means that we can't monitor or own power supply!  Even if we could, the current consumption of the unit varies a bit and as a consequence, so does the unregulated voltage from the monitor supply.  The source of power for the unit itself could be anything that can provide 10-15 volts DC - regulated or not - but these AC->DC transformers were on-hand, plus being simple transformer-rectifier-filter units, they do not generate RF noise - unlike some switching-type devices - a factor important at a radio receive site.

The power from each of the AC-DC adapters enter via a screw terminal strip and immediately passes through a pair of bifilar-wound inductors - the purpose here being to provide RF isolation:  Because this device contains a computer and a low-power transmitter, we don't want any signals on this device from being radiated on the power leads.

The first AC-DC adapter is used to power the unit - a red LED indicating that voltage is present.  Following this is a "bog standard" 5 volt regulator using a 7805 to provide a lower voltage to feed to the "VIN" pin of the ESP8266 board and to run other circuitry on board.

The other wall wart has only a light load - most of the current being consumed by D4, an orange LED used to indicate that the mains voltage being monitored is present.  As you would expect, an unregulated AC-DC supply like this isn't a precision instrument when it comes to measuring line voltage as it is not any sort of RMS measuring device and with its built-in filter capacitor, it's also relatively slow to respond  - but it is "good enough" for the task at hand.

This voltage is divided down via R3 and variable resistor R4 for the 0-3.3 volt input range of the "A0" (analog input) pin on the ESP8266 module.  (Note:  It's reported that A/D range of the "raw" '8266 module is 0-1 volt - apparently this board includes a voltage divider to scale from 3.3 volts.)  Resistor R4 is 10 turn unit used for calibration of the line voltage.

Watchdog timer:

Figure 4:
The completed ESP8266-based temperature, humidity and line voltage
monitoring device.  The CW transmitter portion is in the upper-left corner
of the board with the 555-based watchdog timer below it.  In the lower-
right corner is the 7805 regulator with its heat sink with R4, the
calibration for the line voltage being seen just below the lower-left
corner of the ESP8266 board.  The gray wire at the top connects to the
small board containing the DHT-22 temperature/humidity sensor.
The entire unit is mounted via stand-offs into the lid of the plastic case
depicted in Figure 1.  Inside the lid I placed a sheet of self-adhesive
copper foil that is used as a ground plane to which the input filter
capacitors (C1, C3), the LEDs and the ground connections of the
board are soldered.
Click on the image for a larger version.

Because this device is unattended in a remote location I took the precaution of adding a simple hardware watchdog timer.  The software generates a pulse train (nominally a square wave) on pin "D2" which is then applied to transistor Q1:  Capacitive coupling, via C7, is used as a DC coupled signal would have made it possible that a watchdog reset condition could have been simulated if the pin were stuck "high".  The timer itself is the ubiquitous NE555 "programmed" via C8 and R7/R8 to have an approximately 45 second period.

The pulse train from pin D2 pulses Q1, keeping timing capacitor C8 discharged - but if the pulse train stops, pin 3 will go high after the timing period, briefly pulsing the "RST" (reset) pin of the EP8266 via capacitively-coupled Q2.  A 45 second period was chosen as it takes about 8 seconds for the ESP8266 to "boot up" enough for the software to generate the - and it also allows just enough time to upload the program.

During initial development one would probably not plug a 555 into its IC socket as spurious resets would likely be an annoyance as there may not be code to create the reset pulses, but with the size of the code for this project the reset period is long enough to allow uploading of the code before a reset occurs and the pulses resume.

Temperature/Humidity sensor:

The readily-available DHT-22 sensor is used, chosen over the slightly cheaper DHT-11 as the '22 offers a wider temperature and humidity measurement range - although the software can be configured to work with either one. To avoid erroneous temperature or humidity measurements from the unit's heat generation, this sensor is mounted on its own board as depicted in Figure 3.  On this small board is not only a power supply bypass capacitor (C20) but also a pull-up resistor R19.

The "sensor module" - visible in Figure 1 - was placed inside a small piece of ABS tubing (gray non-metallic electrical conduit) for protection with small pieces of nylon window screen glued to each end to keep out insects, but allow air flow to permit accurate measurements.

CW transmitter:

Because it is a computer - and there was plenty of code space - I decided to add Morse Code generation to provide telemetry that could be picked up by the HF receivers on site.  Stealing my own Morse-generating C code from a 20+ year old PIC project, I made minor modifications to it, using a hardware-derived timer in the main loop to provide a sending clock.  The Morse generating code toggles D3, setting it high to "key" the transmitter.

The signal from pin D3 goes to Q3 which is wired via current-limiting resistor R13 to Q4, a PNP transistor to provide "high side" keying of the unregulated V+ supply.  This voltage is then passed through resistor R14 which provides both a bit of current limiting and, with C12, some R/C filtering to slow the rise/fall of the voltage:  Without it the RF would have been keyed very "hard" causing objectionable "key clicks" on the rise and fall of the RF waveform.  This voltage is used to key both the buffer and the output amplifiers, described below.

The signal source is a 28.57 MHz crystal "can" oscillator module that I found in my junk box.  While I could, in theory, have done CW keying by turning this oscillator on and off, these oscillators aren't designed to be particularly stable and doing so would have caused the oscillator to "chirp" - that is, the short-term frequency drift that occurred when power was applied would have caused an objectionable shift in the received audio tone during keying.

Instead, the oscillator was powered continuously with its output fed to Q5, an emitter-follower buffer:  R15 "decouples" the oscillator from Q5 somewhat and without it, the RF current into the base of Q5 would increase when its collector voltage was switched off causing the oscillator to heat internally, resulting in a frequency shift.  The output of the buffer circuit is then passed via resistive and capacitive coupling to Q6 which is used as the final RF amplifier.  L1 is used to decouple its collector from the power supply while C16 removes the DC voltage from the RF output.  The remaining components - C17-C19 and L2/L3 comprise a low-pass filter resulting in harmonics that are at least 40dB below the fundamental.

This circuit was originally built without Q5, the buffer amplifier, but I had two issues that could only be resolved by its addition:

• Backwave.  Because the oscillator runs continuously, there will inevitably be a bit of leakage - and in CW where it is the very presence and absence of the signal that is used to convey the information, having a rather strong signal when there is supposed to be silence made it difficult to "copy" the code.  When Q6 was turned off, there was enough leakage between its base and collector to offer only about 15 dB of attenuation when the transmitter was "un keyed".

• Oscillator stability.  As noted above, R15 was used on Q5 to limit the current out of the oscillator to prevent frequency drift as buffer transistor Q5 was keyed.  When I'd tried to drive the output (Q6) directly - without a buffer - I had the same problem:  If I coupled enough energy to drive the transistor, the frequency would vary with the CW keying - and the backwave would get worse - but if I increased the resistor enough to reduce the problem, the transistor would be properly driven - which also increased the backwave as the transistor's output would fall in comparison to the signal leakage.

With the circuit built as shown the backwave is at least 40dB below the keyed output  - which is more than adequate for the task.

The code:

As noted above, the basis of the project was that published by Rui Santos - and in the spirit of open source, the code was modified:

• The original code included some small graphics served on the web page - but in line with the KISS principle, this was stripped out in favor of the simplest text, using only standard HTML formatting.

• Additional code was added to read the AC mains voltage via pin "A0".  In the main "loop" routine, this input is read 100 times a second and then averaged, with a new reading made available every second.  This average removes most of the noise on the pin - some of which is internal to the '8266 itself, but the majority of which is due to a small amount of AC ripple on the voltage monitoring line.

• Additional code was added to record the minimum and maximum of all of the monitor parameters - that is, temperature, humidity and line voltage.

• The default of the code was to read the temperature in Celsius, but being in the U.S. I added code to give the readings in Fahrenheit as well. 

• The web server code was modified to display all of the available data - the temperature in Fahrenheit and Celsius, the humidity and line voltage - and their minimum and maximum values.

• Addition modification was made to the web server code to allow each of the data points to be read in simple text format to simplify parsing for remote monitoring and logging of this data.  The nature of the the web server actually made it very easy!

• Yet another modification was made to reset the minimum and maximum readings and to provide information as to how many seconds it had been since a reset had occurred.  The temperature/humidity min/max reset is separate from the line voltage min/max.

• The code also keeps track of mains voltage that falls below a threshold (an outage) or exceeds a threshold (a "surge") - both of which are extremely common at the remote receive site. 

• I added my own Morse generation code, ported from some PIC-based "C" code that I wrote about 25 years ago and interfaced it with the main timing loop.

Source Code:

If you are interested in the source code (sketch) you may find it at THIS LINK.

A few minor changes/improvements - noted in the source code's header - have been made since the original posting.

Implementation:

Figure 5:
 A screen shot of the web page.  This same information is available
from individual links (on the bottom half of the page) that will
return just the information requested, making it trivial to obtain
individual data points using something like WGET.
Click on the image for a larger version.

As mentioned, with the WiFi capability of the ESP8266, the information that this device records is available via a web page on the wireless network:  One only need enter the SSID and wireless password at compile time.  For reasons obvious to anyone familiar with the Internet, this device won't be accessible from the web itself, but only to devices on the local network.

With the CW generator operating on 28.57 MHz, this signal lands within the 10 meter amateur band - and with this device being co-sited with receivers, it is a simple matter of tuning to that frequency to hear the telemetry.  Even though the RF output power is on the order of 50 milliwatts, the actual transmit "antenna" is a very small piece of wire - large enough to radiate just enough signal to be heard via the nearby antennas but not nearly enough to exceed FCC part 15 rules, eliminating the need for this device to transmit an FCC-issued callsign.

Comment:

This device was installed at the Northern Utah WebSDR shortly after this article was originally posted.  You may hear the Morse portion of this device via this link. 

This page stolen from ka7oei.blogspot.com

[END]

A quick look at the QB-300 RF amplifier

Available from many surplus sellers (e.g. via EvilBay) - and (usually) for a reasonable price - is the QB-300 RF amplifier.  Originally made by Q-Bit corporation, this same device has borne several different manufacturers markings over the 30+ years since it was introduced - but it is (pretty) much the same device.

Figure 1:
The BNC-connectorized version of the QB-300.
This appears to be the "original" version, actually made by Q-bit
Corporation.  The voltage specification is slightly ambiguous, being
shown as "+15/24 Vdc".
Click on the image for a larger version.

Having several of these on-hand I decided to take a quick look at its apparent performance - with the general specifications for this device being listed below for your convenience:

• Frequency range:  1 MHz-300 MHz
• Gain:  23dB (or 24.5 +/- 1 dB, depending on source)
• Gain flatness:  1 dB
• Noise figure:  3.8dB (frequency not specified)
• Input/Output VSWR:  <=1.5:1
• Power output (1dB compression):  +22dBm
• 3rd order Intercept:  +37dBm
• Current consumption:  155mA (voltage not specified)

Depending on which data sheet you consult, there are a few discrepancies - for example:

• The data sheet from "API Technologies" shows the input/output return loss as 1 dB - clearly a typo.

• The maximum voltage rating is all over the map:  Some versions of the data sheet show a maximum of 20 volts, others show 24 volts.  The units that I have clearly show the voltage rating as being "+15/24Vdc" and the equipment from which it was pulled provided 24.0 volts.

Knowing the provenance of this equipment, I would have no problem running my amplifier from 24 volts, but based on the ambiguity of the data sheets, I would operate a version that did not explicitly specify 24 volts ONLY from 15 to 18 volts.

A quick test:

Curious about a few aspects of these amplifiers I decided to test it with my DG8SAQ Vector network Analyzer, checking its gain versus frequency in the input matching (e.g. S11) - the results being displayed in Figure 2, below:

Figure 2:
A sweep of the amplifier from 100 kHz to 500 MHz showing the apparent gain and input matching over the frequency range.  Because the DG8SAQ and the interconnecting cables are increasingly imperfect with increasing frequency, expect increasing uncertainty in the S11 readings above 100 MHz or so.
The gain, S11 and VSWR values at specific frequencies can be seen in the upper-left corner.
Click on the image for a larger version.

Of particular interest was the usability of this amplifier above and below its "official" frequency range - and we can see that it's probably useful down to at least 250 kHz and above 450 MHz, albeit at reduced performance (e.g. lower gain, maximum output power, increased noise figure, increased input VSWR.)  Specifically, I measured, below the 1 MHz minimum frequency specification:

• The gain at 285 kHz was still above 21dB - a bit more than a 2 dB drop from the peak gain.  The input VSWR was still below 1.5:1.
• The gain at about 200 kHz was around 18.7dB
• The gain at about 173 kHz was around 16.8 dB and the VSWR had increased to about 2.7:1. 

Above the 300 MHz maximum frequency specification:

• The gain was above 22dB at 400 MHz
• The gain was about 21 dB at 440 MHz
• The gain was around 18dB at 500 MHz.

The reader is reminded that it is likely that above and below the rated frequencies that the maximum output power - not to mention noise figure - is likely to degrade.

Gain versus operating voltage:

Figure 2 was captured with the unit operating at 18 volts and readings were taken at lower voltages, comparing the gain - but your mileage may vary:

• Gain dropped by approximately 0.1 dB at 15.0 volts.
• The gain was about 0.2 dB lower at 12.0 volts than at 18 volts.
• The gain was about 1 dB lower at 8.0 volts than at 18 volts.
• The gain was about 5 dB lower at 5.0 volts than at 18 volts.
• The amplifier began to exhibit signs of low-frequency instability below 5 volts.

Although not directly measured, one should expect the maximum output power (P1dB) and the intercept point to drop below the specifications when operating it from lower than 15 volts:  The amplifier is likely to be perfectly usable in the 12-14 volt range, but it's likely marginal at 8 volts and below.

A peek under the hood:

Popping the top cover, we see this:

Figure 3:
A look inside the QB-300 amplifier:  The input and output is on the left and right sides, respectively.
Click on the image for a larger version.

 It is immediately apparent that this is not a run-of-the-mill consumer device:  Rather than a circuit board, the unit is built onto an alumina substrate with both soldering of components and spot welding of wires being used.  Two RF transistors are obvious:  The black, 3-lead device near the upper-left corner and the white ceramic device marked with "Q-21" just to its right.  The rest of the components are likely related to feedback/equalization as well as regulation of the operating and bias voltages for the RF devices.

Clearly, it's not hermetically sealed or conformally coated, so  weather protection is certainly warranted if this were to be operated outside.

Uses for this amplifier:

This amplifier was designed as a general-purpose gain block in the HF-VHF range, but it is likely useful into the low UHF range meaning that it should work from the 630 meter amateur band (on the low end) into the 222 MHz - and possibly the 70cm - amateur bands on the high end.

For general HF (amateur radio) amplification purposes, it should be an excellent performer - provided that one keeps in mind that it's gain may be a bit too high in certain applications in that a signal input level of a around -5dBm will push it into overload - and off-air signals of this strength might appear from:

• Local AM broadcast stations.  Especially on a long wire antenna (longwire, rhombic, end-fed half-wave) these signals can, by themselves, overload the amplifier if you live anywhere near  a transmitter.  A simple high-pass filter can effectively reduce such signals and prevent overload.

• High-power shortwave stations.  On a good antenna, signals on the 49, 41 and 31 meter band can be extremely strong in Europe and some parts of the U.S.

If you have strong signals that could overload the amplifier, beware using an attenuator on the input of the amplifier in your receive system.  As an example, if you wish to be able to hear the background noise at 10 meters to be able to hear the weakest possible signals you will need to make sure that your system noise figure is no more than about 15dB - but if you had a cable loss of 3 dB in "front" of your amplifier (between the antenna and the amplifier) and you used a 10dB attenuator in this signal path, you are already at 13dB - and the nominal 3.8 dB of noise figure of this amplifier will push that number to about 16.8dB meaning that your system noise will now likely be high enough that you can no longer hear atmospheric noise if you are fortunate enough to be in a very "RF quiet" location.

In short:  If you hear more noise when you connect your antenna system to your receiver system than when you connect it to a dummy load, you are OK - but if you can't hear the difference, your system will not be sensitive enough to hear the weakest signals.

For receive-only purposes it is often the case that with a low-noise amplifier, a good, quiet (in terms of noise) receive antenna will not need to have much gain from the antenna itself - and if the gain is low, you are less likely to intercept enough absolute signal power to overload the amplifier.  Here are just two of the many possible examples of antennas to consider:

• Small receive loop.  This type of antenna - usually around 3 feet (1 meter) diameter for MF and HF use can offer local noise rejection as well as the ability to null signals from directions broadside the plane of the loop.  This type of antenna will have negative gain (e.g. less than 0 dBi) but its performance can be quite good with a decent, low-noise amplifier like the QB-300.  For an antenna like this, one would place the amplifier at the antenna to minimize cable losses.

• Beverage on the ground.  Also known as the "BOG" antenna, this is simply a wire - as long as possible - laying on the dirt and working against a good (and electrically quiet) ground consisting of one or more ground rods and counterpoise wires and its feedline electrically decoupled (with a "current" balun) to prevent noise from the shack from being brought to the antenna.  This antenna - mostly useful in rural areas - is reported to work well overall despite the likely "negative" gain.  As with the receive loop, it's best to place the amplifier at the antenna feedpoint.

Amplifier and receiver protection:

It should go without saying that any amplifier (or receiver) connected to a large antenna should be preceded by adequate lightning protection to prevent damage to the amplifier from wind static/discharge and nearby lightning strikes as depicted in Figure 4, below.  Such filtering should be placed after any filtering that might precede the amplifier.

Decent protection can be had with four ordinary silicon diodes - two series pair connected anti-parallel (back-to-back) with a bleed resistor (4.7-100k) to shunt voltages above about 1.2 volts.  It's worth noting that the amplifier itself would already have overloaded before signals can a high enough level to cause the diode protection to conduct and cause distortion!

Figure 4:
Depiction of simple input protection circuitry.
On the left, the diodes ("D") are ordinary silicon diodes connected in series for approximately 1.2 volts of conduction.
On the right, a common full-wave rectifier module is used with its DC "output" shorted, providing an equivalent to the circuit on the left.  It is suggested that a low current (2-5 amp) rectifier be used.
The voltage rating of the diodes is not particularly important - a 50-100 volt rating being just fine.
Resistor "R" is not critical and can be anything from 4.7k to 100k and it is used to dissipate any accumulated DC in case the antenna itself does not have a DC ground.  An inductor can be used in addition to or instead of "R" - a value of 22-100uH (e.g. 8-10 turns on an FT50-75 toroid) being suitable for 630-10 meters.
Click on the image for a larger version.

Provided that one avoid excessive signal input level, it can also be used as the basis for a receive multi-coupler.  For example, following the amplifier with an 8-way RF splitter - which, itself, will have a loss of around 10dB - the overall gain will be in the range of 14dB while preserving the system's overall noise figure to allow reception of weak signals on the higher HF bands.

This page stolen from ka7oei.blogspot.com

[End]

A "curiously sharp" 40 meter band-pass filter to reduce 41 meter SWBC overload.

At the Northern Utah WebSDR (sdrutah.org) we recently added another server (WebSDR #4) that is connected to an existing east-pointing beam antenna on site. This antenna, it is hoped, will better-allow users to hear stations on the 40-10 meter bands in the eastern U.S. and, to a lesser extent, the DX locations to which it is pointed.

Figure 1:
East-pointing beam antenna at the Northern Utah WebSDR.
This antenna has 10-13 dBi gain and signals in the 39,41
and 31 shortwave broadcast bands can be extremely strong!
Click on the image for a larger version.

As one would expect, this antenna has gain - between 10 and 13 dBi, depending on the frequency - and this has implications when propagation between Utah and the Eastern U.S. is favorable:  Already-strong shortwave broadcast (SWBC) signals become even stronger.

Because the S-meter (signal meter) on the receivers have known calibration it is possible to make an indirect measurement of some of these signals' strength and, at times, individual signals have been observed in the -20 to -15 dBm range at the antenna - these levels being in the "60 over S-9" range.  At times - particularly during evening "gray line" propagation (where both the transmit and receive sites are entering/in twilight/evening) signals can peak significantly.  What's worse is that there may be several such signals, increasing the total RF power impinging on the receiver risking not only receiver overload, but also providing a ready source of multiple, modulated carriers to mix together and reappear within the receiver's passband among the desired signals.

These sorts of signals are far above those that might be expected due to amateur-only transmissions owing to the widely disparate signal levels.  For example, a very well-equipped "DX" stations may be able to run 1500 watts of RF into a (monster!) 15 dBi gain antenna and attain an EIRP (Effective Isotropic Radiated Power) in the area of 50kW, but this does not compare with an SWBC station which may be running 500kW peak (about 125kW carrier) into an antenna with (a conservative) 18dBi gain - an EIRP of about about 32 million watts - a signal level nearly 1000-fold (30dB) stronger than one that would be transmitted by law-abiding amateurs.

What's worse is that some of these SWBC bands are adjacent amateur bands - and the 40 meter amateur - which runs from 7.0-7.3 MHz is no exception as the 41 meter SWBC band is just above it, starting at 7.3 MHz.  With such close spacing, typical filtering in receivers have little hope in effectively rejecting these nearby, strong signals.

Addressing the problem:

There are two time-honored ways of dealing with strong signals impinging on receivers:

• AGC (Automatic Gain Control):  This circuit "monitors" the signal level at the receiver and automatically reduces the gain when they exceed a certain amount.  In the past, this has been applied only to signals within the passband of the receiver's IF to keep the audio level constant regardless of the actual signal strength, but this is also applied to modern SDRs where the level of the entire passband of signals being input to the A/D converter is monitored and adjusted to prevent overload. 

• RF front-end filtering:  With the advent of solid-state radios starting in the 60s and 70s the design of RF filtering used in amateur receivers began to be wideband, typically covering MHz, rather than a narrow peak.  This was done not only because it was easier to do so with these designs, but also because it allowed "general coverage" reception outside the amateur bands and it was significantly less expensive than mechanically-complicated, ganged tuning systems - but it had the down-side that signals some distance away frequency-wise could still cause the receiver to experience overload.  These days - particularly with modern, high-performance direct-sampling Software-Defined Radios (SDRs) - "narrow" filtering is once again being used, along with AGC, owing to the need - more than ever - to strictly control the total amount of RF energy reaching the A/D converter to prevent overload. The receiver used at the Northern Utah WebSDR is a type of SDR where the RF energy is converted directly to audio and then digitized.  This has the advantage of simplicity, but it lacks the "AGC" circuit meaning that it is possible for strong, off-frequency signals to cause overload of not only the RF circuits, but also the audio circuits and the A/D converter.

While it is possible to add an AGC circuit to this receiver system to prevent overload (this has been done with some of the other receivers on site - and is still an option) the first step that we are taking is to build a "sharper" filter.

The "Curiously Sharp" band-pass filter:

Passing signals on the 40 meter amateur band - which ends at 7.3 MHz in the Americas - and filtering out signals on the 41 meter shortwave broadcast band - which starts at 7.3 MHz - is a tricky proposition:  How does one suddenly go from passage of signals to blocking them within just a few 10s of kHz?

Figure 2:
The completed 40 meter band-pass filter
in a Hammond 1590D die-cast box.
Click on the image for the larger version.

The answer is:  You don't - but you do the best that you can!

The limiting factor in constructing a "brick wall" filter - one that has an abrupt transition - is physics and is intrinsic to real-world components:  Real-world inductors have ohmic resistance and capacitors have dielectric losses - to name but two factors - that limit the unloaded "Q" of the circuits.

What does this mean?  A truly "sharp" filter will ultimately be limited in its performance by these factors:  One must trade off insertion loss and/or filter performance in terms of how quickly our band-pass filter cuts off.

Fortunately, the first of these - insertion loss - is pretty easy to mitigate:  Have enough extra signal gain in the receive system to accommodate the insertion loss.  At 40 meters, we have "signal to burn" - partly because our receive antenna has so much gain, but there is also a "strong" (resistant to overload) RF amplifier located near the antenna to mitigate the effects of cable losses at the higher HF bands (10 meters).

Even if we didn't have both antenna and amplifier gain, we could afford to lose a lot of signal at 40 meters:  A system noise figure of about 30 dB (assuming a unity gain antenna) is sufficient to "hear" the noise on even a quiet band, so a significant loss can still be made up by placing an RF amplifier after the filter and still be able to resolve the 40 meter noise floor during quiet band conditions.

Figure 3:
 Schematic of the 40 meter bandpass filter.  This is a 7-element Elliptical (Cauer) filter centered at 7.15 MHz - the middle of the U.S. 40 meter amateur band.  It was originally designed with the aid of the "A.A.D.E. Filter Design" program, version 4.5 being available from the AE5X web site.
The nominal impedance of the filter portion is 800 ohms to permit higher values of inductance and lower values of capacitance in an effort to ease construction and to reduce component losses (e.g. reduce the L/C ratio).
Click on the image for a larger version.

Figure 3 shows the schematic of the filter - and a few explanations are warranted.

• ALL of the capacitors must be either NP0 (a.k.a. C0G) ceramic or silver mica capacitors - preferably the latter.  I did not use any silver mica capacitors, but I used known-good ceramic capacitors from a trusted source (e.g. Mouser-Key) rather than from a random EvilBay seller.
• L1, L4 and L7 were wound using solid 12 AWG copper wire.  The wire that I used happened to be tin-plated, but enameled copper wire will be just fine with only the two ends (and the tap point) being bared for soldering.  If bare copper wire is used it is suggested that it be very clean and sprayed with clear lacquer after construction is completed to prevent oxidation.
• The other inductors were wound using 17 AWG wire, which was on hand, but 18 AWG would be fine.
• All of the inductor/capacitor pairs have their own resonant frequency, noted on the diagram in parentheses.  The 7.15 MHz resonances (C1/L1, C4/L4, C7/L7) will be adjusted very close to the stated frequency but the other resonances (C2/L2, C3/L3, C5/L5, C6/L6) are made adjustable by small ceramic (or air) variable capacitors and must be CAREFULLY adjusted for the proper filter response.
• As can be seen, the filter's in/out ports are terminated with 2dB resistive attenuators to help assure a consistent source/termination impedance to the filter and prevent the likely-imperfect devices to which it is connected from too-badly affecting the response.
• L1 and L7 show taps that are chosen to be at the 50 ohm points.  The "S11" port of a known-calibrated VNA may be used to best-set the 50 ohm points of the taps during filter construction/adjustment.

Constructing the filter:

Figure 5, below, shows the as-built filter:

During construction I used my DG6SAQ Vector Network Analyzer - and a tool such as this is invaluable as it will give "live", dynamic readings to facilitate adjustments.  The more economical (approx. $50 U.S.) "NanoVNA" will work fine (along with the "NanoVNA Saver" program) - and although its update/sweep rate is quite a bit slower than that of the DG6SAQ, it's still usable.  No matter what sort of VNA you might use, be aware that the limited number of data points per scan can "hide" details such as narrow, deep notches - and this is especially true with the NanoVNA.

The "through loss" measurements (in dB) were the most important in this case as the insertion loss versus frequency plots over a range of about 6.5 to 7.8 MHz allowed the "dialing in" of the resonant circuits.  During construction two "bloody ended" coaxial cables were used - one end of each being connected to the VNA and the other end having its ground shield tacked to the ground plane and the center conductor attached to the point under test:  These test cables are visible in Figure 4, below.  This "plywood and foil" test bed allowed an easily reconfigurable circuit design and test bed for ideas - and, most importantly, it helped me determine if a particular design was even practical.

Figure 4:
Early prototype built on a piece of plywood covered with self-adhesive
copper foil.  Originally, L1, L4 and L7 were wound on toroids - but
a switch was made to the larger, air-wound inductors to reduce losses.
This early version used input/output transformers for transformation of
the 50 ohm in/out to the 800 ohm (nominal) impedance of the filter itself -
but this was changed to tapped inductors as that was simpler and
lower loss.  This simple "breadboard" allowed several ideas to be tried
before settling on the final version, giving plenty of room to work.
This picture shows the short pieces of coaxial cable that were
tacked to the foil ground:  These cables connect to the two
ports of the VNA used to analyze the response of the filter.
 Click on the image for a larger version.

The first to be constructed were the large resonators (L1/C1, L4/C4, L7/C7) which needed to be set to 7.15 MHz and for this, two resistors (1k-4.7k - the precise values are unimportant) were connected in series with the center point connected at the "top" end of the parallel L/C network and the "ends" being connected to the VNA's in and out ports.  With this arrangement one can see the "peak" where the L/C circuit resonates - the two resistors minimizing loading - and one compresses/stretches the large inductor using a small screwdriver to increase spacing between turns or a pair of needle-nose to compress them - or, if necessary, removes fractional turns - to "dial it in" at 7.15 MHz.

After these have been adjusted, the other L/C networks are then constructed - and this is where it gets to a bit tricky:  The variable capacitors allow the resulting "notch" to be moved around, but it may be necessary to add/remove turns from the inductor - or add small amounts of capacitance (10pF at a time) to get the circuit's adjustment within range of the variable capacitor.  In some cases, one may temporarily "shunt" (short out) one or more of the series L/C networks to better-visualize the notch that one is trying to adjust.  If you can't find the "notch", don't forget that it may be above/below the sweep range and you may temporarily need to set the start/stop frequencies wired to find it.

As often happens, one's first ideas don't work quite as expected:  You will note that in Figure 4, L1, L4 and L7 are shown as being toroidal inductors, but it became clear that these inductors were just too lossy, a factor that severely affected "Q" and performance.  I ended up using air-core inductors wound from much heavier wire as can be seen in Figure 5 to minimize loss.  Ideally, superconducting inductors would have been used, but for some reason such devices that operate at room temperature aren't available!

Adjusting such a filter requires patience as everything interacts.  An examination of Figure 5 will reveal that each section is connected with jumper wires, allowing isolation of the individual tuned circuits.  Eventually, one can get a "feel" for how the adjustments interact - but it may still be necessary to  disconnect the sections and check/tune them individually back to a starting point if one gets "lost" and the response/tuning gets worse and worse.

Also visible in Figure 5 are shields around the large tuning elements made from pieces of double-sided copper-clad PC board material.  While shielding between the sections isn't really necessary from a performance standpoint, placing the filter - which was constructed on the lid of the Hammond 1590D box - into the box itself causes the filter to detune slightly due to proximity to the enclosure's walls:  The shielding on the sides of the large coils - and the bars across the top - "simulated" the filter being within the die-cast box and almost eliminated the effect, but still allowed access to permit adjustment if the large coils.

Figure 5:
As-built 40 meter band-pass filter.
This filter was constructed on a solid copper ground plane of circuit-board material.  to hold components in place and to isolate junctions from the ground.  "Manhattan" (island) pads were used for junctions that needed to be isolated from the ground:  The "Me Pads" (from "QRPMe") were used.  Blobs of RTV are used to mechanically support some of the larger components.
Click on the image for a larger version.

 

To be clear:  This should NOT be your first band-pass filter as it is VERY tricky to adjust - and you MUST have available a scalar and/or vector network analyzer to properly adjust it!  If both of these do not apply to you it is suggested that you obtain help - or prepare to get this gear and pull your hair out during adjustment!

Did it work?

The answer is Yes.

Figure 6:
A sample passband of the filter during adjustment:  The ultimate adjustment resulted in a somewhat flatter response.
The "upper" notches (L2/C2 and L5/C5) can clearly be seen as can the upper "lower" notch (L6/C6).
The intrinsic insertion loss, including the two 2 dB pads, is around 15dB.  The ultimate rejection is around 65 dB, correlating to a filter rejection of around 50 dB, taking into account the through losses.
Click on the image for a larger version.

 

This filter offers over 20dB of (additional) attenuation below 6.9 MHz and above 7.4 MHz and between careful adjustment of the receive system gain (e.g. just enough signal to comfortably "hear" the noise floor during the quietest part of the day) and the attenuation of the 41 meter signals, overload on the 40 meter receivers on the Northern Utah WebSDR #4 no longer occurs.  If you wish, you can check it yourself, particularly during the evening "gray line" hours when sunset is sweeping across North America at sdrutah.org.

Comments:

The use of a similar filter in ITU Regions 1 and 3:

In Regions 1 and 3 the 40 meter amateur band covers 7.0-7.2 MHz with strong SWBC signals starting at 7.2 MHz.  Narrowing this filter to 200 kHz would require a redesign and would further-push the limits of standard components, but broadly similar results should be possible.  Alternatively, the center frequency of this filter design could be moved down by 100 kHz to 7.05 MHz and offer similar rejection to signals above 7.2 MHz.

Options for even "sharper" filtering:

While the filter described is starting to push the limits in terms of the use of reasonably-obtainable components, there is another option:  A frequency-converting band-pass filter.  For this, a local oscillator and a pair of mixers would be used to convert the 7.0-7.3 MHz 40 meter passband down to a lower frequency where a "sharper" filter would be easier to construct.

For example, using an 8 MHz oscillator would convert the 40 meter band from 7.0-7.3 MHz to 0.7-1.0 MHz, inverting the frequency, meaning that the most critical part of our filtering - that "above" 7.3 MHz - would now be happening below 700 kHz.  Of course, this "converting filter" would have to have decent band-pass filtering of its own to prevent response to undesired signals and the mixer used for the down-conversion would have to be adequately "strong" to withstand the insanely strong 41 meter signals.

Once the filtering is done at this lower frequency, the same 8 MHz oscillator would be used to up-convert back to 40 meters:  With the same oscillator used in both directions, it need not be particularly frequency-stable as any drift would be self-compensated.

This page stolen from ka7oei.blogspot.com

[End]

About the WSPRDaemon noise graphs - and repurposing for use in monitoring WWV(B) signal levels over time.

What is WSPRDaemon?

The WSPRDaemon program is largely a work of Rob, AI6VN and it exists for the purpose of facilitating the reception of WSPR signals off air, processing the data, and making that data available publicly - not only to wsprnet.org, but also via the wsprdaemon.org web site.  The WSPRDaemon script runs on a Linux computer - typically a Raspberry Pi - and typically takes data from a web-connected receiver (typically a KiwiSDR) - but it can also use the ubquitous RTL-SDR dongle or even the "raw" audio input from a receiver via a sound card.

For each of the (approximately) 2 minute receive "cycles" of WSPR transmissions (which are, worldwide, scheduled to start precisely at the beginning of each even-numbered minute) an audio (.wav) file is recorded for each receiver - and this audio is then processed using the "WSJT-X" program to decode the WSPR transmissions found within that audio stream.

Features not found in WSJT-X that are made available in the wsprdaemon script include the scheduling of receivers (e.g. switching a limited number receivers to most favorable bands) and the "merging" of several receivers - possibly using different antennas for diversity - that are listening to the same band so that only the best of the WSPR "spots" is reported, avoiding duplicate reports of a given transmission.

• More information about wsprdaemon may be found on the wsprdaemon Github page.
• For more about WSPR, see the WSPR Wikipedia page and the other pages to which it links - especially K1JT's page.

What it does:

While not precisely calibrated, having long-term, disparate records of WSPR signal reception from all over the world can provide useful information to amateur radio operators and researchers alike, providing a glimpse into the propagation of LF, HF and even VHF (and higher bands) which can help divine when propagation is occurring between two stations and when it occurred.  Having a "live" and past database of these events can help validate/tweak models of the Earth's geomagnetic field and its interaction with the sun on the ionosphere - and to satisfy the intellectual curiosity of anyone who wishes to study this by themselves.

Accumulation of "noise" data:

In addition to the accumulation of WSPR data, the wsprdaemon software is able to measure the apparent noise floor on the specific frequencies associated with WSPR transmissions on the amateur bands and with some hardware it is possible to calibrate this measurement in absolute terms of intercepted noise power.

If receivers taking these readings are found at disparate "RF quiet" sites around the world, this data can be informative of the natural, background noise which can be indicative of the state of the ionosphere and the Earth/Space environment:  In some cases, it is possible to observe the rising/setting of strong radio noise sources such as Sagittarius A and, occasionally, "noisy" planets in our solar system such as Jupiter not to mention the tremendous noise that can result when the Earth's magnetic field is being abused by our local star.

The software makes two separate noise measurements:

• The noise "floor" within the passband.  This reading (the "RMS" level, in Red on the wsprdaemon.org graphs) is the calculated noise floor level and its processing attempts to "remove" the effects of other signals within the detection passband.

• The signal power within the passband.  This reading (the "FFT" level, in Blue on the wsprdaemon.org graphs) is the total power within the passband.  Unlike the RMS reading, this is indicative of the cumulative power intercepted and will always be higher.

Even though the apparent detection bandwidth of these receivers is on the order of 400 Hz, all readings are scaled (by approximately 26 dB) such that the power readings reported are relative to a 1 Hertz detection bandwidth.

A typical graph may be seen below:

Figure 1:
A typical graph showing the last 24 hours of noise on the 40 meter amateur band a receiver at the Northern Utah WebSDR.

The displayed time, in UTC, shows the noise level rising at around local nighttime and then dropping off at night.  This graph is "skewed" somewhat by the overnight presence of strong thunderstorms in the Eastern United States, the intensity of which gradually tapered off overnight and into the day and by the fact that the antenna used is a log periodic beam has significant gain that is pointed in the direction of those storms.

Almost the "inverse" of this is the noise graph from another receiver at the Northern Utah WebSDR site:

Figure 2:
This graph depicts a 24 hour plot of the noise floor on the 20 meter band.

The noise floor can be seen to increase during the daylight hours, but drop to the floor during the night when propagation and ionospheric stimulation by the sun effectively ceased.  As expected - particularly during the period of low sun activity during which this is being written - the noise floor and signals decrease during local nighttime.

In theory, careful analysis of the noise data used to produce the above graphs can provide the opportunity to analyze HF propagation modes and the effects of Earth-Space environment.

Monitoring signals from consistent sources:

It occurred to me that the wsprdaemon script also afforded the opportunity for something else for which it was originally intended - the monitoring of consistent signal sources of known transmitter power and location.  From the Utah location, one source of signals - those from the NIST in the form of the WWV and WWVB  transmitters - was obvious.

A quick modification of the wsprdaemon script allowed the addition of additional frequencies:  While there would clearly be no WSPRnet reporting on these non-amateur channels, the noise measurements would still be posted as the graph below depicts:

Figure 3:
 A noise graph of the WWVB transmission at 60 kHz in Fort Collins, Colorado.

The red line shows the received signal level from WWVB - the same signal used to set many automatic clocks - on 60 kHz.  During daylight hours the signal level is pretty consistent at the "-100dBm" mark while during the night, signal levels vary a bit - particularly during sunrise/sunset where ionospheric perturbations are evident.

The "Blue" line is largely influenced by the 17dB amplitude modulation of the WWVB carrier used to convey time and date information, but it is also prone to being "diluted" by peaks in the background noise as can be seen during the nighttime hours (from about 0400-1100 UTC) where propagated lightning static is evident.

From the same site, on a frequency that is orders of magnitudes higher we get this plot:

Figure 4:
 A noise graph of the signal and noise levels on the 10 MHz WWV/H frequency.

The graph of Figure 4 is a bit more cluttered as one might expect.  The Red "line" shows the wildly varying signal of the signal on 10 MHz - which could be from either WWV in Colorado OR WWVH in Hawaii.  As is the nature of HF, these signals can vary significantly - not only between day and night, but also from one moment to the next.  The blue line generally depicts the noise level at 10 MHz, but this may not be truly representative as it may be being affected by the ever-present modulation on the WWV/H carrier - primarily in the form of the 100 Hz time code modulation.

The utility of the graph in Figure 4 may be debatable as there is not one, single signal source, but it does provide a general perception of the signal levels that one might expect - and how the time of day affects them.

Final comments:

In addition to the general monitoring of the noise floor on the HF bands where WSPR monitoring is taking place, the wpsrdaemon script can also be used to monitor signals from known transmitters.  To be sure, this wasn't the intended use of this software and if such data is useful, it's likely that the utility and accuracy of such measurements could be improved.

* * * * * * * * * * * * * * * * *

Modifications to the wsprdeamon script for version 2.8a:

In the "wsprdaemon.sh" file, one need only add a few lines to the code to produce the "new" bands in the array "WSPR_BAND_LIST", as in:

"WWVB        58.5"
"WWV_2_5  2498.5"
"WWV_5      4998.5"
"WWV_10     9998.5"
"WWV_15    14998.5"
"WWV_20    19998.5"
"WWV_25    24998.5"
"CHU_3       3328.5"
"CHU_7       7848.5"
"CHU_14     14658.5"

Each new "band" is named by the first entry on the line (e.g. "WWVB") and the frequency of the carrier that one wishes to monitor is defined in kHz in the second entry.  Note that the frequency used here is 1.5 kHz lower than the actual carrier frequency to be monitored.

In the "wsprdeamon.conf" file where the receiver and its use is to be defined, these "bands" - defined in wsprdaemon.sh - are used in exactly the same way as any other band in the list.  For example, one might schedule the start a hypothetical receiver called "KIWI_1" on the WWVB signal at 0000 local time as follows:

declare WSPR_SCHEDULE=(
"00:00  KIWI_1,WWVB"
)

Comment:  At present, wsprdaemon will dutifully try to process the audio file for WSPR spots - but it will fail to do so.  It should be possible to modify the code to add an argument that will prevent this from happening to reduce processor loading.

* * * * * * * * * * * * * * * * *

This page stolen from ka7oei.blogspot.com

[END]

Setting/calibrating signal and noise levels for signal/noise level monitoring using the "wsprdaemon" script

The WSPRDaemon script, a work largely of Rob, AI6VN, is designed to automate WSPRNet reporting of signals transmitted using the WSPR protocol - (The script may be found on GitHub, here).  While the WSJT-X program, itself, can report the signals that it hears in terms of callsign, Maidenhead grid locator and apparent signal-noise ratio, the WSPRDaemon script leverages the multi-receiver capability that can exist in various hardware configurations - including the KiwiSDR - which, with the appropriate platform, can receive as many as 15 signals on 15 different frequencies simultaneously.

Comments:

While this discussion is focused on methods and measurements exclusively using the KiwiSDR as that is the hardware with which I am most familiar, where similar capabilities exist, these same methods can be applied to other receive platforms.

Similarly, the methods discussed on this page need not be specific to the software mentioned and may be applied any time receiver signal level measurement is needed.

Care has been taken such that at the Northern Utah WebSDR, all of the WebSDR's receivers' S-meters are calibrated to within a dB or so in the center of their target band using the methods described below.

HF noise floor measurements:

In addition to facilitating the decoding and reporting of WSPR signals, the WSPRDaemon script can also divine the apparent noise floor within the receive bandwidth on some of these platforms.  In the case of the KiwiSDR - where an "S-Meter" reading is available via the remote interface - this reading can be used to derive absolute measurements and such data, taken from multiple locations over disparate geographical locations, can be used to glean information about the noise on the HF bands over time.

In practice, accurate, absolute signal level readings are quite difficult to obtain owing to the issue of meaningful calibration - and the more complicated the antenna system is, the more difficult it may be to get such measurements as one must consider things like:

• Cabling losses
• Splitter losses
• Filter losses
• Amplification
• Calibration of the receiver
• Gain of the antenna (e.g. dBi, dBd)

Fortunately, all but the last on the list may be accounted with a single measurement using a known-accurate signal source.

The "short" version:

The ultimate goal of this exercise is quite simple:

Input a signal of known amplitude into your antenna system - say, -60 dBm - and have the noise logger report that same signal level, +/- your known antenna gain.

The steps below are included in the event you may not be familiar with the methods/techniques involved.

An example receive system:

As an example, the KiwiSDR-based portion of the system in use at the Northern Utah WebSDR (a browser-enabled remote HF receiver found at sdrutah.org) will be discussed and a simplified block diagram of the signal path may be seen below:

Figure 1:
Simplified KiwiSDR Signal Path used at the Northern Utah WebSDR showing several gain blocks and filters.
As noted, the directional coupler is connected in "reverse" as it is being used to inject rather than extract a signal from the signal path.
Click on the image for a larger version.

The components of the receive system depicted above in detail:

As with any antenna system, it begins with an antenna - but for the moment, we will ignore any intrinsic gain - or loss - that the antenna may have.

• Directional coupler:  At the Northern Utah WebSDR, the Mini-Circuit Labs ZFDC-20-3 is used at the main antenna input.  This device has minimal insertion loss (<0.25dB) on the "through" line and there is a separate port (labeled "CPL") that is coupled at 20dB from the through line.  This sort of device is typically used to extract a small sample present on the line while minimally affecting it and "in" and "out" ports are labeled for such purposes, but here it is being used in reverse to inject a signal into the receive signal path.

This device is convenient in that it has the advantage that it may be left in line all of the time with negligible effects at HF and it can put test signals into the receive signal path without affecting off-air performance.

This device is optional in that the same measurements may be obtained without it by connecting the signal source directly to the antenna port - the obvious disadvantage being that the signal path is interrupted - something that we try to avoid on a busy, multi-user system such as a WebSDR.

• Calibrated signal source:   It should be no surprise that having a known-accurate signal source is an absolute necessity when it comes to calibrating the signal level readings with available levels in the range of -60 to -30dBm being the most useful.  Having access to such devices (e.g. Communications Test Set or RF Service Monitor) is helpful - but other means of obtaining a usefully-accurate signal include:

• An "inexpensive" calibrated signal source like the Elecraft XG-3 - the >$300 price tag (when fully equipped) being the reason for the quotes.

• A "transfer" measurement from a signal source with a consistent level.  Some antenna analyzers - including the popular "NanoVNA" - can output a level that is fairly consistent.  Coupled with outboard attenuators (a switchable step attenuator and/or an assortment of fixed attenuators) one can use known-accurate test equipment (an RF power meter, signal level meter, spectrum analyzer) to obtain a table of signal power readings for each of the HF amateur bands.

• Because of the harmonic content of the NanoVNA, it is preferred that a selective means of measuring RF power like a spectrum analyzer be used rather than radiometric detector like a power meter - but the difference is unlikely to be significant.

• Important note:  The amplitude of the NanoVNA's output port is too high to be directly used as a calibration signal - especially if amplification is used in the signal path.  Expect to need attenuation in the range of 20 to 50 dB to provide a usable signal level:  It's recommended that one use a switchable step attenuator and/or a series of fixed attenuators to get reasonable signal levels.

• A transfer measurement from a piece of equipment that is likely to be "pretty close" - such as the KiwiSDR itself:  With the default "factory" S-meter calibration level of -13dB the KiwiSDR to which I have access have typically been within a couple of dB when comparisons were made.
• For calibrating, a signal source (e.g. NanoVNA) would be connected directly to the input of the receiver - via the attenuator(s) - and the S-meter levels noted for different settings on each of the amateur bands.

• AM BCB Filtering:  In many cases, strong AM broadcast band (mediumwave) signals will be present on the antenna port.  Even if these signals are not strong enough to overload the RF amplifier that follows, such signals can overload the receiving device downstream.  As mentioned later, there are advantages to strategically attenuating strong signals to minimize the total power impinging on the A/D converter of any direct-sampling receiver.
  

The AM broadcast-band filter used at the Northern Utah WebSDR is a rather complicated affair (and is described in this article:  Managing HF Signal Dyanmics and preventing overload... (link)).  Its main purpose is to reduce the amplitude of several very strong (50kW) local transmitters to prevent the "wasting of amplifier power and A/D bits" on very few signals.

• Amplification (post AM/BCB filtering):  In general, a 15 dB system noise figure is sufficient to allow the detection of a 10 meter signal above even the lowest expected level of background noise found at this frequency.  For this reason it is suggested that an RF amplifier be placed after the AM BCB filter (to minimize the probability of overload and intermodulation/mixing products).  Remember:  Once loss appears in a receive system due to loss/attenuation, no amount of amplification can restore the system noise figure beyond that point.

Such an amplifier should also be placed prior to any RF splitting as much as is practical as any loss - which can be from the feedline, in filters or in a splitter - will directly contribute to the the system noise figure.  In cases where very strong signals may be present on the antenna system (local AM BCB, strong Shortwave Broadcast signals - when the bands open) it is preferable to have an amplifier at this point that has only modest gain - in the area of 12-15 dB:  A high-gain amplifier (>20dB) may be more prone to overload in such conditions and the typical means of preventing this (addition of an attenuator in front of the amplifier) is not recommended as this will increase system noise figure.

• Limited attenuation high-pass filter:  Any receive system is capable of handling only a certain amount of total signal input power - and the direct-sampling receiver of the KiwiSDR is no exception.  The overload level of the KiwiSDR's 14 bit A/D converter is approximately -13dBm, so the total power arriving at its antenna port should be kept below that level and strategic, selective attenuation of local, strong carriers - often those of AM/mediumwave broadcast transmitters - should be considered - but there is another problem:

The sensitivity of a KiwiSDR is approximately -155dBm (in a 1 Hz bandwidth) over its frequency range which means that in a unity-gain HF antenna system that at 10 meters, the KiwiSDR will be "short" by at least 6-8dB from hearing the textbook "rural quiet" noise floor on that band, requiring 10-15 dB of overall amplification to place the 10 meter noise floor comfortably above that of the receiver's in that environment.  This amplifications has a cost:  With the addition of a 12 dB RF amplifier, the KiwiSDR will now overload at -25dBm.  It is not uncommon for the absolute power level of a single high-power shortwave broadcast station to exceed this level under good band conditions - and there will be many such signals on the bands - not including the wide-band energy from lightning static - that can combine in overall signal power and exceed this overload level.

What system noise figure is required for HF reception? This is a tricky question to answer because conditions on the HF band vary so much - most strongly on a per-band basis. Assuming quiet (e.g. "dead"), bands the lack of storm static. the absence of solar noise and the receiver being located in an area completely absent of man-made noise, noise figures equal to or lower should be attained for the receive system to have a "natural noise" limit: 1.8 MHz:  45dB NF 3.5 MHz:  37dB NF 7 MHz:  30dB NF 14 MHz:  24dB NF 21 MHz:  20dB NF 28 MHz:  15dB NF 50 MHz:  9dB NF 144 MHz:  2dB NF These numbers are approximate, but are a general guideline for system design.

Were the KiwiSDR being used over a very narrow range of frequencies - such as the case for most commercial direct-sampling receivers like the Icom IC-7300 and IC-7610 - this could be managed by gain adjustment in the signal path, but if the KiwiSDR is to be used over the entire HF spectrum, this cannot readily be done.  Instead, one can selectively insert attenuation where the signals+noise are likely to be the highest, namely below approximately 10 MHz and adjust calibration accordingly.

The use of this "limited attenuation" high-pass filter leaves signals above 12 MHz alone but offers significant attenuation (about 12 dB - around 2 "S" units) to signals below 8 MHz.  Because of the relatively high noise levels in the lower HF bands, additional attenuation - and the commensurate increase in system noise figure - can be tolerated without the loss of useful sensitivity.

Note:  This same issue of overload will occur with similar devices - such as the Red Pitaya - if operated as wideband HF receivers, without input band-pass filtering.  Simple math will indicate that even if the KiwiSDR had a 16 bit A/D converter - which would theoretically yield another 12 dB of dynamic range - one would still need to take these same precautions.  It is for this reason why all well-designed direct-sampling receive gear has narrowband filtering that precedes its A/D converter.

Unfortunately, the "limited attenuation" filter fits a niche requirement as is not a commercially-available device, but one may be built without too much difficulty:  Two versions are described here:

• A limited-attenuation high-pass filter for the KiwiSDR - link. This filter starts to attenuate below 12 MHz, offering about 12 dB below 8 MHz.

• Revisiting the limited-attenuation high-pass filter - link.  This version is similar to that above, but its attenuation pass range is limited  to the range of approximately 1700 kHz to 12 MHz, allowing the passage of signals in the AM broadcast band and below for the reasons discussed in the article.

• Because of the simplicity of these filters they exhibit a low return loss (high reflectance) at lower frequencies and  it is recommended that they be preceded with an amplifier to present the input feedline with a more consistent "50 ohm" impedance.

• Amplification (post high-pass filter):  Where high signal levels are likely to exist it is prudent to distribute the signal gain across several blocks rather than have a single high-gain amplifier near the "front" of the signal path.  An amplifier placed after the "limited attenuation" high-pass filter will see lower signal levels from the very strong lower HF-band signals and be less subject to overload.

• Splitter:  A 4-way splitter is depicted in figure 1.  A real-world 4-way splitter will have 6.5-7.5 dB loss between the input and any of its output ports which means that for the higher-frequency bands it is particularly important to have established the system gain and noise figure prior to the splitter.  Like any other block of gain or loss, the impact of such a device must be considered when evaluating the system overall.

Analyzing your receive system:

The procedure following this point assumes the availability of a known-accurate signal source:  See the heading "Calibrated Signal Source", above, for more information.

Again, we are characterizing the signal path beyond the antenna and ignoring any gain/loss of the antenna itself for the moment.

A procedure for obtaining the calibration of the receive signal path is as follows:

• Input a known signal level at the input of the receive signal path.  This signal would be input at the antenna port - or via the "coupling" port of the directional coupler.  This signal should be input at a frequency in the band of interest as calibration values should be obtained for every amateur band to be used.  For WSPR operation it is desirable that one generates test signals within the WSPR passband.

• Tune in the signal and note the signal level on the receiver.  A known signal level of at least 20dB (30dB is better) above the receiver noise floor and other signals in the receiver passband is suggested to minimize effects of the noise and signals - but the level should be no higher than necessary to avoid overload/compression of any amplifiers - or the receiver itself.

• If using a directional coupler as suggested, make sure that it is oriented correctly by comparing the signal levels when the generator output is fed directly into the receive signal path and again with signals fed via the coupler:  If its "in/out" is reversed the coupling level will be inaccurate. (Again, for a coupler used for extracting a signal from a coax, the connection will be reversed.)

• In the case of the 20 dB coupler suggested above, the signal levels reaching the receiver will be 20 dB lower due to the coupling loss than that if the signal generator coupled directly and this should be remembered when taking measurements.

• If the levels of the test signal are low (less than 20dB S/N) and cannot be increased, narrowing of the receive passband can minimize the effects of noise - just observe the S-meter while making such adjustments to verify that the test signal remains within the passband.

• Because we are measuring the amplitude of a coherent signal within a finite bandwidth rather than trying to measure the noise power in a specific bandwidth, we must make sure our test signal is strong enough that the power of the background noise is a small percentage of the total.

• While it is necessary to compensate for detection bandwidth when measuring the noise within that bandwidth, one does not do so for coherent signals as all of the power of that signal is (theoretically) all on a single frequency.  It is for this reason that the measurement obtained of the coherent signal should match the actual power of that signal.

• Knowing the signal level being input to the signal path, observe the S-meter reading on the KiwiSDR on a per-band basis.  It is not important that the S-meter read correctly - except for the benefit of users connected to the KiwiSDR and looking at the meter - as the levels reported by wsprdaemon will be adjusted in its calibration file.

• If you are running wsprdaemon and are producing a signal within the WSPR passband, the signal will show up on the graphs after several minutes as a constant level - and this level can be used to verify calibration.

• The level displayed on the graph will be that of the KiwiSDR plus the amount of attenuation/amplification specified in the wsprdaemon.conf file for that band.  Use either raw data or the interactive Grafana data to read the signal levels precisely.

• At the Northern Utah WebSDR, the KiwiSDR S-meter calibration setting has been adjusted so that for bands above the roll-off of the high-pass filter (e.g. 20-10 meters) the S-meter read the same as the input signal level.  At lower HF bands, the S-meter reads low by the amount of attenuation of the filter.  Since there is currently only one S-meter calibration point that applies to all frequencies, there is nothing to be done to correct this.

Any adjustment of the KiwiSDR's S-meter calibration should be done before completing the steps below.  If it is changed, the final per-band calibration values will need to be adjusted accordingly.

• Record, for each band, the amount of RF you are applying at the antenna port (taking into account the coupler, if used) and the S-meter reading on the KiwiSDR.

• The difference between these two readings will be the amount of gain - or loss - in your signal path.  Again, we are not concerned about the absolute gain of the antenna itself at this point.

• If the reading on the KiwiSDR is above that of the absolute input level, signal gain is indicated at that frequency.

• If the reading on the KiwiSDR is below that of the absolute input level, signal loss is indicated at that frequency.

• We have now characterized the gain/loss of the signal path.  If our antenna had unity (0 dB) gain we would use the numbers that we'd just obtained as an offset in the wsprdaemon.conf file.  For example:

• If -50dBm were present at the input of the signal path and the KiwiSDR read -38dBm on 10 meters, we would know that we had 12 dB gain at that frequency.  To compensate we would put -12 db in the entry for 10 meters in the wsprdaemon.conf file.

• If -50dBm were present at the input of the signal path and the KiwiSDR read -52 dBm on 80 meters we would know that we had 2dB attenuation at that frequency.  To compensate we would put 2 dB in the entry for 80 meters in the wsprdaemon.conf file.

• Again, to verify our calculations we can place a carrier of known signal level within the WSPR passband for the band being tested.  If the measurements have been done correctly the levels on the graph should match those of the level of the RF at the input of the signal path.

Compensating for coax cable losses:

The next step would be to include the losses of any coaxial cable in the system before the point in the signal path where we injected our test signal.  This can be directly measured with a wattmeter and dummy load being used to measure the power lost over its length or - presuming a fairly low VSWR - can be estimated using the manufacturer's loss values on a per-band basis.

These loss values would be added to the per-band numbers in the wsprdaemon.conf file.

Compensating for antenna gain:

With simple equipment, calibrating the signal path for an absolute level is pretty easy - but the gain of the receive antenna at a given frequency is more difficult to determine as very few people have access to a known-accurate antenna range - so we will have to make an educated guess.  Because many HF antennas have definite patterns of of lobes and nulls one will have to decide if the peak gain of the pattern will be used, or some sort of averaged gain value.

Passive antennas:

For simple wire antennas, the gain can be estimated with reasonable (+/- a few dB) accuracy.  For example, a half-wave dipole above typical ground will likely be in the area of 1-2 dBi gain peak while the gain of a commercially made vertical or Yagi antenna might be available from the manufacturer's data sheet.

For multi-wavelength wire antennas such as long wires or rhombics, antenna simulation programs are suggested and may be the only reasonable means of estimating gain - but again, one will have to make a decision on what gain value to apply in light of the likely existence of lobes and nulls.

Active HF antennas:

If the feed from a conventional antenna (beam, dipole) includes an amplifier, the steps mentioned above may be used to estimate its gain and one need only add the loss of the feedline and gain of an amplifier to obtain the desired offset for the wsprdaemon.conf file.

Electrically-short active antennas such as the PA0RDT mini-whip pose a particular problem.  They are essentially a capacitive plate (or wire) that couples RF from the "aether" - but the amount of coupling (which correlates with the apparent gain of the antenna) can vary widely, the determining factors including:

• Height above "ground".  In general, the higher above the ground - earth or metal - the more signal such an antenna will intercept.

• Type of "ground".  The ground above which the antenna is mounted could be earth (dirt. soil) which is lossy or it could be a metal roof (much preferred!) to which the feedline's ground is bonded.

• Size of the "antenna".  The traditional PA0RDT antenna uses a PC board plate for coupling - but there are many variations on what size of coupling conductor - wire or plate - is used, all of which can affect the apparent gain.

• Gain response of the amplifier with frequency.  In most - if not all - implementations of an E-field whip are subject to gain drop-off with higher frequency and this is not easy to measure directly and determining a suitable number is almost entirely guesswork.

• Unfortunately, such drop-off conspires with the high RF levels at low-medium HF frequencies.  With a broadband, direct-sampling receiver like the KiwiSDR, overload will surely occur if enough gain is added to the system to attain the natural noise floor at the higher HF bands (e.g. 10 meters) unless strong compensation (e.g. something like the high-pass filter described above) is used.

However one might obtain the antenna gain, this value would be applied to the wsprdaemon.conf file:  Positive gain would mean a reduction of this number by the dB value while negative gain would mean a commensurate increase in this value.

What to expect:

In theory, the noise floor one sees should generally reflect that predicted in the ITU-R data, reproduced in figure 2, below:

Figure 2:  "Typical" noise floor for various radio environments.  Because the above chart is based on a 500 Hz bandwidth, one would subtract 27dB from its vertical axis to scale to an equivalent 1 Hz noise bandwidth when measuring noise power.
Click on the image for a larger version.

To be sure, the values depicted in Figure 2 are very generalized - and one would be lucky to approach the values of "quiet rural".  In general, these value depicted above would be most representative when the respective frequency band is "dead" (e.g. daytime for the lower bands, nighttime for the higher bands) with a "quiet sun".  While usable only as a general guideline, one should be suspect of their calibration settings if the "quiet band" noise floor values they measure on their system is markedly lower than those of the chart above.

Cross-checking noise level measurements with the receiver:

It is possible to check the noise level measurement provided by wsprdaemon using the receiver in question, but to do this two conditions must be satisfied:

• You must know the receive bandwidth.  For most SDRs this is pretty easy:  The "bandwidth" setting - which is typically at the -3dB or -6dB points - is "fairly close".

• The passband must contain ONLY noise and no other signals.  On the WSPR frequencies this may not be possible, but measuring on a "nearby" frequency (up to a few 10s of kHz away, provided that such measurements aren't affected by narrow band-pass filters that might be present).
• Wider (SSB) bandwidths are fine, but it may be necessary to user narrow bandwidths to avoid signals within the passband.

 The method of making this measurement is approximately thus:

• Note the average signal level reading (in dBm).
• Since we are looking for what amounts to a rolling average over time, the readings of occasional noise bursts must be considered which means that a "fast" AGC should be used and one should not be looking at any "peak" readings.
• Since we are likely doing a "sanity check" simply staring at the signal meter and guesstimating the average.
• It is best to do this during "quiet" conditions on a given band - that is, when it is dead:  Trying it on 80 meters during the night where there may be a lot of summer static is not recommended.

•  Note the receive bandwidth.
• On an SDR, this is likely to be the "bandwidth" figure which is probably taken from either the -3dB or -6dB points - which will be "close enough.

• Calculate the bandwidth noise power ratio.
• Because wsprdaemon's noise power measurements are scaled for 1 Hz detection bandwidth, the reading obtained on any receiver with a wider noise bandwidth than that will be much higher and the reading (in dB) must be offset by that amount.
• As an example:  If you are using a 500 Hz receiver bandwidth (e.g. a CW filter) you need to account for a 500:1 power ratio which is:
• 10 * log10(500) = 26.99 = 27 dB - that is, one would subtract 27 dB from the average level that you observed.
• In other words, to scale to the 1 Hz wsprdaemon bandwidth:
• Your signal level - (10 * log10(RX bandwidth in Hz))

Conclusion:

While it is unlikely that the typical amateur will be able to be absolutely confident of the accuracy of their absolute HF noise floor measurements, the steps above will greatly limit the uncertainty of the system's measurement to the the properties of the antenna itself.

[End]

 This page stolen from ka7oei.blogspot.com