Frequency response of the RX-888 SDR at the high and low ends (Above 30 MHz, below 1.5 MHz)

Figure 1:
The RX-888 Mk2.

For information about improving reception below 1.5 MHz, scroll down to the text surrounding Figure 2.

The RX-888 Mk 2 (hereafter referred to as the  RX-888 or '888) is a versatile device, essentially providing a means by which "all of HF" (0-30 MHz - or even 0-60 MHz) may be sampled and presented to a computer for processing via a multi-gigabit USB3 interface.  As it has no onboard signal processing, this device is practically "future proof" in that as all computations are performed on the host computer and there are no frequency or bandwidth limitations regarding the sort of signals - or how many - may be processed, presuming adequate processing capacity.

Comment:

I've seen at least three different "sub-versions" of the RX-888 Mk2 - each one looking slightly different (different circuit board color and other minor differences) - since this device was released.  It's also very likely that components also vary a bit with different manufacturers so the actual frequency response of units of different "builds" may also change.  Unfortunately, I only "have what I have" and haven't been able to compare possible variations in frequency response with these different "builds" - if any.

The highs and the lows

Like any receiver, it has limits of its frequency response - both at the upper end where the high-pass filter dominates and at the bottom end where the component selection as well as the design itself will limit low-frequency response.

Let's look at the low end first.

The lows

The low end limit to the frequency response (somewhere below 1 MHz) of the '888 has not previously been well defined.  This low frequency response is set by component limitations within the HF signal path, including:

• Coupling capacitors.  DC blocking capacitors in series with the signal path will act as high-pass filters, rolling off the low frequencies.
• The Bias-Tee inductor.  The RX-888 has the ability to supply power via the antenna port to an amplifier.  This inductor has a finite inductance and it, too, will force a high-pass response as well.  This inductor's value was measured as being 10uH (nominal) which presents a reactance of 50 ohms at about 800 kHz.  This is the major contributor to low-frequency roll-off as discussed below.
  
• The coupling transformer.  The RX-888 has a transformer that couples the input of the variable gain amplifier (VGA) from the attenuator.  As with any transformer, this, too, has defined low-frequency response.  This transformer was measured and found to have an inductance of 125uH of its primary (a reactance of 50 ohms at 64 kHz) with the secondary (the side facing the VGA) being about 760 uH.
  

This low-frequency roll-of is not uncommon in broadband receivers:  Most amateur transceivers suffer severe performance degradation at LF and VLF frequencies for the simple reason that the designers presume (correctly!) that very few of the users of that gear would ever be interested in that range - and making this assumption simplifies the design somewhat and reduces cost. 

Using a signal generator with a constant output, the response of the RX-888 (Mk2) was measured, using the signal strength at 1500 kHz as a reference, with and without the Bias-Tee inductor removed:

Frequency (kHz)	Attenuation (db) Unmodified unit	Attenuation (db) Bias-Tee inductor removed
1500 (Reference)	0	0
1250	0.3	0
1000	0.4	-0.1
750	1.0	-0.2
500	2.6	-0.3
475 (630 meters)	2.9	-0.3
400	4.5	-0.2
300	7.4	-0.2
250	10.9	-0.2
200	19.9	-0.2
150	17.9	0
137 (2200 meters)	14.7	0
100 (Loran C)	9.7	0.3
75  (DCF77 approx.)	7.5	0.7
60  (WWVB, JJY)	6.8	1.1
50	6.7	1.9
40 (JJY)	7.4	5.3
30 (Submarine comms)	11.3	7.4
25 (Submarine comms)	12.8	10.5
20 (Submarine comms)	17.5	15.0
15	22.5	22.7
10	30.7	32.8
7.5	36.4	40.0
5	45.5	49.5
2.5	61	60
1	80	87

Table 1:  Attenuation measurements at 1.5 MHz and below using both an unmodified RX-888 and the same one after the bias-Tee inductor was removed.

Comments about the frequency response of the unmodified unit:

As can be seen from the table above, the "stock" RX-888 is flat within about 2 dB or so across the AM broadcast band (520-1700 kHz) but it falls off precipitously between 100 and 300 kHz with a bit of "rebound" in the 40-150 kHz area, likely due a very low "Q" resonance of inductance and capacitance of the aforementioned components (inductors, transformers) in the signal path.

In the "VLF" range (30 kHz and below) the unmodified receiver may be somewhat usable when using an active antenna to overcome losses, but at 20 kHz and below the response drops off like a rock and, as the chart shows, it's pretty much unusable below 5-10 kHz.

The factors above conspire to prevent a flat frequency response at lower frequencies - say, those below 1.5 MHz.  For the table below, my reference amplitude and frequency is 1.5 MHz as it seemed to be more or less representative of the amplitude response above this, in the HF range - and it seemed to be comfortably above that at which the aforementioned high-pass effects of the components were having a significant effect.

Can anything be done to improve LF/VLF response?

Figure 2:
The red arrow points to the location of the
10uH bias-Tee inductor.  As seen in
Table 1, its removal can significantly improve
MF and LF performance.  The yellow
arrow points to the 0.47uF bypass capacitor
that needs to be removed even if the Bias-Tee
jumper (if equipped) were to be removed.
Click on the image for a larger version.

YES, it is possible to modify the RX-888 to improve the low and frequency response by removing the Bias-Tee inductor (see Figure 2) from the HF port and as can be seen from the above data there is a dramatic difference in usable sensitivity at frequencies below 1 MHz - particularly below 400 kHz.

This is the easiest modification as it entails the removal of a single component and the red arrow in Figure 2 shows the location of this inductor.  It may most easily be removed with a hot-air rework tool, but it should be possible to carefully use a solder-wetted iron to heat it and remove with a pair of tweezers (temporarily remove any thermal pad below that portion of the board if it's present) or a very sharp pair of diagonal flush-cut pliers to remove it (perhaps destructively) as well.

If your RX-888 has a "Bias-Tee" Jumper:

Some versions of the RX-888 have a jumper near the location of the red arrow in Figure 2.  While removal of this jumper will prevent DC from being applied to the "HF" antenna port, it does not remove the 0.47uF bypass capacitor.  In other words, removing this jumper does nothing to help low-frequency performance.  If you remove the bias-Tee jumper and don't wish to remove the inductor, you may also remove the 0.47uF bypass capacitor, indicated by the yellow arrow in Figure 2.  (Note that even though Figure 2 shows the version without the jumper, the bypass capacitor is still in the same location.)  If you remove the inductor, you need not remove the capacitor.

There are other ways by which the low frequency response may be improved, including:

• Replacing the coupling transformer.  The transformer used in the RX-888 is likely specified for a low-end frequency response of 1 MHz or so, so it's not surprising that this may be the worst offender (once the bias-tee inductor has been removed) in low-frequency roll-off.  Replacing it with a different unit with larger inductance (a commercial or hand-made unit) would certainly help.  It may also be possible to simply replace the transformer with coupling capacitors (say, 0.1uF) - but this would be at the expense of sensitivity and performance across the entire frequency range, something that might be acceptable if one's primary interest was in the MF/LF/VLF spectrum.  As the inductance of the transformer's primary is known to be about 125uH, we can see that this is likely the main cause of attenuation below 60 kHz.
• Increasing value of coupling capacitors.  The coupling capacitors in series with the signal path are likely not ideal for coupling VLF frequencies.  A value such as 0.1 uF or larger would be suggested.

For VLF use (30 kHz and below) if you have interest in this frequency range you may be better off not trying to use the RX-888 - at least directly.  Some possibilities include:

• Use a VLF up-converter.  Converting the frequencies 0-30 kHz to a higher frequency range will put this spectrum within the useful range of the RX-888 and practically any other modern receiver.  There have been a number of VLF up-converter units for sale in the past, but I don't have a specific recommendation.  If this up-converter is clocked from the same source as the RX-888's clock (e.g. using its onboard 27 MHz oscillator, or both from a common, external clock) then frequency drift could be minimized.
  
• Use a sound card.  A modest computer sound card with a 192 kHz sample rate and a 16 to 24 bit A/D converter is perfectly capable of ingesting frequencies up through at least 80 kHz and down (nearly) to DC.

Having a receiver capable of VLF (3-30 kHz) or ELF (300-3000 Hz) is one thing, but having an antenna system capable of this is a different matter altogether.  There are many available E-field active whips that will work well down into the 10-20 kHz region, but below that frequency you are into the realm of specialized gear - and listening at "audio" radio frequencies in all but the most rural areas devoid of power lines and other forms of civilization can be fraught with frustration and disappointment due to the likely pick-up of mains-related energy and its harmonics.

Here are a few links related to equipment for LF/VLF reception.  Note that I have not necessarily built, bought or used the equipment described below, so your mileage may vary.

• The BBB-4 Natural Radio Receiver.  This is a simple circuit is intended for "Audible" RF frequency ranges of a few hundred Hz to a few 10s of kHz and is suitable for detecting "spherics" like whistlers and chirps.
• Some thoughts on E-Field Whistler Receiver Design.  This article describes design aspects of ELF and VLF receivers
  
• An up-converter for receiving long and very long waves.  This article describes an upconverter that may be built.
  
• AMRAD low-frequency up-converter.  This is another upconverter design for VLF/LF use.  The low-end frequency response is limited by T1 in this design.
  
• Heros VLF-LF-MF up-converter.  This is a commercial LF/VLF upconverter. 
• The AMRAD Active LF Antenna.  This article - from the September, 2001 QST - describes a high-performance antenna capable of reception down to at least 10 kHz.  In lieu of the CP666 transistor, one may also use the Russian KП903 (a.k.a. "KP903") power JFET which may be found on EvilBay and other sites.
  

The effective reception of signals in the LF, VLF and ELF frequency range is highly contingent on having a "quiet" receive site, largely free of local noise sources and also on scrupulous attention to detail when it comes to decoupling the feedline (going to the "noisy" chassis of the receiver) from the antenna to prevent unwanted signals from being conveyed - but that's a topic of its own!

See the article A (semi)-typical suburban E-field whip receive system for the 630 and 2200 meter amateur bands - link. for a few details on how this might be done.

Real-world observations

At the Northern Utah WebSDR - where there are, at the time of writing, full-time WSPR receivers - it so-happens that there are currently some KiwiSDR and RX-888 based receivers sharing the exact, same signal path.  The KiwiSDR - which is capacitively coupled (e.g. you can hear "tinny" audio from the receiver tuned to 0 Hz and you apply the source to the antenna connector) has quite good response well into the VLF range.

Compared to the RX-888, the KiwiSDR performs noticeably better on the 2200 meter amateur band (137 kHz) in decoding WSPR and FST4W signals in which the '888 is about 15dB down.  As the '888 based system can't hear the 2200 meter signals as well, this indicates that signal levels feeding the '888 are a bit too low for it to "hear" the noise floor of the antenna system - but it also indicates that, perhaps, a few dB of boost in the signal path may remedy this:  This RX-888 has NOT had its bias-Tee inductor removed - but that's on the "to do" list:  After the bias-Tee inductor is removed I expect that it will perform comparably to the KiwiSDR at 2200 meters and I'll update this web page after having done so.

As the "LF/VLF" antenna system at the Northern Utah WebSDR is separate from that of the HF signal path - being combined in a special filter/amplifier module - boosting only the LF/VLF path would be the most beneficial as it wouldn't compromise HF reception by potentially overloading the A/D converter as would boosting everything.

The Highs

The RX-888's specifications state that it contains a "60 MHz" low-pass filter - but the precise nature of its response is not noted.

Comment about sample rates and aliasing - and the need for additional low-pass filtering

The use of the 60 MHz low-pass filter implies that the designers intended an A/D converter sample rate of more than twice that frequency - and since the RX-888 will happily sample at more than 130 MHz, this fits the need.  Many users do not operate their RX-888 at 130 MHz, however, as their interest does not extend beyond HF and operate it, instead, at around 65 MHz to reduce CPU and power loading.

A bit of warning here:  With a 65 MHz sample rate, the '888 will happily respond to signals above the Nyquist frequency (half of the sample rate, or 32.5 MHz) and these signals - spectrally "inverted" - will naturally appear at lower and lower frequencies as the original source signal's frequency increases.  Since the '888s low-pass filter is set at around 60 MHz, it will do nothing to prevent this:  The far right column of Table 2, below, shows the aliases of the test frequencies.

What this means is that users of the RX-888 using it at a sample rate lower than 130 MHz should be using an outboard low-pass filter.  With a sample rate of 65 MHz, a good-quality 30 MHz Low-Pass filter is strongly recommended and will suppress aliased signals that would otherwise appear above Nyquist.  Such filters may be found online via the usual retailers, but do not overlook an old 30 MHz transmit-type low-pass filter of the sort used to prevent interference to analog TV by an HF transmitter - often found at amateur radio swap meets or on EvilBay for cheap.

Even if you do run the RX-888 at 130 MHz sample rate, the low-pass filter is rather mediocre (only 30-40dB across most of the band) and you will likely need to add an FM broadcast-band filter to your receive system if there's even a modestly-strong transmitter near you.

 The amplitude response, relative to 30 MHz, is shown below:

Frequency (MHz)	Attenuation (dB)	Alias frequency (MHz) @130 MHz sample rate	Alias frequency (MHz) @65 MHz sample rate
30 (Reference)	0	-	-
40	0.8	-	25
50	3.8	-	15
54	5.5	-	11
60	8.5	-	5
64	10.5	-	1
70	14.1	60	5  (double alias)
75	17.7	55	10  (double alias)
80	21.7	50	15  (double alias)
85	27.1	45	20  (double alias)
90	32.5	40	25  (double alias)
95	37.8	35	30  (double alias)
100	42.8	30	30  (triple alias)
105	48.0	25	25  (triple alias)
110	52.9	20	20  (triple alias)

 Table 2:  Sensitivity response of the RX-888 relative to 30 MHz

Table 2 shows the amplitude response of the RX-888 (Mk2) relative to 30 MHz.  The third and fourth column show the resulting aliased frequencies at sample rates of 130 and 65 MHz, respectively.

"Could I intentionally use aliases to receive higher frequencies than my sample rate would allow?"   After reading this, you might ask yourself "If I operate at a sample rate of 65 MHz, could I intentionally do this to receive spectrally-inverted 6 meter signals between 15 and 11 MHz?"   The answer is yes, you could - and as the chart above shows, they would be only 3.8-5.5dB down from the "real" signals across that same 15-11 MHz range.  Intentionally allowing aliases to occur is often done to allow the detection of signals well above the sample rate.  The caveat here is that one would want to sharply filter the source of the "above Nyquist" frequencies to limit them to the band of interest as well as prevent noise on the aliased frequency (15-11 MHz in this example) by filtering those frequencies as well.   Doing this works just fine as long as proper filtering is done to keep out the "unwanted" signals (at the higher and lower frequencies) along with appropriate amplification make up for losses.   In the example above, the lower part of 6 meters would appear just above the 20 meter band - but if one adjust the sample rate, the alias could be moved farther away from 20 meters and, with proper filtering, one could receive both 6 and 20 meters on the same receiver hardware.

What the above table above also shows is that the 60 MHz low-pass filter isn't very good:  By the time you get to the bottom of the FM broadcast band (88 MHz) we know that the attenuation is only around 32 dB.  Here in North America it's common for an FM broadcast station to have many 10s of kilowatts of ERP which means that if you live anywhere near such a station - even if you are using an antenna that wasn't designed to receive FM broadcast frequencies - you may experience some interference around the alias frequencies noted in Table 2.

No matter the sample rate at which you operate your RX-888, it's recommended that you carefully check for aliased responses of FM transmitters.  If you find them - and even if you don't - I'd recommend a separate FM broadcast band blocking filter be installed to quash ingress from strong signals:  Without it you'll probably get some leakage of moderate-to-strong signals in the 22-42 MHz range (frequency-inverted) if you are running at a 130 MHz sample rate or in the 23-32 MHz range if you are running at a sample rate of 65 MHz.

Figure 2 also demonstrates why - if you operate the '888 with a sample rate of 65 MHz - you should really be using a good 30 MHz low-pass filter with it:  Any signals above 30 MHz - including noise - will be attenuated only to the extent shown in the table and will interfere with the desired 0-30 MHz signals.

* * * * *

Other RX-888 related posts at this site:

• Measuring signal dynamics of the RX-888 -  This page discusses the gain distribution of the RX-888, its apparent sensitivity and steps that one should take to maximize performance when used for simultaneous "all of HF" reception.
• Improving the thermal management of the RX-888 (Mk2) - The internal power dissipation of the RX-888 exceeds its ability to get rid of the heat that it produces, reducing reliability - particularly in environments with elevated temperature.  This page discusses what to do to remedy this.
• Using and external clock with the RX-888 (Mk2) - Although the RX-888's TCXO is pretty good, you may wish to use an external reference to provide very high frequency accuracy and stability - and this page gives advice and warnings about doing so.
• Repairing a dead RX-888 (no A/D converter clocking) - While external clocking of the RX-888 (Mk 2) is desirable, it must be done with a bit of care to protect the circuitry involved.  If you do manage to damage your '888, this page may be helpful in its repair.
  

 

This page stolen from ka7oei.blogspot.com

[END]

Using the jt9 executable to receive FST4W signals

Note: 

Since originally posted, WSJT-X v2.3.0-rc2 was released, adding a feature to the "JT9" executable that simplifies this process (the "-F" parameter) as described below.  Note that most of this post was originally written soon after "rc1" had become available.

* * *

As a heavy user of K1JT's WSPR and operating on the 2200 and 630 meter bands, I have noted with interest the introduction of the "FST4W" mode in the recent (v2.3.0-rc1) wsjt-x release.  Operating using the same detection bandwidth as WSPR (when FST4W is operated in the 120 second mode) it offers a theoretical 1.4dB improvement in detection sensitivity.

Being involved with wsprdaemon (link to that project here ) - an open-source project that automates and optimizes reception of WSPR signals on all bands, particularly if multiple receivers/antennas are used - we have been watching this development with interest, particularly since FST4W has the likelihood of supplanting conventional WSPR operation, especially on the lowest amateur bands (2200, 630 and possibly 160 meters) where minimal of Doppler shift is expected.

Internally, WSJT-X  uses the subordinate wsprd program as the decoding (and encoding) engine.  As a stand-alone program, the wsprd executable code may be invoked with a command line to decode signals contained within a .wav file that was captured during the standard two minute interval - aligned with even UTC minutes - and produce a text file containing the decoded signals.

Why use the executable rather than the entire wsjt-x suite?  The fact is that the use of the wsjt-x suite does not lend itself easily to script-driven, bare-minimum, lightweight implementations where further processing of the decoded data (to remove duplicate decodes from multiple receivers, antennas and to use this same data for further analysis of signal/noise) is desired.

The "jt9" executable:

After a bit of digging about, it was "discovered" that FST4W - being an offshoot of the JT9 protocol - was handled not by the wsprd executable, but the jt9 executable.  Simply executing this program with no arguments will yield a list of command-line arguments which, on the face of it, made it appear that updating the wsprdaemon to include the decoding of FST4W signals would be a relatively simple matter.

Except that it didn't work.

Initial testing with strong, off-air FST4W signals that was known to be decodable (because farther-flung stations were able to decode the very same transmissions) yielded no results when the .wav file was applied to the jt9 program - but automatic execution over many hours yielded the occasional off-air decode.  Confused by this, I sought help on the WSJT-X groups.io forum.  Fortunately, Joe Taylor and several of the developers offered a clue:  The "-f" parameter of the jt9 executable, described minimally as "Receive Frequency Offset".

Apparently, the default center frequency of the jt9 executable - at least when in FST4W mode (and maybe others) is 1500 Hz - a fact implied when one gets the display of command-line arguments.  What is not so clear - and only alluded to in the available documentation - is that the apparent bandwidth of the decoding, at least in the 120 second mode, is on the order of 40 Hz (+/- 20 Hz).  Addendum:  This issue was fixed with the "-F" parameter - see below.

At a quick glance through the source code (file "jt9.f90"), this bandwidth setting appears to be hard-coded into a shared variable (apparently accessible by other programs in the WSJT-X suite) called "ntol" (likely a number referring to the "frequency tolerance" setting in the GUI) that is not available via the jt9 command line - at least, not without modification of the source code.  (The possibility of directly accessing these shared variables exists - but this would be platform-specific, a bit messy and somewhat dangerous!)

Unfortunately, this fixed +/-20Hz bandwidth does not appear to be compatible with the way that the FST4W mode has (already!) found use on 2200 and 630 meters where it is used along-side the WSPR mode in the 200 Hz subbands.

A hell of a kludge:

Update - kludge no longer needed:

As of version 2.3.0-rc2 it appears that a new parameter "-F" was added to allow something other than a +/-20Hz bandwidth (referred to as "tolerance") to be used, likely eliminating the need for multiple decodes, below.  A possible command-line for this would be:

jt9 -W -p 120 -f 1500 -F 200 <wav file to be processed> 

With the center frequency (-f) being the center of the passband (1500 Hz) and the "-F" parameter referred to as"tolerance" (e.g. detection bandwidth) being 200 Hz. 

Initial testing indicates that the -F parameter does what it's supposed to do and the kludge below is now longer required.

This fact implies that in order to use something other than the GUI version of the wsjt-x software, a work-around must be invoked.  The following is a bare-minimum example of how one might do this via the command line:

jt9 -W -p 120 -f 1420 <wav file to be processed> 

jt9 -W -p 120 -f 1460 <wav file to be processed>

jt9 -W -p 120 -f 1500 <wav file to be processed>

jt9 -W -p 120 -f 1540 <wav file to be processed>

jt9 -W -p 120 -f 1580 <wav file to be processed>

(One might include the -H, -L and -d parameters in actual practice.)

In other words, in order to cover the entire 200 Hz WSPR subband, the JT9 executable (v2.3.0-rc1) must be executed - processing the same .wav file - at least five times:  The results of the decoding will, in each case, be found in the file "decoded.txt".  If one wishes to implement an equivalent of the -w parameter of the wsprd executable (e.g. +/- 150 Hz "wideband" mode), you will need even more invocations than above.

The result from the above mess will be five different decoding results, each of which must be saved (e.g. renamed) between subsequent executions to prevent overwriting by the previous instance.  After this, the five results would be concatenated to yield a single file - but there is a catch:  It is likely - particularly if the signal is strong - that the same signal will be decoded more than once.  Apparently, the "+/- 20Hz" limit isn't the result of a "brick-wall" filter:  Signals beyond this frequency range may be decoded, but the reported S/N values will likely be reduced as distance of the received signal from the specified center frequency increases.  In short, this means that the results of the concatenated version of the "decoded" file(s) must be sorted and all but the single, "strongest" decode (e.g. best SNR) for each station must be discarded.

Comment:

It would appear that just five iterations to cover the 200 Hz bandwidth is not enough:  I received correspondence from a reader of this blog that observed that a frequency variation of less than 10 Hz from that defined by the "-f" parameter can affect the S/N reading by about 1 dB.  Make of that what you will!

* * * * * * * * *

If one wishes to integrate the FST4W decodes into the existing WSPR captures for processing, yet another step must be undertaken:  "Fixing" the formatting.  Not surprisingly, the output in the "decoded.txt" is not formatted the same as the results of the decoding from the wsprd executable meaning that one will need to do a few things, after the fact, to "fix" them - particularly if you wish to forward them to wsprnet.org, including:

• Supply the date.  The "decoded.txt" includes the time - but not the date.  Because date of the .wav file may not be the same as the system date (e.g. later processing of the .wav files - or the interval being processed occurred just before the new day) - one must use the actual date of the recording.  The obvious place to obtain this is from the name of the .wav file being processed.
  

• Frequency offset.  The information that one might send to wsprnet.org must include the carrier frequency of the received signal, but the output in the "decoded" file has only the audio frequency:  One must obtain the LO frequency of the receiver being used from "somewhere else" and calculate this on the fly.
  

• Supply missing information.  The "decoded.txt" file does not have all of the same information fields that one might supply when uploading WSPR spots, so this information must be added as necessary.
  

• Arrange the fields in the proper order.  Once the needed information is applied, one will probably want to use "awk" or similar to produce the same order as the wsprd data - assuming this wasn't already done in the process.
  

* * *

There are two outputs from the jt9 executable - one directly from the program itself to the standard console and that output to the file "decoded.txt" - and the latter is the most useful. 

Console output:

0416 -24  0.7 1515 `  KA7OEI DN40 17                                 
<DecodeFinished>   0   1

The fields are:  <time UTC> <SNR in dB> <DT?> <Audio frequency in Hz> <always "`"> <Callsign received> <Grid of received station> <Reported TX power in dBm>

From the "decoded.txt" file:

0416   0  -24   0.7   1515.   0   KA7OEI DN40 17                        FST4

 The fields are:  <time UTC> <unknown - possibly drift in Hz> <SNR in dB> <DT?> <Audio frequency in Hz> <unknown> <Callsign received> <Grid of received station> <Reported TX power in dBm> <Always "FST4">

* * *

There you have it:  The germ of what would be needed if one wishes to supplement the existing WSPR decodes with the newer FST4W mode using just the bare executables.  If one wishes to decode other than the 120 second FST4W mode, things get even more complicated!

Sample audio file:

An audio file containing both FST4W-120 and WSPR transmissions may be found HERE - right-click to download.  This file contains an FST4W-120 transmission by KA7OEI from about 116km distant and (at least) two WSPR transmissions.

* * * 

 

Note:  It appears that the "-F" parameter, above, modifies the default ntol setting as described above.

P.S.:  While it would be pretty trivial tweak the code to allow modification of the ntol variable via command line, this would complicate the ongoing maintenance of the wsprdaemon code.  We can only hope that the current authors see fit to include a means by which the entire wspr subband can be monitored with a single invocation of the jt9 executable.

 

This page stolen from ka7oei.blogspot.com

[End]

Homebrew construction of 2 and 4 port splitters/combiners for the LF-MF-HF(30 kHz-30 MHz) frequency range.

Note:

This is a follow-up of a previous article, "Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range" - link.

Comment:

All of the devices described here could also be used to combine signals from multiple sources.  Unless the signals being combined are "phase coherent" (e.g. from the same signal source) the insertion loss will be the same as that in splitter operation.

"Rolling your own" splitter for LF through HF (<30kHz-30MHz):

Unless you get the Mini-Circuits devices for cheap at a swap meet or via a surplus outlet, their cost may be a bit prohibitive for casual use in the shack.  How about making your own splitter that will work over the 30 kHz-30 MHz range?

Why would one want this?  There are a number of modern Web-Based SDR receivers that cover from (literally!) audio through 30 MHz - and in my case, I have a number of KiwiSDR receivers - link that are connected to an antenna system that is capable of intercepting signals over this range.  If one has several such receivers, it can be a challenge to find a splitter that works well over this range - particularly the low end - as described in the article linked above.

Figure 1:
A two-transformer splitter/combiner.  L1 transforms the impedance at
J1 to half that value and L2 splits the signal itself.  R is twice the system impedance
- 100 ohms in a 50 ohm system.  The tap on L1 is at 0.707 times of the total
number of turns:  A tap at 7 of 10 total turns is "close enough" while C is a high-
frequency compensating capacitor.  L2 is a bifilar-wound transformer with the
two sets of windings connected in series at the common point.
Click on the image for a larger version.

To do this, ferrite - rather than iron-powder - cores would be used, the most common types using mix 31, 43, 61, 73 and 75 - and the most useful of the lot for the low-frequency end are types 73 and 75.  Not having a complete assortment of the ferrite types on hand, I used what I had and the use of a binocular core with mix 43 and a ferrite with mix 75 is discussed here.

Two-transformer splitter/combiner:

A common splitter topology consists of two cores:  One to transform the impedance to half that of the characteristic system impedance and a second to split the signal two ways as depicted in Figure 1.  The inductance of L1 and L2 should be high enough to present a reactance of 3-10 times the system impedance at the lowest frequency.

Figure 2:
Measured insertion loss of a transformer using two BN43-202 cores wound
with 30 AWG wire:  10 turns, tapped at 7 turns for L1 and 10 bifilar turns
for L2.  R = 100 ohms and C = 62pF.
The required number of turns to achieve the desired low-frequency response
increases the stray capacitance, undesirably increasing loss at the
high-frequency end of the HF spectrum.
Click on the image for a larger version.

Figure 2 shows a tested version of this transformer, the details noted in the caption.  According to this diagram the insertion loss is a nominal 3-ish dB from 50 kHz to 1 MHz, dropping down to about 5 dB loss between 20 kHz and 10 MHz and nearly 6dB at 30 MHz.  Capacitor "C" was made variable, adjusted for lowest loss, improving the highest end by a bit over 1dB, and its value measured.

For LF and HF use, this splitter is just "OK" - the loss being an extra 3dB at the high end of the spectrum:  If preceded with amplification, this loss may be tolerable - but note that even the nominal 3dB loss of a 2-way splitter should be of concern at the higher HF bands as signals - and the natural noise floor - can be quite weak and additional loss can drop the receiver's noise floor below that, potentially causing the loss of reception of weaker signals.

Much of the high-frequency loss is due to the inter-winding capacitance.  Experimentally, versions were constructed using wire with PTFE ("Teflon") insulation and comparing it with another with the same number of turns of the 30 AWG enamel and the losses for the PTFE wire version were 1.5-2dB lower - but fewer turns could be passed through the core and low-frequency response suffered.

Figure 3:
In this form, the primary (connected to J1) has 1.414 times as many turns
as each of the two identical secondary windings.  The value of R is half that
of the characteristic system impedance, or 25 ohms for a 50 ohm system:
Parallel 51 ohm resistors were used for a nominal 25.5 ohms.
Typical turns values are 10/7 turns, 14/10 turns and 20/14 turns for the
primary and bifilar secondary, respectively.  For the center-tap, the windings
of the secondary are connected as if they were in series.
Click on the image for a larger version.

If a higher-permeability material (like 73 or 75 mix) were used for the core rather than 43, fewer turns could have been used to maintain the inductance and low-frequency response and it is likely that the high-frequency loss would be reduced.

Single-transformer splitter-combiner:

Another common splitter/combiner is the form depicted in Figure 3, using a single core - and potentially this can reduce loss compared with a device with two cores.

In this system the primary should consists of 1.414 times (e.g. the square root of two) as many turns as each of the secondary windings.

Figure 4:
The insertion loss of the described two-way splitter using 24 AWG wire on
an FT50-75 core:  It is well below 4dB over the range of 10 kHz to 60 MHz.
Click on the image for a larger version.

Both a binocular and toroidal core were tried and better results were obtained with the FT50-75 core than the available BN43-202 binocular core - both because the higher permeability improved the low-end response and larger wire could be used for the toroid:  Capacitance was reduced on the toroid because the turns could be spread out rather than being tightly overlaid as the case of the binocular core, and high-end losses were further-reduced by laying the turns of the secondary next to each other rather than the higher capacitance resulting from the two conductors being twisted as is commonly done with Bifilar windings.

The results of this work are visible in Figure 4.  For this transformer, two parallel secondary "bifilar" windings consisting of 14 turns each were carefully and neatly laid down using 24 AWG enamel wire with 20 turns of 24 AWG over the top.  As can be seen, the results are excellent:  The insertion loss is below 3.6dB from 10 kHz to 60 MHz and the overlaid Smith chart shows the VSWR to be pretty well-behaved, never exceeding 1.5:1 over this range.

Figure 5:
The port-to-port isolation is quite good over the range of 100 kHz to
30 MHz.  The peculiarly-flat isolation limit of the bottom "trough"
of the graph is a result of the value of "R" not being exactly 1/2 of the
impedance value of the system used for testing:  If R is made variable,
higher isolation may be obtained in the middle of the range - but the
difference between that and the fixed resistors used was only an
ohm or two.  In practice, high values of isolation can be obtained only
if the source and load impedances are purely resistive, but since
practical antennas, amplifiers, receivers and filters will not be perfect
sources and loads, such high isolation cannot be achieved in practice.
Click on the image for a larger version.

Additional tests were run to determine the port-to-port isolation of this splitter - the results being visible in Figure 5.  Over the range of 100 kHz through 60 MHz, the isolation exceeds 15dB, exceeding 25dB from abut 100 kHz through 30 MHz.  During testing, the same device had been constructed using smaller 30 AWG wire and the results were worse above 10 MHz (by 2dB at 30 MHz) - likely a result of the skin effect losses of this smaller wire.

A four-way splitter:

I happened to have a need to take signals over a wide frequency range and split it four ways - specifically, to several KiwiSDR receivers, stand-alone web-based receivers capable of reception over the 5kHz-30MHz range - so I decided to construct a splitter using the configuration described above.  To do this, I would need three splitters:  A pair of splitters to feed the four outputs and one more splitter to feed the aforementioned two splitters.  This splitter is depicted schematically in Figure 6:

Figure 6:
For a 50 ohm system, resistors "R" are 25.5 ohms (two 51 ohm resistors
in parallel) and capacitors "C" are 47pF NPO/C0G types used to "flatten"
the response to 30 MHz.
The as-built splitter uses FT50-75 cores wound with 24 AWG, the dual
secondary windings consisting of 14 turns and the primary with 20 turns.
The dual secondaries are laid parallel rather than twisted to minimize stray
capacitance.  The center-tap is connected as if the two secondary windings
were placed in in series.
Click on the image for a larger version

This splitter consists of three of the two-way splitters connected as described:  FT50-75 cores wound with 14 turns, each of two parallel 24 AWG conductors for the secondary overlaid with 20 turns of 24 AWG for the primary.  During testing it was observed that the addition of capacitors "C" slightly reduced (by nearly 1 dB) the insertion loss at 30 MHz at the expense of increased loss (about 2dB) at 60 MHz - but because the target high-end limit was 30 MHz, this was considered to be acceptable.

The end result was an insertion loss (see Figure 7) of less than 7 dB from 20 kHz through 30 MHz, rising to 8 dB and 9.3 dB at 10 kHz and 60 MHz, respectively, being under 6.3dB between 50 kHz and 10 MHz.  In testing port-to-port isolation, the worst case results were those obtained from the same transformer (e.g. T2 or T3) and this value was at least 15dB from 50 kHz to 30 MHz.

Figure 7:
This shows the typical insertion loss of the as-built four-way splitter
depicted schematically in Figure 17.  The insertion loss is less than 7 dB
from at least 20 kHz through 30 MHz with the VSWR being 1.5:1
or less over that same range.
Click on the image for a larger version.

This four-way splitter was built into a small die-case box for mechanical rigidity and electrical shielding.  Inside the box, pieces of plastic tape were affixed to the bottom and the lid to eliminate the possibility of inadvertent shorting of connections to the case:  Details of the mechanical construction may be see in Figure 8.

To reiterate:  It was determined that with the number of turns required to obtain good response into the LF range (e.g. below 30 kHz) that the use of twisted bifilar windings was NOT indicated:  Doing so resulted in excess loss (3-6dB) by the time one got to 30 MHz.  As indicated, the use of thicker insulation (e.g. PTFE versus enamel) reduced this loss somewhat, but using the smallest wire on hand with the only available 75 mix toroid, too few turns could be wound to afford the needed inductance for the desired low frequency respons:  The best-results with the materials on-hand were obtained by simply laying the "bifilar" windings parallel to each other.  In this case, 24 AWG enamel wire was used, a compromise between lower skin-effect losses and the ability to fit the required number of turns on the FT-50 core.

Comment:  There are other splitter topologies available that have their own sets of advantages and disadvanges.  While some of these may be discussed in (a) future article(s), they are beyond the scope of this article - which is the construction of a very simple, straightforward device that is suitable for the task at hand.

Conclusion:

If one needs a very wide-range splitter for broadband receivers that cover from LF through HF - such as some modern "Direct Sampling" SDRs (e.g. the KiwiSDR) there are some commercially-available devices that may be found that will work well - if you can find them surplus, or are willing to pay for them.  If you are willing, a perfectly suitable device may be constructed inexpensive using a minimal complement of components.

Figure 8: 
The as-built 4-way splitter. capable of useful operation from below 20 kHz to above 30 MHz.
As described in the text, the cores are FT50-75 wound with 24 AWG wire, wired "dead bug"
inside a small die-cast aluminum box.  The resistors "R" and compensating capacitors "C"
may be easily seen.  The bottom of the box and the lid (not visible) are insulated with a piece
of plastic tape - this this case, 1" (25mm) wide PET (Polyester) tape.  If I'd had some on hand,
I would have wound the transformers on slightly larger toroids to spread out the windings a bit.
Click on the image for a larger version.

* * *

This is a follow-up of a previous article, "Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range" - link.



Stolen from ka7oei.blogspot.com


[End]

Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range

Figure 1:
The collection of devices to be tested - plus a few 50 ohm terminators.
Click on the image for a larger version.

Note: 

There is a follow-up article, "Homebrew Construction of 2 and 4 port splitter-combiners for the LF/MF/HF frequency range" link.


Rummaging through a box of RF stuff I ran across several multi-port devices made by Mini-
Circuits Labs that I'd picked up over the years - typically at amateur radio swap meets.

The "official" specs of these devices are easy to find (at least for the newer "plus" versions) but what if, like me, one was interested in using them at frequencies below their official design specs - such as the lower amateur bands, including 80, 160, 630 and 2200 meters?  How much "extra" design margin was built into these device?

Wielding my DG8SAQ Vector Network Analyzer, I decided to find out.  For these measurements I limited the range to between 10 kHz and 60 MHz.  I also built several homebrew versions to see if I could, for little cost, come up with suitable versions of my own - and these are described in the follow-up article linked at the top of this page.

Comment:

All of the devices described here could also be used to combine signals from multiple sources.  Unless the signals being combined are "phase coherent" (e.g. from the same signal source) the insertion loss will be the same as that in splitter operation.  They will be referred to only as "splitters" in this article to minimize clutter.

Figure 2:
 Insertion loss of the ZFSC-4-3 from 10 kHz to 60 MHz.
Even though the "official" low-frequency specification is 10 MHz,
it should be quite usable on 160 meters (down to at least 1.8 MHz).
Click on the image for a larger version.

ZFSC-4-3 four-way splitter:

This device, equipped with BNC connectors, has an "official" frequency range of 10-300 MHz (the currently-offered "plus" version has the same ratings), splits the signal 4 ways with a theoretical insertion loss of 6 dB, but practically speaking, the actual loss is rated as being closer to 6.4 dB over the lower end of the design range.  Although this device - and others below - are billed as a splitters, they may be used to combine disparate signals from multiple sources onto a single line with the same amount of insertion loss.

Figure 2 shows the measured insertion loss over the range of 10 kHz to 60 MHz.

Figure 3:
 Isolation of the ZFSC-4-3 from 10 kHz to 60 MHz between ports 1 and 2.
Click on the image for a larger version.

This shows us that down to about 1.8 MHz (160 meters) that the insertion loss (blue trace) is only slightly (0.15dB) higher than the rated specs - and the Smith chart (red trace) is also reasonably well-behaved.  The next marker to the left (#3) is placed at 100 kHz and we see that the insertion loss is closer to 9 dB and that the impedance has dropped to around 12 ohms - getting worse at 20 and 10 kHz where the losses are 20dB or more and the measured impedance is only a few ohms.

What this tells us is that this device is likely to be useful down to about 500 kHz, below which point the insertion loss and impedance mismatch start to become significant - likely due to the fact that the intrinsic impedance of the ferrite devices within the splitter has dropped too low at these frequencies to remain "transparent".

Figure 4:
 Isolation of the ZFSC-4-3 from 10 kHz to 60 MHz between ports 1 and 3.
Click on the image for a larger version.

Figure 3 shows the port-to-port isolation between ports 1 and 2 and the scene is similar:  The insertion loss curve is pretty flat to about 500 kHz where it starts to vary and much below 100 kHz, the isolation seems to increase, but this correlates to the insertion losses.

Figure 4 shows the port-to-port isolation between ports 1 and 3.  This is different from that in Figure 3 because a 4-way splitter actually consists of three two-way splitters:  One to split two ways, and this path is then split two more ways with ports 1 and 2 on one splitter and 3 and 4 on another - and cross-coupling to the "other" splitter is apparently not as good at frequencies below the design.

It is worth noting that all of the above measurements are contingent on all ports "seeing" a 50 ohm source and load - either from the instrument itself doing the port-to-port measurements, or by terminating the "unused" ports (e.g. those not involved in the measurements) with known-good 50 ohm loads.  It is likely that real-world devices (antennas, receivers, amplifiers, filters) connected to any splitter will not have as good a return loss (effectively, VSWR) as a load and this will affect the isolation and apparent insertion loss.

Despite what the "official" ratings say, this device would be suitable down to at least 160 meters (1.8-2.0) MHz and likely usable through the entire AM broadcast band and, possibly, the 630 meter band.

Figure 5:

Insertion loss of the ZFSC-4-1 splitter between 10 kHz and 60 MHz.
Click on the image for a larger version.

ZFSC-4-1-BNC+ four-way splitter:

This device, also equipped with BNC connectors, has an "official" frequency range of 1-1000 MHz, splits the signal 4 ways with a theoretical insertion loss of 6 dB, but practically speaking, the actual loss is rated as being closer to 6.4 dB over the lower end of the range.

Figure 6:
 Typical port-to-port isolation of the ZFSC-4-3 from 10 kHz to 60 MHz.
Click on the image for a larger version.

As Figure 5 shows, this device does a much better job at the low end of things than the ZFSC-4-3:  At 100 kHz, the insertion loss is only slightly (0.2dB) higher than at 1.8 MHz and Marker #3 at this frequency on the Smith chart shows a reasonable (approx. 1.5:1) VSWR.  By the time one gets to 20 and 10 kHz, the VSWR and insertion loss have risen - but not as bad as that of the ZFSC-4-3.

Figure 6 shows the typical port-to-port isolation (ports 1 and 2 in this case) showing that down around 100 kHz, the isolation has dropped to about 20dB - still reasonable, and comparable to the isolation to be expected at the high end (published specs, near 1 GHz) of the design frequency range.

Clearly, if one has the amateur 2200 and 630 meter bands in mind - or one is splitting signals above about 100 kHz to feed several receivers - this is a much better choice than the ZFSC-4-3.
 

Figure 7:
 Insertion loss of the ZFSC-2-1+ two-way splitter between 10 kHz
and 60 MHz.
Click on the image for a larger version.

ZFSC-2-1+ two-way splitter:

This device, equipped with BNC connectors, has an "official" frequency range of 5-500 MHz (the "plus" version has the same ratings), splits the signal 2 ways with a theoretical insertion loss of 3 dB, but practically speaking, the actual loss is rated as being closer to 3.3 dB over the lower end of the range.

Figure 8:
Port-to-port isolation of the ZFSC-2-1+ two-way splitter between 10 kHz
and 60 MHz.
Click on the image for a larger version.

Figure 7 shows the measured insertion loss and surprisingly, it looks quite good down to 100 kHz - probably due, in part, to the fact that unlike the four-way splitters, there is likely only a single ferrite device contained within to incur losses at the low end where it "runs out" of inductance on the transformer.  Down at 20 kHz the loss has gone up by about 2dB and the impedance is in the area of 20 ohms, but this device may still be fairly usable in some applications.

Figure 8 shows the port-to-port isolation and this remains above 20dB down to about 250 kHz, quickly dropping to about 14dB at 100 kHz.

What this tells us is that this device is still likely to be usable down to 100 kHz if one is able to tolerate a couple of extra dB of loss and only mediocre isolation.

Figure 9:
Insertion loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.  Because the "Couple" port is pulling a slight amount of
energy from the through line, a small amount of insertion loss
is to be expected.
Click on the image for a larger version.

ZFDC-20-3 20dB directional coupler:

This device is not a splitter, but rather a device designed to directionally "siphon" a small amount of signal from the "through" line - but do this only in one direction.  This device is typically used to sample (with 20dB of attenuation) a signal on a given line, or if turned around to couple in the opposite direction it can insert a signal on this same line.  A common application of this device is to measure return loss (or VSWRm using a pair of these devices), allow non-intrusive monitoring of signals on a cable and it can be used to insert a signal on that same line - say for receiver sensitivity testing - on a cable that cannot be interrupted.  Unlike a splitter, connecting/disconnecting a device on the "Couple" port will have a very small effect on the through-signal.

Figure 10:
Forward coupling loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.
Click on the image for a larger version.

The "official" specs of the ZFDC-20-3 indicate a frequency range of 200 kHz to 250 MHz, but one can see in Figure 9 that the insertion loss is well below 1 dB down to around 20 kHz - although the VSWR at this frequency climbs to nearly 3:1:  At 50 kHz, the insertion loss is still only about 0.25dB and the VSWR is about 1.5:1 - still within the usable range for applications that can tolerate a small amount of degradation.

On the sample port we can see on Figure 10 that the coupling is ruler-flat down to at least 100 kHz and still staying within 1dB of the nominal value down to 10 kHz - but one should keep in mind the fact that the insertion loss and the varying impedance will likely affect the through-line's signals below around 50 kHz.

Figure 11:
Reverse coupling loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.  Because the coupling is 20dB in the forward direction,
the attenuation values depicted in the above graph should be reduced
by that amount.  The "bump" at the extreme low end is an artifact
of the configuration of the test instrument.
Click on the image for a larger version.

Figure 11 shows the "reverse" coupling loss (e.g. "directionality").  Ideally, no signal should be detectable when the "load" is a perfect, non-reflective 50 ohms but due to imperfections in the load, device, cabling and measurement will reduce this.

This shows that the absolute directionality+coupling exceeds about 60dB (about 40dB of directivity compared to the "forward" coupling) at all frequencies below 60 MHz down to about 20 kHz:  Values below about 70dB (the "floor" between 20 kHz and 10 MHz) are representative of the limits of the test instrument and its configuration so they may actually be greater than this.  Below about 15 kHz, the "bump" is mostly due to measurement artifacts - but this still indicates that the relative directionality is at least 30dB.

These measurements indicate that this device is usable down to 50 kHz with only minor degradation, and would probably work down to 25 kHz in applications where one can tolerate a bit of extra insertion and return loss.

Final comment about the Mini-Circuits devices:

In reviewing the above tests, it would appear that these Mini-Circuits four-way splitters and the directional coupler are generally useful down to about 1/10th of their "official" low frequency rating and that down to 1/5th of their low-frequency rating, they more or less meet their "official" specs.


Follow-up article:

I have built several homebrew versions of 2 and 4 way splitters to see if I could, for little cost, come up with suitable versions of my own that will work from below the LF range through HF - and these are described in this article:  "Homebrew Construction of 2 and 4 port splitter-combiners for the LF/MF/HF frequency range" link.

* * *

Stolen from ka7oei.blogspot.com


[End]

A transmit converter (and amplifier) for 630 and 2200 meters

There is a dearth of commercial equipment "out there" designed to allow operation on the new (to U.S. Amateurs) 630 and 2200 meter bands.

Figure 1:
Complete (except for antenna and matching network) 630
and 2200 meter transmit station.  The IF radio (a Yaesu FT-817)
is tuned to the frequency for 2200 meter WSPR operation -
136.0 kHz + WSPR audio offset.
Click on the image for a larger version.

There have been some attempts to use commercial amateur transceivers to produce transmit RF at these frequencies but due to the 630 meter band being less than 1/3rd the frequency of 160 meters, the filtering and circuitry within simply isn't designed for this - and that's if you can even get around the radio's inhibition to transmit outside its designed frequency range!

Here are a few different radios and their attempts to be used at these frequencies:

• Flex 6000 series:  Several amateurs successfully use radios in the Flex 6000 series for receive and transmit on the 630 and 2200 meter bands - but with a caveat:  A low level transmit signal on these bands is available only from the transverter port and an external power amplifier and filtering is required.  I don't know to what degree earlier Flex radios may have supported 630 and/or 2200 meter operation.
  

• Kenwood TS-590S(G):  From the "Drive" output on the rear panel is available, at a level between -10 and 0 dBm - much like the Flex 6000.  This output is usable from (reportedly) 2200 meters and up:  Several amateurs use this radio - receiver, too - on 630 meters.  (The TS-590SG works this way - not 100% sure about the TS-590S.)
  

• Icom IC-7300:  Several have reported that the IC-7300 will seem to "go" down to 630 meters, but while this radio may coaxed to "tune" down here - and the wattmeter may even show output power - analysis has shown that not only is output at this frequency loaded with harmonics, but that attempted operation at this frequency may well stress other components (e.g. things get warm!)  Receive performance is reportedly rather poor, requiring strong band-pass filtering for 630 meters and (possibly) some receive signal amplification.  At the time of writing, I am unaware of anyone who has successfully used this radio for transmitting on the 630 or 2200 meters bands.  It has been reported that a diplexer is suitable for providing both band-pass filtering and appropriate RF termination to allow it to provide low-level (no more than a few watts) output on 630 meters.  This diplexer is mentioned on KB8U's blog - link.
  

• Drake TR-7/A:  The TR-7 - a solid-state all band HF transceiver from the late 1970-early 1980s - has an "LF Input" pin on a rear panel connector which allows, with some external circuitry (amplifier, filtering) reception down to almost DC.  A slight modification of the radio can permit a transmit signal to be produced on this pin (in the sub-milliwatt range) down to a few 10s of kHz with appropriate amplification and filtering being required to make this useful.  Because this radio natively uses an analog VFO, a stable, outboard digital VFO is required to obtain the stability necessary for the narrow-band digital modes often used on these band. (I own a TR-7A and have done this in the past.)

• Icom IC-735:  Some have reported the ability to "transmit" at 630 meters, but like the IC-7300 there is very little output at the desired frequency and there is the possibility of stressing components in the attempt.  Reception requires strong filtering and some amplification.

• Elecraft K3/K3S:  The K3S can reportedly produce low power (approx. 1 mW) at 630 meters on its transverter port.  It would appear that doing similar for 2200 meters is not possible and that most K3S owners that operate 630/2200 meters seem to use transverters, anyway.  I do not know about the receive performance on these bands.  For more information about using the K3 at 630 meters, read this app note from Elecraft.

In short:

• Even if the radio can be made to go into transmit mode at a frequency below 500 kHz, it is likely that it is producing very little power at these frequencies and is stressing transmit components:  The radios' power amplifiers simply cannot be used as-is.  In many radios, if they allow transmitting at all, the desired 630/2200 meter signal may be among harmonics and spurious signals, requiring good filtering if it is to be at all usable.

• Many receivers are somewhat "deaf" at these frequencies - particularly at 2200 meters.  Even if they are not, strong band-pass filtering for the band of interest is usually warranted along with appropriate amplification, particularly if there are local AM (mediumwave) broadcast stations that can overload the front end.

• If you have a radio that can tune below 500 kHz you may find that it is badly overloaded by local AM/Mediumwave broadcast signals.  A practical 500 kHz low-pass filter is described here:  Low-Pass filter for LF/MF (2200 meter and 630 meter) reception.  This filter works well for preventing AM broadcast station overload to a receiver and it may also be used for low-pass filtering in low-power (<1 watt) transmit circuits.

Off-the-shelf 630 and 2200 meter converters:

What all of the above means is that some sort of transmit converter may be warranted.  There are a number of transmit-capable converters out there designed for operation on one or both of these bands.

Here are a number of kits or pre-built units that are available for the 630 and/or 2200 meter bands.  I have no experience with any of these devices and cannot offer advice as to how well they might work - I will leave it up to you to do that! 

• Minikits - An Australian company that makes several kits, including a 630 meter transverter kit (link).
• Monitor Sensors - This company offers both a 630 meter (link) and 2200 meter (link) transverter.

It is likely that there are more than the above transverters available and I will update this list if supplied information.  Again, I have not used any of the above and can make no specific recommendation.

A practical transmit converter:

As the name implies, a transmit converter takes another frequency - such as that produced by a conventional HF transceiver - and converts it to another frequency.  In my case I use an FT-817 - a low-power (5 watt) all-mode, all-band transceiver that is a favorite for VHF, UHF and microwave enthusiasts that use transverters.  Because of its small size, feature set and already-low output power, it is a natural to be used in this application.

If you don't have an FT-817 (or FT-818) on hand that can be modified to transmit "everywhere" you may have an HF transceiver that has a transverter output port that can produce a few milliwatts.  If your transceiver doesn't have a low-power transverter output you will either need to modify the transceiver to have one or use a 100 watt dummy load in conjunction with a 20 dB tap (or a 20 dB pad capable of handling 100 watts) to drop the power to a "safe" level.

I constructed my transverter from parts that were on-hand, but these parts are readily available:  A schematic diagram of the circuit may be seen below.

Figure 2:
Diagram of the transmit converter.  This circuit uses a 10 MHz local oscillator that is divided-by-two to yield a 5 MHz IF which can yield better overall frequency stability.
The circuit in the upper-right corner is used to convert a lower-output (3.3 volt) TCXO or OCXO to TTL level - see text.
The above circuit works well for receive conversion as well if the 20 dB attenuator is removed or relay-switched and the nominal 7 dB loss of the mixer is taken into account.  
Click on the image for a larger version

Circuit description:

Local oscillator:

The local oscillator frequency chosen for this converter is 5 MHz - a frequency band available on many HF transceivers that have been "opened up" to allow operation on the 60 meter amateur frequencies.  The choice of this frequency was also influenced by the convenience being able to use a readily-available 10 MHz oscillator, which could be a 10 MHz TCXO, an "ovenized" oscillator or 10 MHz from an available in-shack reference such as a GPSDO.

The use of a "low-ish" IF frequency like 5 MHz can also enhance the stability:  With many modern transceivers, a single, internal reference sets its frequency stability and accuracy and the lower the frequency, the greater the stability.  I took advantage of the availability of an inexpensive (<$20) EvilBay TCXO for my FT-817 to give it an overall stability that is better than one part per million over a wide temperature range.

Figure 3:
The transmit converter board.  The large can is the 10 MHz OCXO, the RF input and attenuator are in the lower-right corner and the mixer/transformers are in the bottom-center.  The driver amplifier is visible in the upper-left corner.
Click on the image for a larger version.

The output of the 10 MHz oscillator is amplified/buffered if necessary and then divided-by-two by U102 - a 74HC(T)7474 - a chip that is still readily available as a DIP part.  This divide-by-two step is necessary as the mixer requires a 50% duty cycle for best balance and efficiency.

Inexpensive, stable TCXOs are readily available with 1ppm ratings or better:  One such a part is the Taiten TXETALSANF-10.000000 (Digi-Key 1664-1262-1-ND) which, at the time of writing, costs $2.92 in single quantities and has a rated stability of 0.5 ppm.  This is a tiny 3.3 volt surface-mount device, but it can be easily adapted for this circuit:  The use of a device like this - with an output that is too low to drive TTL directly - would utilize the single-transistor converter seen in the upper-right corner of Figure 2.  Even though this is a 3.3 volt device, the 1.6-1.8 volt drop through a standard (not "ultra-bright) red LED from the 5 volt supply will yield the correct operating voltage.

For an example of using a small SMD TCXO like the Taiten device mentioned above, see the 20 February, 2018 entry of this blog - Better frequency stability for the QRL Labs ProgRock synthesizer - link.  Note that this article describes the use of a 27 MHz TCXO in the same, tiny SMD package as the 10 MHz TCXO noted above.


Switching mixer:

The heart of the converter is U201, a 74HC4066 quad bilateral switch, used as a commutating switching mixer.  While the popular FST3251 (or similar) could have been used, that chip is available only in a surface-mount package while the 74HC4066 is available in DIP and works at least as well in this application - much better than an integrated solution like the NE602.


On the input and output ports of this mixer are simple transformers used to assure a balanced signal in and out and these are trifilar-wound on small ferrite toroids.  For my version I used some FT37-43B toroids because they were on-hand,  but the more-common FT37-43 or FT50-43 could have been used instead with equal results.  The exact number of turns is not particularly important, but a general rule of thumb is for such a transformer's winding to have at least three times the inductive reactance as the operating circuit at its lowest operating frequency:  More inductance is better - within reason.

Because our lowest intended frequency will be 136 kHz, we would calculate the inductance thusly, designing for an inductance that yields at least 3 times the operating impedance at the lowest frequency (e.g. 3x 50 = 150 ohms):

Because:

Z = 2*Pi*F*L

Where:
   Z = Inductive reactance in ohms
   F = Frequency in Hz
   L = Inductance in Henries
   2*Pi = approximately 6.28

To get inductance we rearrange the equation as:

L = Z/(2*Pi*F)

So, for 137kHz and an assumed "Z" of 150 ohms (3x 50 ohms input/output), L =

150 / (6.28 * 136000) = 175uH

Let is now refer to a handy online toroid caculator - toroids.info.  If we have some FT-50-43 cores on-hand we can find this particular toroid, enter the desired inductance and we'll need 20 turns to get 175uH.  After this circuit was completed it was tested and found to provide useful output down to at least 60 kHz, indicating plenty of design margin.

Although a bit difficult to tell from the schematic, the "inside" windings of T201 and T202 are really two of the trifilar windings connected in series and this is used to quadruple the impedance seen by the switch U201 and minimize losses.  Practically speaking, this is probably unnecessary in this application, but it's easy to do.

Figure 4:
A close-up view of the mixer and other support components.  Right to left:  Input attenuator and high-pass filter, input transformer, 74HC4066 mixer (with 74HC74 mixer above it), output transformer and output low-pass filter.
The close-eyed observer will note that the 100 ohm, 2 watt resistors (bottom right, blue devices) are slightly browned from having accidentally set the FT-817 to 5 watts:  No real damage was done!
Click on the image for a larger version.

In some cases builders have been known to apply a mid-voltage DC bias (2.5 volts in this case) to the center of the input/output windings on such a mixer, but that was not done here as testing showed that it didn't seem to make a measurable difference in performance as either a transmit or receive conversion mixer.  If you don't use bias, make sure that these windings can "float" with respect to DC and the local ground.

On the input side may be seen a high-pass filter that nominally blocks signals below 5 MHz.  Perhaps this is overkill, but this was included to eliminate any signals below 5 MHz that might enter the mixer - specifically any local AM broadcast stations that might have strong enough signals to ingress the cable between transceiver and the converter - not to mention the (possibly) very strong MF/LF signal from the output amplifier driven by this converter that might re-enter the signal path and produce spurious signals!

Preceding the mixer is a simple 20dB attenuator pad that is used to reduce the nominal 1 watt from an FT-817 to about 10 milliwatts.  This attenuator was designed to be able to withstand the full 5 watts from the '817 in the event full power was accidentally used.  As noted in the text, a 5-watt 62 ohms non-inductive resistor is ideal, but I didn't have one so I used the resistor combination shown in the diagram, which is more than "good enough".

Following the mixer is a low-pass filter that removes signals above approximately 500 kHz - which includes leakage from the 5 MHz local oscillator and the mixer images in the 10+ MHz area.  Included in this circuit are R206 and C205 which form a crude diplexer to terminate those image frequencies while minimally affecting the desired LF/MF signals.

Bilateral use:

By this time the reader may have noticed that J201 and J202 are labeled as both inputs and outputs.  When this circuit was first built I envisioned making it usable as both a transmit and receive mixer - and this is possible because the signal path is completely passive.  In other words, if one connected a receiver tuned to the 5 MHz area to J201 and LF/MF signals to J202, it would function as a high-performance receive converter as well, albeit with the expect 6-8dB insertion loss of a passive mixer.

The only caveat with its use as transmit-receive mixer is the presence of the 20dB attenuator - but this isn't as much of a problem as one might think:   Receive antennas at LF/MF are typically amplified and the sub-microvolt sensitivity of modern HF receivers means that, in many cases, this additional 20dB of attenuation will not put the LF/MF noise floor below the receiver's noise floor.

Practically speaking, a relay could be inserted at this point, keyed by the transmitter to put the attenuator inline, which would eliminate this loss, but I chose to omit this circuit as it would have been inconvenient to wire this to the transmitter as well - plus it is likely that I would have accidentally transmitted into the mixer when it was in "receive" mode (e.g. no attenuator) and destroyed U201!

Ultimately, I decided to use other receive gear for 630 and 2200 meter reception rather than use this mixer:  An RFSpace SDR-14 is used on 630 meters and a SoftRock Ensemble II (the LF/MF version) along with a 192 kHz sound card is used for 2200 and 1750 meter reception, each sharing a connection from an low-pass filtered, active E-field whip.

Driver amplifier:

This converter will produce a few 10s of milliwatts of linear RF at most so some "help" is needed for driving an external amplifier.  A suitable driver amplifier is depicted in the schematic below:

Figure 5:
Transmit driver amplifier.  This amplifier is linear up to about 200 milliwatts.
Click on the image for a larger version.

This amplifier is based on the venerable 2N5109, a very linear UHF RF amplifier transistor designed for CATV amplifier use and it is still available in a through-hole case for a reasonable price.  This amplifier has moderate-gain (15-20dB) and presents a reasonable 50 ohm load to the mixer and has linear output to at least 200 milliwatts when powered from a 12 volt supply, producing nearly 500 milliwatts when saturated.

As noted in the diagram, the transistor should be heat-sinked as is it is running in the linear range and is pulling a fair amount of current when idle.

The output of this amplifier is intended to be passed along to a high-power amplifier, although it can be used directly if operating QRP (e.g. low power).  On 630 and 2200 meters many operators use amplifiers that are not linear because most of the communications uses modes that transmit only single tones (e.g. CW, JT-9, WSPR) where a nonlinear amplifier will suffice:  Linearity is usually traded for the higher power efficiency of a class D or E power amplifier.

Comment:

The use of a 20dB attenuator with 1 watt of RF output from the FT-817 yields approximately 10-15 milliwatts of drive power which, in conjunction with the amplifier depicted in Figure 5, can drive the amplifier described below to (more or less) saturation.

If you are using a radio with a "transverter output" that is markedly lower than 10 milliwatts, an additional amplification stage may be required to "max out" the power amplifier.

Since this article was written the 2N5109 transistor has become unavailable.  Suitable through-hole replacements - available from DigiKey as of January, 2025 - are the 2N3053 and 2SD822.  With a bit of re-working of the circuit , a power FET such as the IFR510 or IRF511 would also work well as a driver amplifier.

An example power amplifier:

In the figure below, a typical single-ended FET-type power amplifier that can be operated linearly is depicted schematically:

Figure 6:
Typical single-ended power amplifier with an example low-pass filter for 630 meter operation and an autotransformer-type matching network.  Not shown in the diagram is a series 10 ohm resistor between C401 and the gate of Q401 and a series 1k/1watt resistor and 0.1uF capacitor between the drain and gate of Q401 - these having been added to improve stability.
Click on the image for a larger version.

This amplifier has been designed to operate equally well on both 630 and 2200 meters - mostly by making sure that the coupling transformer (T401) and the coupling/decoupling capacitors are chosen appropriately for operation at 137 kHz.  A low-pass filter specific for 2200 meter operation is not shown, but links to proven designs may be found below.

The drive signal is applied via J401, which is capacitively coupled to the gate of Q401, a high-power N-channel switching FET with R401 offering a ground reference a bit of RF "swamping".    T401 transforms the lower impedance of the drain (10-15 ohms) to 50 ohms and this is coupled to the output via capacitors C410 and C411.  Capacitors C406-C408 together form a low-impedance RF bypassing network to remove RF from the power supply lead.

Comment:

Suitable FET devices that may be used for Q401 include, but are not limited to:

• Infineon IPP17N25S3-100 - This device has a rating of 250 volts and 17 amps (Mouser P/N:  726-IPP17N25S3-100).  This device is useful with a power supply voltage of up to 20 volts.

• D3 semiconductor D3S080N65B - This device has a rating of 650 volts and a current rating of 38.3 amps  (Mouser P/N: 488-D3S080N65B-U)  This device is more appropriate when operating the amplifier on a >20 volt supply.
  

• OnSemi NTHL070N120M3S - This is a SiC (Silicon Carbide) power FET rated at 1200 volts and 70 amps.  This device will happily run on a >20 volt power supply is more rugged and fault-tolerant than either of the above.
  

Either of the above devices can tolerate a 33 volt supply (and an operating current of 3.4 amps) while producing 75+ watts into a wide variety of loads - but neither of them are completely impervious to abuse.  As noted above, use of the the 650 volt FET or the SiC FET is recommended at higher supply voltages.

An optional bias supply is shown below the main circuit, using a 78L05 5 volt regulator as a stable voltage reference which is then made adjustable via potentiometer R402.  The bias is applied to the gate of Q401 via resistor R401, a 100 ohms, 2 watt resistor, the "cold" (non-RF) end of which is RF-grounded by C402 and C403.  If the bias supply is omitted,  the "bottom" of R401 would be connected directly to ground.

Figure 7:
The power amplifier portion, built on perforated prototype board, using a Hammond 1590D enclosure as the heat sink.  The layout allows the (relatively) easy replacement of the RF output transistor (upper-right corner of the board.)  Because one is likely to blow up the occasional output transistor, one should make it easily replaceable and keep several on hand.
Click on the image for a larger version.

Transistor Q401 is a high-power N-channel FET of the sort found in mains-powered switching power supplies and as such, it should have a voltage rating of at least 200 volts (a higher voltage rating like 400 volts is better!) and a current rating of at least 15 amps.  This transistor should be well heat-sinked:  The body of the Hammond 1590D enclosure has proven adequate for continuous duty operation with the amplifier operating into a reasonably-well matched load at power levels of up to 80-100 watts DC input:  Higher input power levels than this should be used with a "proper" heat sink and/or forced-air cooling.

A power supply voltage of 12-15 volts will produce RF power output in the 15-20 watt range while a 30 volt supply limited to 3.5 amps will yield 60-90 watts of RF power.

Particularly at this higher supply/output end of this range, a higher-voltage (>=400 volt) power FET is recommended to be able to withstand mismatch conditions that could occur if the antenna system is detuned.  It is strongly recommended that a current-limited power supply be used with its threshold current set just above the maximum current pulled by the amplifier when driven to full output into the intended load:  If a poor antenna match occurs, the transistor is somewhat protected and if there is a transistor failure, the damage to other components will be minimized.

In typical "non-linear" use the bias is either set to zero volts (R402's wiper to ground or R401 grounded) or increased such that there is only a few milliamps of FET no-signal idle current:  This latter condition slightly reduces the RF drive requirement and may yield slightly higher RF output.

Amplifier stability:

It is an unfortunate fact that while inexpensive power FETs can be used as inexpensive, high-power amplifiers that they are also easy to blow up when operated at radio frequencies.  As noted in the caption of Figure 7, several components were added to improve overall stability - namely the 10 ohm resistor in series with the gate and the RF drive and a series-connected 1k resistor and 0.1uF capacitor between the drain and gate.

This amplifier is powered from a current-limited adjustable bench-top supply (a Tenma 72-6628) that can produce 34 volts at a bit more than 3.25 amps.  Having strict current limiting goes a long way toward protecting the amplifier under fault conditions (mistuned, open or shorted antenna) and has likely prevented the need to replace the transistor several times - but care is still warranted.

On the air, the amplifier has been quite reliable - never having failed while in service unless something went amiss with the antenna or match system - but if "proper" reverse power protection had been included, it's likely that I'd still be running the original transistor.

Linear operation at reduced power:

If linear operation is desired it is strongly recommended that the power supply voltage be limited to around 18 volts as the amplifier circuit can become unstable at higher voltages (30 volts) when biased into the linear range, instantly destroying the gate-source junction of the FET.  Using an 18 volt supply, approximately 25 watts PEP of RF was produced with the transistor biased at about 200mA:  On the air, the audio report during a 630 meter SSB QSO was good and the observed spectra using a waterfall display appeared to be clean.


Similar RF amplifier circuits may be found at the "472kHz.org" Useful Links web page - see the "transmitting" section, near the bottom of that page.

The low-pass filter:


Figure 6, above, also depicts a 630 meter low-pass filter that adequately removes the 2nd and higher harmonics and this filter is shown in the figure below.

Figure 8:
Low-pass filter for 630 meters using 17 AWG wire wound on PVC forms oriented to minimize cross-coupling.  Silver-mica capacitors were used but high-quality polypropylene units will work as well.  If one attempts the use of ceramic capacitors, use only C0G/NP0 types with a 500 volt rating or greater.  Green PET insulating tape can be seen under the coils to provide insulation to the ground plane.  The filter was built into the lid of the amplifier's aluminum enclosure.
Click on the image for a larger version.

Additional harmonic suppression will occur in any practical antenna matching network (e.g. series loading coil) of reasonable "Q".

Other low-pass filter designs suitable for high-power 630 and 2200 meter transmitting are described by W1VD at his web site:

• LF and MF kilowatt+ low-pass filters at W1VD.com

  
Other entries on related topics found at this site:

• KA7OEI now QRV on 630 and 2200 meters -  This page describes the actual use of this transmit converter as well as a practical matching network and antenna system.
• A (semi)-typical suburban E-field whip receive system for the 630 and 2200 meter bands - link.  This page describes various techniques to effectively use an E-field whip in a noisy environment for LF/MF reception.
• Low-pass filter for LF/MF reception - link.  Many HF rigs, while capable of reception at 630 meters (and possibly 2200 meters) are overloaded by signals from local AM broadcast stations.  This filter can help reduce this problem.
• Completely containing switching power supply RFI - link.  Switching power supplies are the bane of the LF/MF listener and this page - and the pages linked to it - contain information about how these devices may be quieted.

Other web sites that have information on 630 and 2200 meters: 

This list is by no means comprehensive.  Peruse the "links" sections on the sites below for even more information.

• NJD Technologies - link to archive.org capture  - This web page has a wealth of information related to 630 meter operation, propagation and reports of activity, plus lists of known-active operators on both 630 and 2200 meters.  This web site also has many links to others that have credible information on LF and MF band topics.
• W1TAG's web site - link  - John, W1TAG, has long been an experimenter and operator on the MF and LF bands.  This site has details on equipment both for operating and measuring performance at these frequencies.
• W1VD's web site - link - Jay, W1VD, has long been an experimenter on the LF/MF bands and this page offers a lot of information on equipment for transmitting and receiving on these bands.
• Antennas by N6LF - link - The callsign gives you the clue that this guy likes LF/MF operation.  This page includes detailed information on LF/MF antennas and how to characterize/improve them.

This page stolen from ka7oei.blogspot.com

[END]