Resurrecting my FE-5680A Rubidium frequency reference

Fig 1:
The Hammond 1590 aluminum case
housing the FE-5860A rubidium osc-
oscillator and other circuitry - the
markings faded by time and heat.
Click on the image for a larger version.

Recently I was getting ready for the October 14, 2023 eclipse, so I pulled out my two 10 MHz rubidium frequency references (doesn't everyone have at least one?) as I would need an accurate and (especially) stable frequency reference for transmitting:  The details of what, why and how will be discussed in a post to be added in the near future.

The first of these - my Efratom LP-101 - fired up just fine, despite having seen several years of inactivity.  After letting it warm up for a few hours I dialed it in against my HP Z3801 GPSDO and was able to get it to hold to better than 5E-11 without difficulty.

My other rubidium frequency reference - the FEI FE-5680A - was another matter:  At first, it seemed to power up just fine:  I was using my dual-trace oscilloscope, feeding the 'Z3801 into channel 1 and the '5680A into channel 2 and watching the waveforms "slide" past each other - and when they stop moving (or move very, very slow) then you know things are working properly:  See Figure 2, below, for an example of this.

That did happen for the '5680A - but only for a moment:  After a few 10s of seconds of the two waveforms being stationary with respect to each other, the waveform of the '5680A suddenly took off and the frequency started "searching" back and forth, reaching only as high as a few Hz below exactly 10 MHz and swinging well over 100 Hz below that.

My first thought was something along the lines of "Drat, the oven oscillator has drifted off frequency..."

Fig 2:
Oscillogram showing the GPS reference (red)
and the FE-5680A (yellow) 10 MHz signals
atop each other.  Timing how long it takes for the
two waveforms "slide" past each other (e.g. drift
one whole cycle) allows long-term frequency
measurement and comparison.
Click on the image for a larger version.

As it turns out, that was exactly what had happened.

Note: 

 I've written a bit more about the aforementioned rubidium frequency references, and you can read about them in the links below:

• A portable 10 MHz Rubidium Frequency Refrence using the Efratom LPRO-101  - LINK 

• A portable 10 MHz Rubidium Frequency Reference using the FE-5680A - LINK.
  

Oscillator out of range

While it is the "physics package" (the tube with the rubidium magic inside) that determines the ultimate frequency (6834683612 Hz, to be precise) it is not the physics package that generates this frequency, but rather another oscillator (or oscillators) that produce energy at that 6.834682612 GHz frequency, inject it into the cavity with the rubidium lamp and detect a slight change in intensity when it crosses the atomic resonance.

In this unit, there is a crystal oscillator that does this, using digital voodoo to produce that magic 6.834682612 GHz signal to divine the hyperfine transition.  This oscillator is "ovenized" - which is to say, the crystal and some of the critical components are under a piece of insulating foam, and attached to the crystal itself is a piece of ceramic semiconductor material - a PTC (positive temperature coefficient) thermistor - that acts as a heater:  When power is applied, it produces heat - but when it gets to a certain temperature the resistance increases, reducing the current consumption and the thermal input and the temperature eventually stabilizes.

Because we have the rubidium cell itself to determine our "exact" frequency, this oven and crystal oscillator need only be "somewhat" stable intrinsically:  It's enough simply to have it "not drift very much" with temperature as small amounts of frequency change can be compensated, so neither the crystal oven - or the crystal contained within - need to be "exact".

Fig 3:
The FE-5680A itself, in the lid of the
case of the 1590 box to provide heat-
sinking.  As you can see, I've had this
unit open before!
Click on the image for a larger version.

What is required is that this oscillator - which is "pullable" (that is, its precise frequency is tuned electronically) - must be capable of covering the exact frequency required in its tuning range:  If this can't happen, it cannot be "locked" to the comparison circuitry of the rubidium cell.

The give-away was that as the unit warmed up, it did lock, but only briefly:  After a brief moment, it suddenly unlocked as the crystal warmed up and drifted low in frequency, beyond the range of the electronic tuning.

Taking the unit apart I quickly spotted the crystal oscillator under the foam and powering it up again, I kept the foam in place and watched it lock - and then unlock again:  Lifting the foam, I touched the hot crystal with my finger to draw heat away and the unit briefly re-locked.  Monitoring with a test set, I adjusted the variable capacitor next to the crystal and quickly found the point of minimum capacitance (highest frequency) and after replacing the foam, the unit re-locked - and stayed in lock.

Bringing it up to frequency

This particular '5680A is probably about 25 years old - having been a pull from service (likely at a cell phone site) and eventually finding its way onto EvilBay as surplus electronics.  Since I've owned it, it's also seen other service - having been used twice in in ground stations used for geostationary satellite service as a stable frequency reference, adding another 3-4 years to its "on" time.

As quartz crystals age, they inevitably change frequency:  In general, they tend to drift upwards if they are overdriven and slowly shed material - but this practice is pretty rare these days, so they seem to tend to drift downwards in frequency with normal aging of the crystal and nano-scale changes in the lattice that continue after the quartz is grown and cut:  Operating at elevated temperature - as in an oven - tends to accelerate this effect.

By adjusting the trimmer capacitor and noting the instantaneous frequency (e.g. adjusting it mechanically before the slower electronic tuning could take effect) I could see that I was right at the ragged edge of being able to net the crystal oscillator's tuning range with the variable capacitor at its extreme low end, so I needed to raise the natural frequency a bit more.

If you need to lower a crystal's frequency, you have several options:

• Place an inductor in series with the crystal.  This will lower the crystal's in-circuit frequency of operation, but since doing so generally involves physically breaking an electrical connection to insert a component, this is can be rather awkward to do.

Fig 4:
The tip of the screwdriver pointing at the added 2.2uH
surface-mount inductor:  It's the black-ish component
at sort of a diagonal angle, wired across the two
crystal leads.
Click on the image for a larger version.

• Place a capacitor across the crystal.  Adding a few 10s of pF of extra capacitance can lower a crystal's frequency by several 10s or hundreds of ppm (parts-per million), depending on the nature of the crystal and the circuit.

Since the electrical "opposite" of a capacitor is an inductor, the above can be reversed if you need to raise the frequency of a crystal:

• Insert a capacitor in series with the crystal.  This is a very common way to adjust a crystal's frequency - and it may be how this oscillator was constructed.  As with the inductor, adding this component - where none existed - would involve breaking a connection to insert the device - not particularly convenient to do.
  

• Place an inductor across the crystal.  Typically the inductance required to have an effect will have an impedance of hundreds of ohms at the operating frequency, but this - like the addition of a capacitor across a crystal to lower the frequency - is easier to do since we don't have to cut any circuit board traces.
  

With either method of tweaking the resonance of the oscillator circuit, you can only go so far:  Adding reactance in series or parallel will eventually swamp the crystal itself, potentially making it unreliable in its oscillation - and if that doesn't happen, the "Q" is diminished, potentially reducing the quality of the signal produce and furthermore, taking this to an extreme can reduce the stability overall as it starts to become more temperature sensitive with the added capacitor/inductor than just the crystal, alone.

In theory, I could have placed a smaller fixed capacitor in series with the trimmer capacitor  - or used a lower-value capacitor - but I chose, instead, to install a fixed-value surface-mount inductor in parallel with the crystal as it would not require cutting any traces.  Prior to doing this I checked to see if there was any circuit voltage across the crystal, but there was none:  Had I seen voltage, adding an inductor would have shorted it out and likely caused the oscillator to stop working and I would have either reconsidered adding a series capacitor somewhere or, more likely I would have placed a large-value (1000pF or larger) capacitor in series with the inductor to block the DC.

"Swagging" it, I put a 2.2uH 0805 surface-mount inductor across the crystal and powered up the '5680A and after a 2-3 minute warm-up time, it locked.   After it had warmed up for about 8 minutes I briefly interrupted the power and while it worked to re-establish lock I saw the frequency swing nearly 100 Hz below and above the target indicating that it was now more less in the center if its electronic tuning range indicating success!  As can be seen from Figure 4, there is likely enough room to have used a small, molded through-hole inductor instead of a surface-mount device.

Fig 5:
The crystal is under the round disk (the PTC
heater) near the top of the picture and the
adjustment capacitor is to the right of the
crystal.
Click on the image for a larger version.

With a bit of power-cycling and observing the frequency swing while the oscillator was hot, I was able to observing the electronic tuning range and in so-doing, increase the capacitance of the trimmer capacitor very slightly from minimum indicating that I now had at least a little bit of extra adjustment room - but not a lot.  Since this worked the first time I didn't try a lower value of inductance (say, 1uH) to further-raise the oscillator frequency, leaving well-enough alone.

Buttoning everything back up and putting it back in its case, everything still worked (always gratifying!) and I let the unit "burn in" for a few hours.

Comparing it to my HP Z8530 GPS Disciplined oscillator via the oscilloscope (see Figure 2) it took about 20 minutes for the phase to "slide" one entire cycle (360 degrees) indicating that the two 10 MHz signal sources are within better than 10E-10 of each other - not too bad for a device that was last adjusted over a decade ago and as seen about 15000 operational hours since!

 

* * *

 

Follow-up:  A few weeks after this was originally posted I had this rubidium reference with me at the October, 2024 Eclipse event as a "hot standby", its frequency being compared to the LPRO-101 - which was the active, on-the-air unit - using an oscilloscope.

 

This (repaired) unit fired up and locked within 5 minutes at the cool (45F/7C) ambient temperature and remained stable for the several hours that it was powered up.

 

I used this unit again form the April, 2024 Eclipse - this time, from a location in central Texas and, again, it worked perfectly.

* * * * *

 

This page stolen from ka7oei.blogspot.com

 

[END]

 

Exploring the NDK 9200Q7 10 MHz OCXO (Oven-controlled Crystal Oscillator)

Figure 1:
The NDK 9200Q7 OCXO.  This unit, pulled from
used equipment, is slightly "shop-worn" but still
serviceable.  The multi-turn tuning potentiometer
is accessible via the hole at the lower-left.
Click on the image for a larger version

The NDK 9200Q7 (pictured) is an OCXO (Oven-Controlled Crystal Oscillator) that occasionally appears on EvilBay or surplus sites.  While not quite as good a performer as the Isotemp 134-10 (see the 17 October, 2017 Blog entry, "A 10 MHz OCXO" - Link) it's been used for a few projects requiring good frequency stability, including:

• The 146.620 Simulcast repeater system.  One of these is used at each transmitter site, which are held at 4 Hz apart to eliminated "standing nulls" - and they have stayed put in frequency for over a decade. (This system is described in a series of previous blog entries starting with  "Two Repeaters, One System - Part 1" - Link).
  

• 10 GHz transverter frequency reference.  One of the local amateurs used one of these units to hold his 10 GHz frequency stable and it did so fairly well, easily keeping it within a  hundred Hz or so of other stations:  This was good enough to allow him to be easily found and tuned in, even when signals were weak.
  

• Frequency reference for a 2200/630 MHz transverter.  While a low frequency, the use of very low-rate digital modes like FST4W-300, 900 and 1800 require sub-Hz frequency stability.  (This transmit converter was described in the 19 January, 2019 entry, "A transmit converter (and amplifier) for 630 and 2200 meters - Link).

At least some of these units were pulled from scrapped VSAT (Very Small Aperture SATellite) terminals so they were designed for both stability and the ability to be electronically tuned to "dial in" the frequency precisely.

Testing and experience shows that given 10-15 minutes to thermally stabilize, these units are perfectly capable of holding the frequency to better than 1 part in 10⁸ - or about 1 Hz at 100 MHz - and since any of these units that you are likely to find about are likely to be 25-30 years old, the intrinsic aging of the quartz crystal itself is going to be well along its asymptotic curve to zero.

Figure 2:
The bottom of the OCXO, annotated to show the various
connections.
Click on the image for a larger version.

Using this device

In its original application, this device was powered from a 12-15 volt supply, but if you were to apply power and give it 5-15 minutes to warm up, you would probably be disappointed in its accuracy as it would not have any sort of external tuning input to get it anywhere close to its intended frequency.

Because of the need for it to be electrically tuned, this device is actually a VCXO (Voltage-Controlled Crystal Oscillator) as well and as such, it has a "Tune" pin, identified in Figure 2.  Nominally, the tuning voltage was probably between 0 and 10 volts, but unless a voltage is applied, this pin will naturally drift close to zero voltage, the result being that at 10 MHz, it may be a dozen or two Hz low in frequency.

Adding a resistor

The easiest "fix" for this - to make it operate "stand-alone" - is to apply a voltage on the pin.  If your plans include locking this to an external source - such as making your own GPSDO (GPS Disciplined Oscillator) then one simply need apply this tuning voltage from a DAC (Digital-to-Analog Converter) or filtered PWM output, but if you wish to use this oscillator in a stand-alone configuration - or even as an externally-tuned oscillator, a bit of modification is in order.

Figure 3:
This shows the 10k resistor added between the internal 5 volt
source and the "TUNE" pin to allow "standalone" operation.
Click on the image for a larger version.

The OCXO may be disassembled easily by removing the small screw on each side and carefully un-sticking the circuit board from the insulation inside.  Once this is done, you'll see that there are two boards:  The one on the top is part of the control board for the heater/oven while the bottom houses some of the oscillator components.

Contained within the OCXO is a 78L05 five-volt regulator which is used to provide a voltage reference for the oven and also likely used as a stable source of power for the oscillator - and we can use this to our advantage rather than need to regulate an external source which, itself, is going to be prone to thermal changes.

Figure 3 shows the addition of a single 10k resistor on the top board, connecting the "TUNE" pin to the output of this 5 volt regulator.  By adding this resistor, the TUNE pin allows one to use this OCXO in a "standalone" configuration with no connection to the "TUNE" pin as it is is automatically biased to a temperature-stable (after warm-up) internal voltage reference and can then be used as-is as a good 10 MHz reference, using the onboard multi-turn potentiometer to precisely set the frequency of operation.

Figure 4:
More pictures from inside the OCXO
Click on the image for a larger version.

Another advantage of adding the internal 10k resistor is that it's easy to reduce the TUNE sensitivity from an external voltage:  This value isn't critical, with anything from 1k to 100k likely being usable.  Testing shows that by itself, the oscillator is quite table and varying the TUNE voltage will adjust it by well over 10 Hz above and below 10 MHz.

The inclusion of the 10k internal resistor may also be of benefit.  In many cases, having a much narrower electronic tuning range than this will suffice so a resistor of 100k (or greater) can be used in series with the TUNE pin, between it and an external tuning voltage, acting as a voltage divider.  Doing this will reduce the tuning range and it can also improve overall stability since much of the tuning voltage will be based on the oscillator's already-stable 5 volt internal source.  The stability of the OCXO itself is such that even with a 10-ish:1 reduced tuning range due to a series 100k resistor, there is still far more external adjustment range than really necessary to tune the OCXO and handle a wide range of external temperatures.

The actual value of the added internal resistor is unimportant and could be selected for the desired tuning/voltage ratio based on the external series tuning resistor and the impedance of the tuning voltage.

When reassembling the OCXO, take care that the insulation inside the can is as it was at the time of disassembly to maximize thermal stability and, of course, be sure that the hole in the can lines up with the multi-turn potentiometer!

Operating conditions

Figure 5:
Even more pictures from inside the OCXO.
Click on the image for a larger version.

The "official" specifications of this OCXO are unknown, but long-term use has shown that it will operate nicely from 12-15 volts - and it will even operate from a 10 volt supply, although the reduced heater power at 10 volts causes warm-up to take longer and there may not be sufficient thermal input for the oven to maintain temperature at extremely low (<15F, <-9C) temperatures unless extra insulation is added (e.g. foam around the metal case.)

It is recommended that if one uses it stand-alone, the voltage source for this device be regulated:  While the on-board 5 volt regulator provides a stable reference without regard to the supply voltage, the amount of thermal input from the oven will change with voltage:  More power and faster heating at higher voltage.  While you might think that this wouldn't affect a closed-loop system, it actually does owing to internal thermal resistance and the fact that due to loss to the environment, there will always be a thermal gradient between the heater, the temperature-sensitive circuitry, and the outside world - and changing the operating voltage and thus the amount of heater power will subtly affect the frequency.

Finally, this oscillator - like any quartz crystal oscillator that you are likely to find - is slightly affected by gravity:  Changing orientation (e.g. turning sideways, upside-down, etc.) of this oscillator affects its absolute frequency by a few parts in 10E-8, so if you are interested in the absolute accuracy and stability, it's best to do the fine-tuning adjustment with it oriented in the same way that it will be used and keep it in that orientation.

* * * * * * * * *

This page stolen from ka7oei.blogspot.com

[End]

Locking the Icom IC-910H to an external 10 MHz (GPS) reference

In late 2009 my friend Bryan, W7CBM, came to me with a project that he had in mind:  "Can we lock my Icom IC-910H to my 'Z-box'?" - in other words, could the 10 MHz output from his Z-3801 GPS Disciplined Oscillator - known to be accurate to better than one part in 100 million - be used to lock his tri-band (2 meters, 70cm and 23cm) all-mode radio to frequency?

Figure 1:
The front panel of the modified Icom IC-910H.
Click on the image for a larger version.

During the initial discussion he'd brought with him an article where Rex, VK7MO, had done a similar thing (see the article on the VK3HZ site from the web archive - link) using an external box to provide a precise version of the radio's 30.2 MHz reference - but he wanted it to be contained entirely within the radio.  

In looking at the requirements and designing the circuit in my head, I decided that we could make it simpler, smaller and easier to use - and with these ideas in mind, I wrote down the specifications for a 30.2 MHz fundamental-mode crystal and he sent an order off to International Crystal.

About 2 months later - in early 2010 - we got back together in my ham shack, crystal in hand, and it was then that I decided that I'd better get around to designing the circuit, so I scribbled the vestiges of a schematic onto a piece of paper and built several circuits that would fit into the aluminum box that Bryan had milled out. At the end of about 3 hours we had a circuit that would faithfully lock the 30.2 MHz crystal oscillator to a 10 MHz external source.  This circuit was fairly small and consisting of two boards:  The amplifier/counter/PLL section wired on prototype board while the VCXO itself was constructed "dead bug" on to a piece of copper-clad PC board material as seen in Figure 3.

"Patience is a virtue - but this is ridiculous!"

And that was where it stopped.  In a case of "out of sight, out of mind", "other fish to fry" - or any number of other excuses - the partly-completed lock unit stayed on a shelf in Bryan's ham shack for a decade, in plain sight.  When I'd go over to his shack, I'd see it as a reminder of a project yet to be completed, but it had become a fixture and was often overlooked.

Until recently.

As it happened, we both had more time available with the onset of winter and we carved out Wednesday evenings to get together to work on various projects and this, being the most senior and nearest completion, came to the top of the pile.  Over the course of a couple evenings we worked on the it, having to pause occasionally to get a part, modify some aspect the circuit's implementation, do some physical machine work, or because we ran out of time - but it is now complete!

How it works:

The schematic diagram is depicted in Figure 2, below.

Figure 2:
The schematic of the lock unit for the IC-910H.  This schematic is a reverse-engineered  version of the (now lost) originals and is likely to be mostly correct.
Click on the image for a larger version.

The VCXO:

The heart of the unit is Y201, a 30.2 MHz fundamental mode crystal in a Colpitts oscillator.  Using D201, a varactor diode (approx. 5-20pF) its frequency is made variable, the center of the electronic tuning range being adjusted by trimmer capacitor C201.  The output of the oscillator is buffered by emitter-follower Q202 to isolate the oscillator from the load.

The 30.2 MHz output goes two places:  To Q103, the 30.2 MHz amplifier, and also to a low-pass filter consisting of C208, L201 and C209 which is then output to the IC-901H's synthesizer.

The 10 MHz chain:

Figure 3:
The lock unit under test prior to installation in the case.
Unfortunately, this is the only picture that I got
of the oscillator portion.  The small PCB is the RF sense
circuit, built using SMD components by Bryan.
Click on the image for a larger version.

The 10 MHz input - which can come from a GPS Disciplined Oscillator (GPSDO), a 10 MHz oven-controlled oscillator (OCXO) or a Rubidium source - is input to and amplified by Q101 to a logic level and buffered by U1a, one section of a 74HC86 quad XOR gate.  

The output of U1a is also applied to a 74HC40103 which is wired as a divide-by-50 counter to yield a 200 kHz output - and this is applied to U2a, a 74HC7474 divide-by-two counter to yield a 100 kHz square wave.

The RF sense circuit:

A sample of the 10 MHz signal from U1a is also applied to the input of Q301, which amplifies it:  This RF gets rectified to DC by D301 and D302 and its presence turns on Q301 which pulls R303 to ground and turns off Q301 which is connected to U301 - a 5 volt regulator that is connected to the +5 volt lead of the original TCXO in the IC-910H:  In this way, the internal oscillator in the IC-910H is enabled when there is no 10 MHz signal, but disabled when it is connected.

Also connected to the emitter of Q301 is PNP power switch Q203 which, when R208 is pulled to ground when Q301 turns off, applies power to U201 - a 9 volt regulator - to power up the 30.2 MHz oscillator when the external 10 MHz source is applied, preventing both oscillators from being turned on at the same time.

The Harmonic mixer: 

A sample of the 30.2 MHz signal applied to Q103 is amplified and applied to U1b, another XOR gate buffer, which is then applied, along with the 10 MHz from U1a, into U1d - yet another XOR gate.  This gate acts as a harmonic mixer:  By virtue of the multiplying action of the XOR gate, the 3rd harmonic of the 10 MHz input mixes with the 30.2 MHz input and at the output of this gate is a small amount of the difference frequency - 200 kHz - which easily is filtered by L101 and C103 and amplified by Q102.

Figure 4:
The unit in place - final test.  SMA connectors are used for
10 MHz input and 30.2 MHz output and
feedthrough capacitors are used for the 13.8 volt DC
input and the switched 5 volts for the TCXO.
Click on the image for a larger version.

The use of a harmonic mixer is a very old technique and it has an advantage of simplicity over a more "conventional" digital divider network - albeit more "analog".

A more "conventional" way of doing this might be to divide both the 30.2 MHz and 10 MHz signals down to a common sub-multiple - say, 200 kHz - but to do so would require both a divide-by-50 (to take the 10 MHz down to 200 kHz) and a divide-by-151 (to take the 30.2 MHz down to 200 kHz).  This method works, but adds the a bit of hardware (an additional divider) and, more importantly, these divider steps and subsequent comparisons reduce the PLL loop gain.

By contrast, directly using the 3rd harmonic of the 10 MHz reference to mix with the 30.2 MHz, the 200 kHz difference (ultimately 100 kHz - see below) may be used directly - and loop gain preserved, potentially improving PLL performance and simplifying the design.

The comparison with the reference frequency:

The 200 kHz "difference" signal from the harmonic mixer, filter and amplifier is applied to the divide-by-to circuit U2d to yield to yield a 100 kHz square wave.  The 100 kHz square wave from the divided-down 10 MHz reference signal and that from the 100 kHz "difference" signal are applied to U1c, an XOR gate, which is used as a phase detector.  As the phases of the 100 kHz from the reference signal and that of the difference signal "slide" past each other, the voltage - smoothed by R107 and C107 - will vary from 0 to 5 volts.  If, as an example, C201 in the 30.2 MHz crystal oscillator is adjusted so that 2.5 volts applied to the "VCXO Tune" line, this will cause the crystal oscillator to lock to the reference when the two signals are 90 degrees apart, being steered back onto frequency if they start to drift apart.  

I chose to use an XOR gate as a phase detector over a conventional phase/frequency detector because other than the desired DC component, the lowest-frequency component from its output cannot be lower than the comparison frequency - 100 kHz, in this case, with the vast majority of the energy being 200 kHz and harmonics.  In comparison, many of the flip-flop phase/frequency detectors tend to output "occasional" pulses at very low frequency when at/near lock, which are nearly impossible to filter out.  There is a minor penalty, though:  An XOR gate phase detector requires use of 50% duty cycle square waves to work most efficiently, so each of its inputs is divided-by-two by a single 74HC74 dual flip-flop.

Figure 5:
Power connection to the original TCXO - L511 was removed.
Click on the image for a larger version.

Interfacing to the IC-910:

Switching the internal oscillator:

Bryan's IC-910 has the standard TCXO - X512 (the "CR-452") rather than the "High Stability" option (the "CR-293").  Either unit operates at 30.2 MHz, but there is a difference:  The standard TCXO operates from 5 volts while the high stability unit operates directly from the 13.8 volt supply.  Because the internal oscillator must be disabled when another source is applied, one will need to do one of two things, depending on how the radio is configured:

• Because this radio had the standard TCXO (CR-452 a.k.a. X512), inductor L511 (on the IC-910H's PLL board) was removed to make the power externally switchable and L510 and C501 (the "L510" on Figure 2, above) was connected to power X512.  It is this voltage that is switched by Q303 and regulated by U301 to provide switchable 5 volts.

• If the "High Stability" option ("CR-293") had been present (as described in the VK7MO case) we would have interrupted the 13.8 volt supply at C511/C512 (on the IC-910H PLL board) and switched it using Q303 directly rather than regulated to 5 volts by U301.  Comment:  It is unknown how much current the high stability oscillator consumes so a slight modification of the Q302 circuit might be required to do this.
  

Another difference between the way the two oscillators are interfaced appears to have something to do with the output level.  The standard TCXO outputs an RF signal of about 1.2 volts peak-to-peak while it can be seen from the IC-910H service manual that R515 is in series with the output of the high stability oscillator - presumably to reduce its level.

Injecting the locked 30.2 MHz signal:

Figure 6:
Connection of the 30.2 MHz to the PLL board showing
 the added D501 and L502.
Click on the image for a larger version.

Initially we simply connected the external 30.2 MHz in parallel with the output of the original TCXO, hoping that it would go "Hi-Z" when it was powered down - but that did not work:  When the original oscillator was powered down, its output was effectively shorted to ground, dropping the 30.2 MHz signal down to about 100 millivolts, so this signal was applied, instead, to the junction of variable resistor R570 and R572, using R570 to isolate it from the powered-down oscillator.  For this reason, diode D501 was implemented:  This diode - and L502 to provide a DC return - are connected directly at the junction of R570/R572:  When the external reference is activated, diode D501 is biased via R207, turning it on and connected the output of the 30.2 MHz VCXO to the IC-910H's PLL circuit.

If the external reference is not activated, Q203 - the VCXO power switch - is off and no voltage is applied to diode D501 via R207 and it remains "off", effectively isolating the original oscillator from the powered-down VCXO:  By placing the diode at the end of the coax, farthest from the external reference, there is minimal effect on the signal by that coax to the IC-910H's internal oscillator when the external reference is not being used.

Mechanical installation within the IC-910H:

Figure 7:
The back panel of the modified IC-910H.  The added BNC
connector is in the lower-left corner - the location of the
original ground screw, now relocated to the opposite corner.
Click on the image for a larger version.

Bryan had machined the box out of a chunk of aluminum back in 2010, sizing it to just fit (in all three dimensions) on the lid of the PLL unit.  As originally equipped, there are two brackets screwed down to the lid - apparently for the mounting of an optional voice synthesizer and DSP board - but these brackets were removed to make room.  Two SMA connectors were then mounted to the new box - one for the 10 MHz input and the other for the 30.2 MHz output, into the PLL board.  A pair of 1000pF feedthrough capacitors provide passage for the DC power into the box and the switched 5 volt output to the original TCXO on the PLL board.

Not shown (because I forgot to take the photo) is the connection to the switched 13.8 volt supply:  This was connected to the same point on the PLL board as depicted in the VK7MO document mentioned above - except, of course, that the trace did not need to be cut as would have been necessary to switch the power if the high-stability oscillator had been fitted.

The hole on the rear panel for the ground post was drilled out to permit mounting of a single-hole BNC connector with an already-fitted cable with attached SMA connector as can be seen in Figure 4.  This location for the BNC connector was slightly problematic as it somewhat blocked the screw to hold down the cover, but maneuvering of the connector, the use of tweezers and a small-diameter screwdriver permitted its installation.  In the opposite corner (the far-right in Figure 7) a new hole was drilled and tapped for the grounding post.

Spectral purity:

There was a small of concern that the spectral purity of the transceiver with the new reference oscillator would be worse than the original as I'd made no attempt to construct a very low noise oscillator (e.g. a lightly-loaded Butler or similar) so I compared the spectrum with both the internal oscillator and the "new", externally-locked oscillator on the various bands  - particularly on 23cm.

Figure 8:
Transmitter spectrum +/-500 kHz of a CW
carrier on 23cm as seen on an HP-8562A.
Click on the image for a larger version.

On 2 meters and 70cm, very weak (-70dBc) spurs at +/- 200 kHz - the main component of the output of the phase detector - were noted, barely above the broadband noise floor of the transmitter itself - but these were pretty much absent on 23cm as can be seen in Figure 8.  If these had been of concern, it would have been easy to further-improve the loop filter - which is currently a very simple R/C design as evidenced by Figure 2.  The fact that the plot in Figure 8 was made with the analyzer's resolution bandwidth set to 300 Hz should be an indication as to how low these 200 kHz components really are!

Another possible concern was closer-in phase noise:  Would various noise sources of the new circuits (VCXO phase noise, counter jitter, 1/F noise from regulators, loop noise, etc.) cause notable degradation?

Figure 9 gives a clue:  For this test, trace "A" is the original TCXO and trace "B" (the slightly fainter one corresponding with the peak on the right) was produced using the new, externally-locked reference.  As can be seen, the "close-in" phase noise performance of this radio isn't super great, anyway, but the two "noise humps" on either side of the carrier appear to be identical.

This trace also shows a slight difference in frequency, with the original TCXO (the peak on the left) being slightly low in frequency compared to the GPS-referenced, externally locked version - both showing identical amounts of phase noise indicating that the IC-910H is not degraded by this addition.

Figure 9:
A comparison of the close-in phase noise
using the original TCXO (left peak) and the
new, externally locked oscillator (right peak).
Click on the image for a larger version.

Conclusion:

Even thought it has been a long time in the making, this project is complete - and working as well as we hoped that it would.  I'm gratified that a mere decade ago, the circuit that I scribbled onto a piece of paper - and then built one evening works just as it was expected, with no significant modification!

* * *

 P.S.  Alas, if you wanted to order a crystal, yourself, International Crystal Mfg. is no more, but custom crystals are still available via Quartslab - link, and Krystaly - link - to name but two places.

NOTE:  "Quartslab" stopped doing business in 2021 and "Klove" appears to have acquired the business - link.

(You would have to check with your chosen manufacturer to see if they will make a 30.2 MHz fundamental crystal, though - either that or modify the oscillator to use a 3rd overtone crystal.)

Comment:  Since this article was originally published, devices like the Leo Bodnar GPS reference link have become available that can produce the 30.2 MHz reference required by this radio directly.  This device uses GPS-based timing to set an oscillator to the desired frequency that is programmable - typically with accuracy in the range of 10E-10 or so:  In other words, it can directly synthesize the 30.2 MHz frequency needed by the IC-910H.

If you wish to generate a 30.2 MHz from a stable 10 MHz source that you already have, the Bodnar device really won't help you.

* * *

This page stolen from ka7oei.blogspot.com.

[End]

 

A 10 MHz OCXO (Oven-controlled Crystal Oscillator)

Figure 1:
The 10 MHz OCXO (lower right) in use with my homebrew
24 GHz transverter.  At 24 GHz, the oven provides excellent frequency
stability, suitable for SSB or even digital modes, while providing a
frequency uncertainty of a few hundred Hz at most.
Click on the image for a larger version.

Why a frequency reference?

When operating on the microwave amateur radio bands, narrowband modes (such as SSB or CW) are often used to maximize the link margin - that is, to be able to talk when signals are weak - and when we use microwave frequencies and narrowband modes such as SSB or CW one must maintain pretty good frequency stability and accuracy:

• Stability is important as a drift of even a few hundred Hz at the operating frequency (in the GHz range!) can affect intelligibility of voice - or, if CW is being used for weak-signal work, such drifting can move the received signal outside the receiver's passband filter!  Having to "chase" the frequency around is not only distracting, but it complicates being able to communicate in the first place.

• Accuracy is also important because it is important that both parties be confident that their operating frequencies are reasonably close.  If a contact is arranged beforehand it is vital that both parties be able to find each other simply by knowing the intended frequency of communication and as long as the two parties are within several hundred Hz of each other it is likely that they will be able to find each other if the path "works" in the first place.  If the error was on the order of several kHz, "hunting" would be required to find the signal and if those signals are weak, they may be missed entirely.

Because achieving such stability and accuracy requires some effort, it is more convenient if our gear is constructed such that it can use a common, external frequency reference and lock to it.  In that way, we need only have one "master" reference rather than several individual references.

Figure 2:
The 10 MHz Isotemp 134-10 OCXO - one of many similar units that
often show up on EvilBay.  A 200uF, 16 volt capacitor is soldered
directly to the supply terminals of the OCXO to provide low-impedance
filtering of any noise that might appear on it - any value from 2000 and
up (to several thousand uF) would be just fine.  The green device is a 10-turn
trimmer potentiometer soldered directly to the OCXO's pins.  This
potentiometer is used to adjust the tuning voltage to precisely set the
frequency and locating it at the OCXO practically eliminates the possibility
of external noise pick-up on the tuning lines and the possibility of the I*R
drop on the wires causing a slight tuning shift as the oven power changes.
The OCXO is mounted in the case using rubber/metal shock mounts with "blobs"
of RTV (silicone) on the sides that prevent it from hitting the inside of the box
should the unit be accidentally dropped.
The corners/edges of the OCXO could be mounted in some stiff foam,
instead - but it should not be thermally insulated by this foam unless you have
demonstrated to yourself that doing so will not reduce the oven's stability.
Click on the image for a larger version.

Having one common frequency reference can also be convenient if one is operating portable using battery power since it can mean that one doesn't need to keep all of those individual pieces of gear "warmed up" all of the time to maintain stability.  If a particular piece of gear can accept an external 10 MHz input, this would allow one to turn on that gear (and drain battery power) only when it is needed.

At this point I might mention that Rubidium frequency references (such as one described here) are also readily available in the surplus market as well that provide at least an order or magnitude greater accuracy and stability and warm up in less time than the crystal reference, so why not always use a Rubidium reference instead of a crystal-based one?  The crystal-based unit is cheaper, easier to package and consumes significantly less power than a Rubidium reference, and the stability/accuracy of a good-quality crystal-based reference is more than "good enough" through at least 24 GHz.  When I go out in the field to do portable microwave work I'll often power up the OCXO after putting it in the car knowing that by the time that I get to my destination and set up, it will be warm and on-frequency.  (To be sure, I bring a Rubidium reference as a "backup"!)

About this frequency reference:

The oscillator:

The goal for this project was to have a "reasonably stable and accurate" reference:  Based on an Isotemp OCXO 134-10 this particular unit has a rated stability of about +/-1.0x10^-8 (+/-1 Hz at 100 MHz) or better after it has warmed up for a while with short term variations approaching +/-1.0x10^-10 (+/-1 Hz at 10 GHz).  In-field observations appear to confirm this stability with tests having shown that this unit seems to be able to hold the 24 GHz local oscillator to within 500 Hz or better with no obvious frequency "warble" once it has had 15-20 minutes or so to warm up -  and it seems to be fairly stable across a range ambient temperatures from "hot" to "below freezing."  The Isotemp unit - and others like it - are readily available on both the new and surplus markets, available via EvilBay and similar and other than having different voltage and stability specifications, they, too, can be integrated into a stand-alone project such as this.

The oven module itself is rated to operate from 13 volts, +/- 2 volts, implying a minimum of 11.0 volts.  Even though testing indicated that it seemed to be "happy" with a supply voltage as low as 9.8 volts or so, it was decided to adhere to the published specifications and in looking around I noticed that most readily-available low-dropout regulators (and those that I had onhand) were not specified to handle the maximum "cold" current of this oven - about 800 mA - so I had to "roll my own" 11 volt "zero-dropout" regulator.  More on alternative regulators, below.

Figure 3:
The inside of the enclosure containing the OCXO, regulator and driver.
On the left is the shock-mounted OCXO while the circuit on the perfboard
is the "zero drop-out" regulator and the 10 MHz distribution amplifier.
The P-channel FET pass transistor can be seen along the top edge of
the die-cast enclosure, bolted to it to dissipate any heat while along
the right edge, inside the enclosure is a piece of glass-epoxy circuit
board material to provide a solid, solderable ground plane for the
distribution outputs and the DC input filtering.

A "zero-dropout" regulator:

Why regulate?  I noted in testing that slight variations of supply voltage (a few hundred millivolts) would cause measurable disturbances in the oscillator frequency due to the changes of the power applied to the heater, taking several minutes to again reach (thermal?) equilibrium.  Since battery operation was anticipated, it is expected that the supply voltage would change frequently between periods of transmit and receive - as well as due to normal battery discharge.  Because I had chosen to use an OCXO that required (at least) 11.0 volts to be run from a "12 volt" lead-acid battery, I needed a circuit that would reliably produce that 11.0 volts even when the battery voltage dipped below 11.5 volts - as it could during heavy transmit loads and the end of a power cable with the battery near the end of its charge.

Referring to the schematic U101, a standard 5 volt regulator (the lower-power 78L05 is a good choice) provides a stable voltage reference for U103, a 741 op amp, which is used as an error amplifier.  A 7805 was chosen as it is readily-available but a Zener diode and resistor could have been chosen:  If a Zener is used, a 5.6-6.2 volt unit is recommended with 2-5 milliamps of bias as this voltage range offers good temperature stability.

If the output voltage is too low, the voltage on pin 3 (the non-inverting input) drops, along with pin 6, the op amp's output which turns on Q103, a P-Channel power MOSFET by pulling it's gate toward ground, which increases the voltage and once the voltage on the wiper of R119 reaches 5 volts - that of the reference, which is applied to pin 2, the non-inverting input - the circuit comes to equilibrium.  A P-Channel FET (a slightly less-common device than an N-channel) was used because it takes 3-5 volts of drain-gate voltage to turn on a FET and it would have been necessary to have at least  3-5 volts above the power supply (about 16 volts) to bias the gate "on" if an N-Channel FET were used whereas we can pull the gate voltage "down" from the supply voltage with a P-channel device.  Furthermore, with the use of a P-Channel power MOSFET the dropout voltage of the regulator is essentially limited to the channel resistance of the that FET.  In theory a PNP (possibly a complimentary pair arrangement) could be used instead if one can tolerate closer to a volt of dropout, but the FET was chosen to minimize the dropout voltage.

In testing, once the oven was warm (a condition in which the OCXO was drawing approximately 250 mA at normal "room temperature") the dropout of the regulator was approximately 50 millivolts - a voltage drop that is a result of the resistance of the wires used to power the unit and the on-resistance of the FET.  This rather simple regulator seems to work quite well, holding the output voltage steady to within a few millivolts over the input voltage range of 11.1 to 17 volts with good transient response.

Figure 4:
The end panel of the OCXO module.  The power feedthrough/capacitor
is on the left, obscured by the red/white power cable with the yellow-ish
"ready" light to the right of it.  The three BNC connectors are the 10 MHz
outputs, allowing multiple devices to be connected while in use and/or while
its calibration is being checked.
Click on the image for a larger version.

"Faster warmup" feature:

This OCXO has a "status" output that, when "cold", outputs about 0 volts and in this state, Q101 is turned off, allowing R112 and R113/D102 to pull its collector high - turning on Q102 - which pulls the gate of Q103 low through R118, turning it fully "on."  In this state the voltage applied to the oven is nearly that of the battery supply and this higher voltage increases the power applied to the oven, allowing it to heat more quickly.  Once the oven's "status" line goes high, Q101 is turned on, illuminating the LED and turning off Q102, allowing the regulator to operate normally.

Note:  When the unit is warming up, the OCXO's voltage is unregulated which means that the supply should be kept below 15.0 volts to stay within the "safe zone" of the ratings of the oscillator itself.

Does the "boosted" voltage actually help the oven warm up faster?  Probably only a little bit, but it took only 4 additional components to add this feature!

Status indicator:

It should be noted that this status line doesn't indicate that the oven has fully warmed up, but only that it's still warming:  At "room temperature" it takes at least another 5 minutes before the frequency will be stable enough for use and another 5 minutes or so after that until it's "pretty close" to the intended frequency and it can be used at microwave frequencies without others having to chase you around.

Why have the indicator light if it doesn't indicate that the unit is actually "ready"?   While this indication isn't perfect if the light isn't on, you can be sure that the frequency output won't be valid for one reason or another.

Because the OCXO itself is somewhat load-sensitive (about +/-1.0x10^-9 - perhaps a few 10s of Hz at 24 GHz) U102 - an LM7171 - is used as a distribution amplifier to both isolate the oven from its loads and to provide fan-out to allow multiple outputs to be driven simultaneously.  The LM7171, a high-output, high-speed op amp, is configured for a gain of 2, providing about 2 volts peak-to-peak output with the drive provided by the OCXO.

Mounting the oven:

Because this unit is intended to be used "in the field" it was decided to mount the OCXO module itself to prevent mechanical shock from affecting the reliability, frequency stability and accuracy and this was done using some rubberized mounting pillars from scrapped satellite equipment while some "blobs" of silicone were placed on the wall of the die-cast enclosure to prevent the OCXO housing itself from directly impacting it should the unit be accidentally dropped.

Figure 5:
Schematic of the OCXO-based unit, including the zero-dropout regulator and 10 MHz distribution amplifier.  It is important that the connection of the "ground" side of the 10 turn calibration potentiometer be made at the OCXO and not elsewhere, this to minimize possible frequency shifts due to I*R losses as the oven's heater power changes.  The diagram shows a
value of 82 ohms for R108-R110 because that what was easily found when it was constructed (and it really doesn't
matter much) but anything between 47 and 100 ohms will be fine, based on your preference.
Click on the image for a larger version.

A few bits of stiff foam could also be used to provide some shock mounting in the corners of the OCXO but be aware that some oven-based oscillators have been known to become less accurate and stable if they are over-insulated and can't radiate at least some of their heat, so don't go overboard.

Important:

Like any crystal oscillator, it is somewhat "position sensitive" in that a frequency shift of 10s of Hz (at 24 GHz) can be observed if the unit is placed on its side, upside-down, etc. due to the effect of gravity on the quartz crystal itself.  While this effect is very minor, it's worth noting when it's being set to frequency and in operation.

In other words, when you calibrate it (see below) do so in the same physical orientation that it will be when it is in use.

DC input protection and filtering:

The input supply is RF-bypassed using a feedthrough capacitor to prevent the ingress or egress of extraneous RF along the power lead.   For power-supply short-circuit and reverse-polarity protection, R101, a 1.1 amp, self-resetting PTC fuse is used in conjunction with D101, a 3-amp diode.

Why not use a forward-biased diode for reverse-polarity protection?  If you recall, we are going through the trouble of minimizing voltage drop-out with our "special" voltage regulator and we could diminish this if we inserted something that caused a voltage drop - even the 0.3-ish volts of a Shottky diode would undermine this effort.

By using the "reverse-biased diode" and the self-resetting PTC fuse we get:

• A means of current limiting should something to wrong:  If we accidentally short something out, the fuse resets itself when the fault is cleared - and no need to worry about not having a spare fuse when one is out in the hinterland trying to operate!
• If the polarity is somehow connected backwards, the diode will conduct and the PTC fuse will "open" - no harm done, returning to normal once the fault is rectified.
• There is minimal voltage drop related to the fuse as its resistance is a fraction of an Ohm under normal conditions which means that we won't compromise the voltage "headroom" of a 12-volt lead-acid battery.

Calibration:

The best way to calibrate this device is to use a GPS disciplined oscillator or a known-good rubidium frequency reference.  If you have access to one of these, connect the output of the OCXO to one channel of a dual-trace oscilloscope and the known-good frequency reference to the other, triggering on one of two signals - it really doesn't matter which one.

Note:  If you have an analog dual-trace oscilloscope with sufficient bandwidth you can use the "X/Y" mode to produce a Lissajous pattern (obligatory Wikipedia reference here) - but this doesn't always work well on modern, digital scopes when high frequencies are involved due to sample aliasing.

Adjusting the 'scope to see one of the waveforms, one should see a stationary wave (the one on which the 'scope is triggered) while the other will be "sliding" past the first.  Adjust the OCXO's frequency (after the OCXO has warmed up for at least 30 minutes - preferably more) while it is sitting in the same physical orientation in which it will be used as this can (slightly) affect frequency.  To assure a more consistent thermal environment it is suggested that the cover of the enclosure containing this circuitry be left on except during the brief periods to access the 10-turn potentiometer unless provisions are made to access it (via a hole) from outside the box.

The OCXO's frequency is then adjusted to minimize the rate at which the two waveforms are moving with respect to each other:  It's sometimes easier to make this adjustment if the 'scope is adjusted so that the two waves are atop each other and about the same size.  With careful adjustment it should be possible to set the frequency so that the two waveforms that take more than 10 seconds to "slide" past each other - maybe longer.  The Isotemp OCXO should, in theory, be able to hold to that "10 second" slide rate over a wide variety of temperature conditions.

If you don't happen to have access to a rubidium reference or a GPS Disciplined oscillator, you can do "reasonably" well by zero-beating the 10 MHz output with the signal from WWV or WWVH, be note that Doppler shifts can cause their apparent frequencies to shift by 1 Hz or more.  I'll leave the explanation of methods of successfully zero-beating an off-air signal to others on the GoogleWeb.

The best time to attempt this is when you are hearing only one of these two stations (assuming that you can ever hear them both) and when it's signal is the most "solid" - that is, it's fading in and out is at minimum.  Often, the worst time to make this sort of measurement is when any part of the radio path between you and WWV (or WWVH) is within a hour or two of sunrise or sunset as this is when the ionospheric layers are in a state of flux.  If you are hearing both WWV and WWVH, don't try this as the two frequencies and signal strength will not likely be consistent and the results will probably be confusing.

If you don't happen to live in an area where you have a reasonable signal from WWV or WWVH then I suggest you ask around to find someone who has appropriate gear to help with this task.

Comments about alternative schemes for low-dropout regulation for the OCXO:

There are a number of "low-dropout" adjustable regulator ICs on the market that may be suitable for your this project - but there are a few caveats.

For example, there is the Linear Technologies LT1086-Adj which is rated for up to 1.5 amps of current.  While lower dropout than a conventional adjustable regulator such as an LM317, it does have approximately 1 volt of dropout which means that if you set the OCXO's supply voltage to 11.0 volts - the minimum recommended in the OCXO's specification - your battery voltage must be at least 12.0 volts:  While this represents a lead-acid battery that mostly depleted it is likely that a small, but healthy, lead acid could drop to such a voltage under transmit load - particularly if the resistance of power leads is taken into account.  This 3-terminal regulator is used in a manner very similar to the LM317 - except that you really must have some good quality, low-ESR capacitors (probably tantalum) very close to the regulator itself - see the data sheet.

Also made by Linear Technologies is the LT1528 that is rated for up to 3 amps that has a (nominal) 0.6 volts of dropout - more typically in the 0.3 to 0.5 volt area for the amount of current consumed by the OCXO, particularly once it has warmed up:  This extra margin would keep one in the "safe" region of the OCXO's operating voltage range down to around 11.5 volts from the batter allowing both "deeper" discharge and more voltage drop on connecting wires.  This part is somewhat more complicated to use than the LT1086, above, but it is, overall, simpler than the op-amp based regulator described earlier in this page.

If the "fast warmup" were to be implemented on either of the above regulators it would take a different form than the above - likely using several resistors and a transistor or two to "switch" the resistor-programmed voltage setting to something higher than the normal voltage.

There are a number of other, similar, low-dropout regulators that are made by different manufacturers, but very few have as low a dropout voltage (e.g. about 50 millivolts) as the simple FET/Op-amp circuit described on this page.

Additional comments:

• It is recommended that one not use a switching regulator to power the OCXO unless it has been extremely well filtered and bypassed.  Unless such a regulator is a buck-boost type it will probably have a higher drop-out voltage than even a standard low-dropout linear regulator.  Because of the rather low overhead voltage involved, there is not much loss in the linear regulator - only 10-15% or so with a 12.5 volt supply with a 11.0 volt output - a loss comparable to a garden-variety switching regulator.
  

• If you are interested in an example of this project being built with an etched PC board with surface-mount parts, visit VK4ABC's 10 MHz OCXO Web Page.

* * *

This is a revised version of one of my web pages, the original being found at http://www.ka7oei.com/10gig/10meg_oven_1.html


[END]

This page stolen from ka7oei.blogspot.com

 Note:  This post is partially an attempt to test means of reducing the "scraping" of content of this blog by sites such as "rssing", who seem to "swipe" content and "load" search engines' result with unwary readers NOT ending up at my page.     xe2XV6SJ9914C50H08S8  QY2IU7TU0C11c57804Q8

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