Measuring signal dynamics of the RX-888 (Mk2)

As a sort of follow-up to the previous posting about the RX-888 (Mk2) I decided to make some measurements to help characterize the gain and attenuation settings.

The RX-888 (Mk2) has two mechanisms for adjusting gain and attenuation:

• The PE4312 attenuator.  This is (more or less) right at the HF antenna input and it can be adjusted to provide up to 31.5dB of attenuation in 0.5dB steps.
  

• The AD8370 PGA.  This PGA (Programmable Gain Amplifier) can be adjusted to provide a "gain" from -11dB to about 34dB.

Note:

While this blog posting has specific numbers related to the RX-888 (Mk2), its general principles apply to ALL receivers - particularly those operating as "Direct Sampling" HF receivers.  A few examples of other receivers in this category include the KiwiSDR and Red Pitaya - to name but two.

Other article RX-888 articles:

RX-888 Thermal issues:  I recently posted another article about the RX-888 (Mk2) discussing the thermal properties of its mechanical construction - and ways to improve it to maximize reliability and durability.  You can find that article here:  Improving the thermal management of the RX-888 (Mk2) - link. 

Using an external clock with the RX-888:  The 27 MHz external clock input to the RX-888 is both fragile and fickle.  To learn a bit more about how to reliably clock an RX-888 from an external source, read THIS article.

* * * * *

Taking measurements

To ascertain the signal path properties of an RX-888 (Mk2) I set its sample rate to 64 Msps and using both the "HDSDR" and "SDR Radio" programs (under Windows - because it was convenient) and a a known-accurate signal generator (Schlumberger Si4031) I made measurements at 17 MHz which follow:

Gain setting (dB)	Noise floor (dBm/Hz)	Noise floor (dBm in 500Hz)	Apparent Clipping level (dBm)
-25	-106	-79	>+13dBm
+0	-140	-113	+3
+10	-151	-124	-8
+20	-155	-128	-18
+25	-157	-130	-23
+33	-158	-131	-31

Figure 1:  Measured performance of an RX-888 Mk2.  Gain mode is "high" with 0dB attenuation selected.

For convenience, the noise floor is shown both in "dBm/Hz" and in dBm in a 500 Hz bandwidth - which matches the scaling used in the chart below.  As the programs that I used have no direct indication of A/D converter clipping, I determined the "apparent" clipping level by noting the amplitude at which one additional dB of input power caused the sudden appearance of spurious signals.  Spot-checking indicated that the measured values at 17 and 30 MHz were within 1 dB of each other on the unit being tested.

Determining the right amount of "gain"

It should be stated at the outset that most of the available range of gain and attenuation provided by the RX-888's PE4312 step attenuator and AD8370 variable gain amplifier are completely useless to us.  To illustrate this point, let's consider a few examples.

Consider the chart below:

Figure 2:  ITU chart showing various noise environments versus frequency.

This chart - from the ITU - shows predicted noise floor levels - in a 500 Hz bandwidth - that may be expected at different frequencies in different locations.  Anecdotally, it is likely that in these days of proliferating switch-mode power supplies that we really need another line drawn above the top "Residential" curve, but let's be a bit optimistic and presume that it still holds true these days.

Let us consider the first entry in Figure 1 showing the gain setting of 0dB.  If we look at the "Residential" chart, above, we see that the curve at 30 MHz indicates a value very close to the -113dBm value in the "dBm in 500 Hz" column.  This tells us several things:

• Marginal sensitivity.  Because the noise floor of the RX-888 (Mk2) and that of our hypothetical RF environment are very close to each other, we may not be able to "hear" our noise floor at 30 MHz (e.g. the 10 meter amateur band).  One would need to do an "antenna versus no antenna" check of the S-meter/receiver to determine if the former causes an increase in signal level:  If not, additional gain may be needed to be able to hear signals that are at the noise floor.
  

• More gain may not help.  If we do perform the "antenna versus no antenna" test and see that with the antenna connected we get, say, an extra S-unit (6dB) of noise, we can conclude that under those conditions that more gain will not help in absolute system sensitivity.
  

Thinking about the above two statements a bit more, we can infer several important points about operating this or any receiver in a given receive environment:

• If we can already "hear" the noise floor, more gain won't help.  In this situation, adding more gain would be akin to listening to a weak and noisy signal and expecting that increasing the volume would cause the signal to get louder - but not the noise.  
  

• More gain than necessary will reduce the ability of the receiver to handle strong signals.  The HF environment is prone to wild fluctuations and signals can go between well below the local noise floor and very strong, so having any more gain that you need to hear your local noise floor is simply wasteful of the receiver's signal handling capability.  This fact is arguably more important with wide-band, direct-sampling receivers where the entire HF spectrum impinges on the analog-to-digital converter rather than a narrow section of a specific amateur band as is the case in "conventional" analog receivers.

Let us now consider what might happen if we were to place the same receiver in an ideal, quiet location - in this case, let's look at the "quiet rural" (bottom line) on the chart in Figure 2.

Again looking at the value at 30 MHz, we see that our line is now at about -133dBm (in 500 Hz) - but if we have our RX-888 gain set at 0 dB, we are now ((-133) - (-113) = ) 20 dB below the noise floor.  What this means is that a weak signal - just at the noise floor - is more than 3 S-units below the receiver sensitivity.  This also means that a receiver that may have been considered to be "Okay" in a noisy, urban environment will be quite "deaf" if it is relocated to a quiet one.

In this case we might think that we would simply increase our gain from 0 dB to +33dB - but you'll notice that even at that setting, the sensitivity will be only -131dBm in 500 Hz - still a few dB short of being able to hear the noise in our "antenna versus no antenna" test.

Too much gain is worse than too little!

At this point I refer to the far-right column in Figure 1 that shows the clipping level:  With a gain setting of +33dBm, we see that the RX-888 (Mk2) will overload at a signal level of around -31dBm - which translates to a  signal with a strength a bit higher than "S9 + 40dB".  While this sound like a strong signal, remember that this signal level is the cumulative TOTAL of ALL signals that enter the antenna port.  Thinking of it another way, this is the same as ten "S9+30dB" signals or one hundred "S9+20dB" signals - and when the bands are "open," there will be many times when this "-31dBm" signal level is exceeded from strong shortwave broadcast signals and lightning static.

In the case of too-little gain, only the weakest signals, below the receiver's noise floor will be affected - but if the A/D converter in the receiver is overloaded, ALL signals - weak or strong - are potentially disrupted as the converter no longer provides a faithful representation of the applied signal.  When the overload source is one or more strong transmissions, a melange of all signals present is smeared throughout the receive spectrum consisting of many mixing products, but if the overload is a static crash, the entire receive spectrum can be blanked out in a burst of noise - even at frequencies well removed from the original source of static.

Most of the adjustment range is useless!

Looking carefully at Figure 1 at the "noise floor" columns, you may notice something else:  Going from a gain of 0 dB to 10 dB, the noise floor "improves" (is lower) by about the same amount - but if you go from 25 dB gain to 33 dB gain we see that our noise floor improves by only 1 dB - but our overload threshold changes by the same eight dB as our gain increase.

What we can determine from this is that for practical purposes, any gain setting above 20 dB will result in a very little receiver sensitivity improvement while causing a dramatic reducing in the ability of the receiver to handle strong signals.

Based on our earlier analysis in a noise "Urban" environment, we can also determine that a gain setting lower than 0 dB will also make our receiver too-insensitive to hear the weakest signals:  The gain setting of -25dB shown in Figure 1 with a receive noise floor of -79dBm (500 Hz) - which is about S8 - is an extreme example of this.

Up to this point we have not paid any attention to the PE4312 attenuator as all measurements were taken with this set to minimum.  The reason for this is quite simple:  The noise figure (which translates to the absolute sensitivity of a receiver system) is determined by the noise generation of all of the components.  As reason dictates, if you have some gain in the signal path, the noise contribution of the devices after the gain have lesser effects - but any loss or noise contribution prior to the gain will directly increase the noise figure.

Note:

For examples of typical HF noise figure values, see the following articles:

• Measurements on a Multiband R2P Pro Low-Noise Amplifier System - link to the Web Archive.  Refer to page 9 section 7.
• Antenna and Receiver Noise Figure - link.
  

Based on the articles referenced above, having a receiver system with a noise figure of around 15dB is the maximum that will likely permit reception at the noise floor of a quiet 10 meter location.  If you aren't familiar with the effects of noise figure - and loss - in a receive signal path, it's worth playing with a tool like the Pasternack Enterprises Cascaded Noise Figure Calculator (link) to get a "feel" of the effects.

I do not have the ability to measure the precise noise figure of the RX-888 (Mk2) - and if I did do so, I would have to make such a measurement using the same variety of configurations depicted in Figure 1 - but we can know some parameters about the worst-case:

• Bias-Tee:  Estimated insertion loss of 1dB
  
• PE4312:  Insertion loss of 1.5dB at minimum attenuation
• RF Switch (HF/VHF):  1dB loss
  
• 50-200 Ohm transformer:  1dB loss
  
• AD8370 Noise figure:  8dB (at gain of 20dB)

The above sets the minimum HF floor noise figure of the RX-888 (Mk2) at about 12.5dB with an AD8370 gain setting of 20dB - but this does not include the noise figure of the A/D converter itself - which would be difficult to measure using conventional means.

On important aspect about system noise figure is that once you have loss in a system, you cannot recover sensitivity - no matter how much gain or how quiet your amplifier may be!  For example, if you have a "perfect" 20 dB gain amplifier with zero noise, if you place a 10 dB attenuator in front of it, you have just turned it into an amplifier with 10 dB noise figure with 10dB gain and there is nothing that can be done to improve it - other than get rid of the loss in front of the amplifier.

Similarly, if we take the same "perfect" amplifier - with 20dB of gain - and then cascade it with a receiver with a 20dB noise figure, the calculator linked above tells us that we now have a system noise figure of 3 dB since even with 20dB preceeding it, our receiver still contributes noise!

If we presume that the LTC2208 A/D converter in the RX-888 has a noise figure of 40dB and no gain (a "ballpark" value assuming an LSB of 10 microvolts - a value that probably doesn't reflect reality) our receive system will therefore have a noise figure of about 22dB.

What this means is that in most of the ways that matter, the PE4312 attenuator is not really very useful when the RX-888 (Mk2) is being used for reception of signal across the HF spectrum, in a relatively quiet location on an antenna system with no additional gain.

Where is the attenuator useful?

From the above, you might be asking under what conditions would the built-in PE4312 attenuator actually be useful?  There are two instances where this may be the case - and this would be applied ONLY if you have been unable to resolve overload situations by setting the gain of the AD8370 lower.

• In a receive signal path with a LOT of amplification.  If your receive signal path has - say - 30dB of amplification (and if it does, you might ask yourself "why?") a moderate amount of attenuation might be helpful.
  

• In a situation where there are some extremely strong signals present.  If you are near a shortwave or mediumwave (AM broadcast) transmitter that induces extremely strong signals in the receiver that cause intractable overload, the temporary use of attenuation may prevent the receiver from becoming overloaded to the point of being useless - but such attenuation will likely cause the complete loss of weaker signals.  In such a situation, the use of directional antennas and/or frequency-specific filtering should be strongly considered!
  

Improving sensitivity

Returning to an earlier example - our "Quiet Rural" receive site - we observed that even with the gain setting of the RX-888 (Mk2) at maximum, we would still not be able to hear our local noise floor at 30 MHz - so what can be done about this?

Let us build on what we have already determined:

• While sensitivities is slightly improved with higher gain values, setting the gain above 20dB offers little benefit while increasing the likelihood of overload.
  

• In a "Quiet Rural" situation, our 30 MHz noise floor is about -133dBm (500 Hz BW) which means that our receiver needs to attain a lower noise floor than this:  Let's presume that -136dBm (a value that is likely marginal) is a reasonable compromise.

With a "gain" setting of 20dB we know that our noise floor will be around -128dBm (500 Hz) and we need to improve this by about 8 dB.  For straw-man purposes, let's presume that the RX-888 (Mk2) at a gain setting of 20dB has a noise figure of 25dB, so let's see what it takes for an amplifier that precedes the RX-888 (Mk2) to lower than to 17dB or so using the Pasternak calculator above:

• 10dB LNA with 7 dB noise figure:  This would result in a system noise figure of about 16 dB - which should do the trick.

Again, the above presumes that there is NO  loss (cable, splitters, filtering) preceding the preamplifier.  Again, the presumed noise figure of 25dB for the RX-888 (Mk2) at a gain setting of 20 is a bit of a "SWAG"  - but it illustrates the issue.

Adding a low-noise external amplifier also has another side-effect:  By itself, with a gain setting of +33, the RX-888 (Mk2)'s overload point is -31dBm, but if we reduce the gain of the RX-888 to 20dB the overload drops to -18dBm - but adding the external 10dB gain amplifier will effectively reduce the overload to -28dBm, but this is still 5 dB better than if we had turned the RX-888's gain all of the way up!

Taking this a bit further, let's presume that we use, instead, an amplifier with 3dB noise figure and 8 dB gain:  Our system noise figure is now about 17dB, but our overload point is now -26dBm - even better!

The RX-888 is connected to a (noisy) computer!

Adding appropriate amounts of external gain has an additional effect:  The RX-888 (and all other SDRs) are computer/network connected devices with the potential of ingress of stray signals from connected devices (computers, network switches, power supplies, etc.).  The use of external amplifiers can help override (and submerge) such signals and if proper care is taken to choose the amount of gain of the external amplification and properly choose gain/attenuation settings within the receiver, superior performance in terms of sensitivity and signal-handling capability can be the result.

Additional filtering

Only mentioned in passing, running a wideband, direct-sampling receiver of ANY type (be it RX-888, KiwiSDR, Red Pitaya, etc.) connected to an antenna is asking a lot of even 16 bits of conversion!  If you happen to be in a rather noisy, urban location, the situation is a bit better in the sense that you can reduce receiver gain and still hear "everything there is to hear" - but if you have a very quiet location that requires extra gain, the same, strong signals that you were hearing in the noisy environment are just as strong in the quiet environment.

Here are a few suggestions for maximizing performance under the widest variety of situations:

• Add filtering for ranges that you do not plan to cover.  In most cases, AM band (mediumwave) coverage is not needed and may be filtered out.  Similarly, it is prudent to remove signals above that in which you are interested.  For the RX-888 (Mk2), if you run its sampling rate at just 65 MHz or so, you should install a 30 MHz low-pass filter to keep VHF and FM broadcast signals out.
  

• Add "window" filtering for bands of interest.  If you are interested only in amateur radio bands, there are a lot of very strong signals outside the bands of interest that will contribute to overload of the A/D converter.  It is possible to construct a set of filters that will pass only the bands of interest - but this does not (yet?) seem to be a commercial product.  (Such a product may be available in the near future - keep a lookout here for updates.)
  

• Add a "shelving" filter.  If you examine the graph in Figure 2 you will notice that as you go lower in frequency, the noise floor goes UP.  What this means is that at lower frequencies, you need less receiver sensitivity to hear the signals that are present - and it also means that if you increasingly attenuate those lower frequencies, you can remove a significant amount of RF energy from your receiver without actually reducing the absolute sensitivity.  A device that does just this is described in a previous blog article "Revisiting the limited-attenuation high-pass filter - again (link)".  While I do not offer such a filter personally, such a device - along with an integrated 30 MHz low-pass filter - may be found at Turn Island Systems - HERE.
  

Conclusions:

• The best HF weak-signal performance for the RX-888 (Mk2) will occur with the receiver configured for "High" gain mode, 0 dB attenuation and a gain setting of about 20dB.  Having said this, you should always to the "antenna versus no antenna" test:  If you see more than 6-10dB increase in the noise level at the quietest frequency, you probably have too much gain.  Conversely, if you don't see/hear a difference, you probably need more gain - taking care in doing so.
  

• For best HF performance of this - or any other wideband, direct-sampling HF SDR (RX-888, KiwiSDR, Red Pitaya, etc.) additional filtering is suggested - particularly the "shelving" filter described above.

• In situations where the noise floor is very low (e.g. a nice, receive quiet location) many direct-sampling SDRs (RX-888, KiwiSDR, Red Pitaya) will likely need additional gain to "hear" the weaker signals - particularly on the higher HF bands.  While some of these receivers offer onboard gain adjustment, the use of external high-performance (low-noise) amplification (along with filtering and careful adjustment of the devices' gain adjustments) will give improved absolute sensitivity while helping to preserve large-signal handling capability.
  

• Because the RX-888 is a computer-connected device, there will be ingress of undesired signals from the computer and the '888's built-in circuitry.  The use of external amplification - along with appropriate decoupling (e.g. common-mode chokes on the USB cable and connecting coaxial cables) can minimize the appearance of these signals.
  

 

This page was stolen from ka7oei.blogspot.com.

[End]

 

RF Quieting a Samlex 150 watt Sine Wave inverter

A few weeks ago I was on vacation in remote Eastern Utah - in Canyonlands National Park, to be precise and because we had some "down time" in the evenings, after hiking, after sunset, I was able to set up a portable HF station.  Using the homebrew end-fed halfwave antenna (EFHW) of Mike, K7DOU - one end of the rope tied around a rock laying on a shelf of slick rock some 40 feet above ground level and the other end tied to a bamboo pole attached to my Jeep - I connected my FT-100 through a manual tuner as the VSWR of the EFHW wasn't necessarily very low on some of the higher bands.

Figure 1:
150 Watt Samlex sine wave inverter, sitting on the workbench.
Click on the image for a larger version.

For whatever reason, I had brought along my old lap top and sound-card interface so I could work some digital modes, specifically FT-8 - a mode that I was familiar with, but had personally never worked.  The battery in my laptop had discharged, so I needed an alternate source of power and I connected my 150 watt Samlex Sine Wave inverter (a PST-15S-12A) to the battery to power the computer's power supply.

The (expected!) result of this was a tremendous "hash" all across the HF spectrum - an obvious result of the various high-power converters contained within the inverter.  On some bands the interference wasn't too bad, but on others the result was unusable.  While the battery charged, I operated on the band (20 meters, IIRC) that wasn't as badly affected.

I left the inverter running and the laptop battery charging during the cooking and eating of dinner, and with a reasonable amount of power banked I could turn off the inverter and get a zero noise floor while operating.

Why so noisy?

Modern AC inverters first convert the DC input power to something around the peak voltage found on the AC output - typically around 155 volts for 120 volt mains.  This conversion is done using a switch-mode inverter with a transformer, typically operating in the 20-60 kHz range and this output is rather rich in harmonics.

For the less-expensive "modified sine wave" inverters, the DC output is chopped, typically using an "H" bridge switch using FETs (Field Effect Transistors) with the duty cycle being varied to provide the equivalent of a 120 volt sine wave - and this switching can also add a bit of extra RFI, most notably in the form of a "buzz" - but this action produces less energy at radio frequencies than the initial voltage conversion.

The "Sine Wave" inverters perform the same step of producing the high DC voltage, but will chop the output into much smaller bits.  The method that this is done can vary, but it's sometimes done by using a "buck" type switching converter to transform the higher voltage into a varying - usually lower - voltage to simulate a sine wave on the output.  This second conversion adds yet another source of RF interference atop what is likely already the significant source that already present in the high voltage converter.

Comment:  The power converter (wall wart) that I was using to charge my laptop is particularly quiet, so I did verify that the vast majority of noise was, in fact, from the AC inverter.

Figure 2:
Various mains filtering components:  All of these are bifilar,
common-mode chokes, except for that in the upper-left with is
a combination filter and IEC power connector.
Click on the image for a larger version.

Quieting the inverter:

Fortunately, the internal space of this inverter wasn't terribly cramped so there was just enough room to add the necessary components to suppress the RF "hash" that was being conveyed on both the DC and AC lines.  While the methods of doing this sort of RF quieting have been discussed in previous blog posts (see the references at the end of this article) I'll review them in detail here.

Snap-on chokes won't do!

It's worth noting (several times!) that simply winding the power cord (DC and/or AC) around a ferrite device (e.g. a clamp-on or even a large toroid) would likely NOT be enough to solve this problem.  While doing so may knock down RFI by, perhaps, 6-10 dB - maybe 20 dB if one is really lucky - this sort of noise egress must often be attenuated by several 10s of dB to effectively quash it.  In other words, knocking down the "grunge" by 1-2 S-units is nice enough, but there will still be a lot of hash left over to bury the weakest signals! 

Internally, this inverter did pass through some rather large ferrite cylinders the DC input and (separately) AC output connections, but this very small amount of inductance would have practically no effect at all at HF - likely having been added to make a dent in the noise at VHF so that it would pass muster when subjected to EMC compliance tests.

Filtering the AC output:

I presumed (but didn't actually measure) that the majority of the noise being radiated would be from the AC output as it is "closest" to the circuits most likely to generate a lot of noise, so I concentrated most of my effort there.

The most helpful component in filtering the mains voltage output is the bifilar choke - several varieties of these being displayed in Figure 2.  This component consists of two windings in parallel on the same ferrite core - typically both leads of the mains voltage.  For the low-frequency AC currents, the halves of the choke carry equal and opposite current so there is no DC component to magnetize the core and reduce its efficacy due to saturation, but because RF energy is likely not flowing in a differential manner as is the AC mains voltage, the inductance of the two parallel windings come into effect - the magnitude of this typically being in the 10s of microHenries to milliHenries range.

Where does one get these things?  They can be found at surplus outlets if you look around, but perhaps the easiest source is from defunct PC power supplies:  These devices, found in supplies made by reputable manufacturers, are typically the first things through which the AC mains voltage pass (after any fusing) before going to the rest of the circuitry.

Figure 3:
Schematic of the output filter.  While it's likely that just one bifilar inductor would have sufficed, I decided that since there was room to do so, a second one would be added for even more filtering of the "grunge" that can emanate from such a noisy circuit.
Click on the image for a larger version.

 

This much inductance has significant impedance to RF energy - but inductance alone will have only limited efficacy and intrinsic capacitance of the windings will also reduce the amount of attenuation that would otherwise happen - as would have winding the mains cord/cable on a ferrite toroidal core as noted previously - so capacitors are also required to be placed strategically to help shunt away some of the residue.

Figure 4:
The AC output filter in the process of being installed.  L1 and
C1-C4 are mounted to the outlet itself while the connection
to L2 is made using the orange leads.
Click on the image for a larger version.

The diagram in Figure 3 shows the as-installed filter.  As can be seen, two separate bifilar filters (both of them being the sort as seen as the second from the lower-right in Figure 2) were used to maximize attenuation.  In this circuit, C3 and C4 are used to force any RF on the two wires to be common-mode to maximize the efficacy of the bifilar chokes' attenuation and any residual RF - which will be at rather low level and high impedance - will then be shunted to the metal case of the inverter by capacitors C1 and C2.

Figure 4 shows the installation of the filtering components in the inverter.  C1 and C2 are the disk-shaped blue capacitors seen in the upper-left, mounted directly to the inverter's single AC outlet and capacitor C3 is just in "front" of the two round disks, also mounted directly to the socket.  The first inductor, L1, can be seen in the shadows, connected to the outlet with very short, flexible leads to the plug.

Earlier, I had removed this outlet from the body of the inverter and mounted C1, C2, C3 and L1 to it and with a bit of "tetris" action, was able to reinstall the outlet back in place with the components attached.  From that point I installed C4 (to the "other" side of L1) and the (orange) connecting wires from C4 to L2, which is shown floating in space.

You might ask why there isn't another capacitor (like C4) across the "inverter" side of L2 - or other capacitors to ground other than C1/C2:  There is already a degree of filtering on the AC output of the inverter, so there is little point in adding another capacitor like C4.  As for other capacitors to "ground" like C1/C2 elsewhere in the circuitry:  These were deemed unnecessary - and doing so, particularly at the "inverter" side of L4 would simply put relatively strong RF currents onto the ground lead (e.g. inverter's case) - and our cause won't be helped in making RF currents appear where don't need them to be.  

Figure 5:
Noise filter on the DC input.  It looks suspiciously like the filter on the AC output - because it's the same type, although the current-carrying capacity of L1 is much higher and the values of the capacitors are orders of magnitude larger.
Click on the image for a larger version.

Filtering the DC input:

While I would presume that most of the noise would be emitted via the AC output port, filtering the DC port must be considered as well.  With the inverter's rating being 150 watts, the maximum current on the AC output would be around 1.25 amps and rather light-gauge wire could be used in the inductors - but because this same power level represents 12.5 amps at 12 volts (likely more if the battery voltage is on the low side) the filtering inductance must be made using much larger wire.

Rummaging around in my box of toroids, I found a ferrite device that was about 1" (2.54cm) in outside diameter and wound as many turns of 14 AWG flexible wire onto it as would fit (about 6 bifilar turns) and measured it to have about 30 uH of inductance per winding.  This may not seem like much, but at 1 MHz, this represents about 180 ohms of reactance.   

In referring to Figure 5, above, you'll notice that it is pretty much identical to that of the output filter - except that there is only one section of filtering.  The capacitor values are different, too:  C1 and C2 are 0.1uF units that shunt residual RF getting through L1 to ground (the case) while C3 is a low-ESR electrolytic connected across the DC leads to help force any residual AC noise on the DC lead to common-mode.  Compared to the 180 ohms of reactance of the DC bifilar choke (at 1 MHz) a good-quality, monolithic ceramic capacitor like the 0.1uF units are likely to have well under an ohm of impedance and very little of the RF hash will remain after they do their job to bypass it to the chassis ground.

Figure 6:
The DC input filter.  The capacitors (not visible) are mounted
to the bottom side of the terminal strip, which serves as the
RF "grounding" point to the case.  L1 is just visible.
Click on the image for a larger version.

Because of the limited amount of room, only one inductor was used - although it would likely be possible to have crammed another in the limited space should the above filter have proved to be inadequate (it wasn't).

As can be seen in Figure 6, a small terminal strip is visible and to it is mounted C1-C3 (not visible as they are obscured by the strip itself).  The mounting point for this strip is the ground lug near the DC input cable and the center lug is the common point for C1 and C2.

An important point to mention is the fact that this inverter - like many - have their DC and AC lines isolated from the case - and that's also important here:  Because the DC has no connection to the inverter's metal case, ALL of the DC current passes through L1 of Figure 5 - but with both halves carrying the same current, the core is not magnetized:  Magnetizing the core would likely cause it to saturate and the result would be its effective inductance plummeting - possibly reducing its efficacy as an RF filter.  It is for this reason that a bifilar choke was used on the DC input as well.

As with the AC output, the "inverter" side of L1 of Figure 5 also lacks a common-mode capacitor, but this is well represented on the input of the inverter itself with its own, built-in capacitor.

Figure 7:
The final arrangement of the added filtering components.  Liberal use of RTV (silicone adhesive) was used to stabilize the components as it works well, and can be removed should repairs/modifications be required.  On the left, a generous blob of RTV has been used to keep the terminal strip's lugs at the DC input from touching the inverter's bottom cover.
Click on the image for a larger version.

Additional comments:

Figure 7 shows the final arrangement of the added components.  In the upper-left corner can be seen the components of the DC input filter with come clear RTV (silicone adhesive) added to the top of the terminal strip to insulate it and keep any metal parts of it from touching the bottom cover when it was reinstalled.

On the right side is the AC output filter and on the foreground can be seen L2, now with the "hot" terminals covered by heat-shrink tubing.  This choke was first attached "temporarily" to the inverter's end plate using instant (cyanoacrylate) glue - and then several large blobs of RTV were later added to permanently hold it in place.  Just above it can be seen the orange wires that connect L2 to L1 and these components were also stabilized with rather large blobs of RTV to keep them from "flapping in the breeze".  It's worth noticing that the original ferrite cylinder is still on the AC output connection (on the black and white wires) where it connects to L4 - mainly because there was still room for it, and its efficacy, such as it is, is likely only enhanced by the addition of the new filtering components. 

Did it work?

You might ask the question:  Did this filtering work?

The answer is yes.  Placing a portable shortwave radio next to either the DC or AC power leads from the inverter, one can't detect that it is running at all.  If the radio is placed right atop the inverter, some hash can be detected, but this is likely from direct radiation of magnetic fields from the inductors/transformers within, but detectable amounts do not appear to be emanating from DC and AC wires themselves - and that's the important part as they would otherwise be acting as antennas.

Perhaps the most important part of this modification is the fact that any bypass capacitors are placed on the "quiet" (not the inverter) side of the filtering inductances and that these bypass capacitors are connected, with short leads, to a large, common-point ground - namely the case of the inverter.  If any of the "ground" leads had been more than an inch or two long, it's likely that the impedance of it would have reduced the efficacy of the filtering - but the case, being a solid chunk of extruded aluminum, forms a nice, low-impedance tie point - effectively a single-point ground, preventing an RF current differential between the DC input and AC output leads.

* * * 

Follow-up:

I've used this (modified) inverter several times, running strings of LED (Christmas) lights to illuminate the area around the camp site - often within a few feet (a meter or so) of the HF antenna and attached to the lower portion of the antenna mast and there was NO noise audible whatsoever from the inverter in the form of birdies from switching harmonics.

What was slightly audible was "space modulation" caused by the diode action of the LEDs themselves at the 60/120Hz AC frequency causing a small amount of amplitude modulation on received signals.  Interestingly, there was no audible noise generated by the LEDs themselves across the HF spectrum.

* * *

Links to other articles about power supply noise reduction found at ka7oei.blogspot.com:

• A high-current DC noise filter for UPS or RV use.  What if you have an inverter or UPS that is large - as in the multi-kilowatt class?  This article describes quieting one such device.
  

• Containing RF noise from a "pure" sine wave UPS.   Even when it is not operating your sine wave UPS may be producing a lot of HF radio interference!
  

• Completely containing switching power supply RFI - link.  Sometimes it can be difficult to quiet a switching power supply, so it may be necessary to put it in a box with strong filtering on all of the conductors that enter/leave.
  

• Minimizing VHF (and HF) RFI from electronic ballasts and fluorescent tubes - link.  Electronic light ballasts, like many switching power supplies, operate in the LF frequency range so "cleaning them up" at VLF/LF/MF frequencies can be a challenge.
  

• Quieting high current switching power supplies used in the shack - link.  This page describes techniques that can be used to reduce the amount of RF energy produced by switching power supplies that you may be using to power your radios.  Again, higher-inductance chokes may be required at VLF/LF/MF frequencies.
  

• Reducing switching supply racket - link.  This describes techniques that can be used to beef up the filtering for switching supplies in general.

 

This page stolen from ka7oei.blogspot.com

[End]

Characterizing the RTL-SDR Blog (Version 3) for HF reception using the "direct" input.

An inexpensive option for SDR (Software Defined Radio) reception on the HF (low frequency) bands is a device sold by "RTL-SDR Blog" - the current iteration being Version 3.  Originally intended for digital VHF/UHF TV reception - and that of FM broadcast - the hardware is also capable of tuning much lower frequencies.

Figure 1:  An RTL-SDR Blog V3 USB receiver "dongle".

Unlike most other inexpensive RTL-SDR dongles, this has - via a single SMA port - the ability to operate in "direct" mode where RF below the VHF frequencies is passed straight to the A/D converter rather than via a down-converter, allowing reception from (theoretically) a few hundred kHz to around 30 MHz.

How does it do this?

The typical RTL-SDR Dongle actually consists of two tunable devices:

• The Rafael R820T.  This is simply a frequency converter, capable of handling an input signal from somewhere below 60 MHz into the GHz range and converting it to what we'll call an "IF" (Intermediate Frequency) which is much will likely be below 12 MHz.  In addition to having a programmable oscillator and mixer for frequency conversion, his device has some built-in filtering that provides some protection to strong-ish off-frequency signals, and it has an AGC (Automatic Gain Control) that can adjust the level being output from it to prevent overload of the A/D converter as well as some front-end attenuation control to reduce the likelihood of overload on the input.
  

• The Realtek RTL2832U.  The down-converted output of the R820 chip is passed to this device, which consists of two 8 bit A/D (Analog-to-Digital) converters that are clocked at 28.8 MHz (meaning that signals above 14.4 MHz will be "aliased"), a USB interface, a (reported) 8051-type microcontroller and a digital frequency "converter" that is also capable of being "tuned" to produce a quadrature "baseband" signal that is output onto the USB port - the rate being programmable from around 250 ksps to 2880 ksps.  This device does NOT have an AGC or gain/attenuation control.
  

VHF/UHF operation:

Briefly, reception on the VHF/UHF frequencies is done the following way:

• A simple high pass filter/diplexer passes only the VHF/UHF signals (e.g. those above approximately 40 MHz) directly to the R820T.
  
• The filtering of the R820T is programmed for the desired characteristics at the operating frequency.
• The frequency converter in the R820T is offset from the input signal to provide an output (IF) signal in the 1-14 MHz range - more likely somewhere between 3 and 6 MHz.
• The level of this output signal may be automatically controlled by the AGC system of the R820T to keep it within the optimal range of the A/D converter.  Similarly, input gain adjustment on the R820T can prevent it from being overloaded by strong signals.
  
• The tuner within the RTL2832U is set to about the same frequency being output by the R820T (likely in the 5-12 MHz range) so that it is within the range of output ("baseband") sample rate, and output via the USB port.

HF operation:

• The signal to be received is applied via the RF antenna port and diverted from the R820T to a very simple low-pass filter/diplexer to an RF amplifier.  This diplexer's effect begins at approximately 22 MHz - See the discussion near table 1, below.
  
• The output of from the RF amplifier is passed through additional filtering and applied to one of the two A/D inputs of the RTL2832U - typically the "Q" input.
• The tuner within the RTL2832U is set near the frequency to be received so that it is within the range of the output ("baseband") sample rate, and output via the USB port.
• Because the sample rate of the RTL2832U is 28.8 MHz, any signal above 14.4 MHz will also appear "below" it as well - see the discussion below.
  

There's a penalty to pay for simplicity:

For those familiar with receiver topology - digital or analog - several things the above description of the HF operation of the RTL-SDR dongle without proper measures (e.g. filtering, gain control) should give cause for concern - namely:

• There is NO bandpass filtering at all.  Whatever is being intercepted by the antenna system will be input directly to the A/D converter via the preamplifier.
  

• ANY signal applied to the antenna input - no matter the frequency - can contribute to overload.  Because the inputted RF goes into the A/D converter, a strong signal well away from where you are listening can cause overload.  For example, if you are listening to 14 MHz, a strong AM broadcast signal (e.g. mediumwave) could well be the culprit if you are experiencing overload.
  

• There is no AGC in the HF ("direct") signal path.  Considering that the A/D converter is only 8 bits, this means that compared to even low-end shortwave receivers, the range of signal levels over which this device will operate is very narrow.  The lack of an "AGC" (automatic gain conttrol) means that there will likely not be enough gain when signals are very weak and overload of the A/D converter is likely if the signals are very strong.
  

• The A/D converter's sample rate is 28.8 MHz.  What this means is that the Nyquist limit - the frequency above which the digitized output can no longer represent the input signal - is half this, or 14.4 MHz, uncomfortably close to the 20 meter amateur band, and entirely below the 17, 15, 12 and 10 meter bands - and this does not even consider the fact that the sample frequency is within the 10 meter band itself. 

What this means is that there are unsuppressed image responses all across the HF spectrum.  For any frequency you tune below 14.4 MHz, you can also hear any signal above 14.4 MHz, the frequency of which is calculated by subtracting its frequency from the sample rate (e.g. 28.8 MHz).  For example, 15 meter signals will also appear, spectrally inverted (e.g. USB = LSB) 7.80-7.35 MHz.  Image frequencies appear in table 1 - along with MDS and clipping values - below.

In other words, if you are tuned to any HF frequency, you are actually "hearing" TWO frequencies at the same time unless you have specific filtering to prevent such image response.  See the right-hand column of Table 1 (below) for the frequency at which you will see an image.

The above puts strict limits on the performance of the RTL-SDR dongle as an HF receiver and these realities must be considered when the configuring any system that might use them.

The usable dynamic range:

In theory, 8 bits of A/D sampling would indicate about 48dB of useful signal range, but the reality is more complicated than this.  Compared to the bandwidth of the narrow signals typically sought on HF, the overall sample rate of the A/D converter - and even the rate after the RTL2832U's converter has reduced the signal to the sample rate being sent to the USB port - are very much higher, effectively improving the bit depth via oversampling - and the fact that the HF spectrum is backgrounded with what amounts to white noise works to our benefit, helping to spread discrete, spectral artifacts that are the inevitable result of the imperfect signal acquisition.

This also works the other way:  Very strong, discrete signals (e.g. SWBC transmitters) can also cause mixing products which can find their way in other signals - particularly weak ones - but this effect may be minimized if one manages to keep the input level "high enough" so that at least several bits of A/D conversion are involved and also limited such that A/D converter overloads are minimized as much as practical - a balancing act that makes it difficult to handle both weak and strong signals at the same time.

All of this makes the actual, usable range a bit difficult to divine.  To this end, an RTL-SDR (V3) unit was put on the workbench, using the "HDSDR" program as a receiver, and its operation was analyzed by observing CW (unmodulated) signals on narrow (SSB) bandwidths.

Methodology:

Ideally, we would be able to determine the RF level at which the A/D convert clipped directly, but the HDSDR program does not provide a means to see the peak A/D level, requiring us to infer it by noting the level from a known-accurate signal generator at which the S-meter starts to decrease at the same rate that the signal level is reduced - and also by noting the disappearance from the waterfall many of spurious signal caused by overload.  Typically, this is about 3dB below the "maximum" S-meter reading.

For sensitivity, DL4YHF's "Spectrum Lab" program was used to measure the SINAD in a 500 Hz bandwidth set by the HDSDR program, the "minimum discernible signal" being equivalent, in this test, to 3dB S/N.

Table 1:  Measured signal levels for A/D clipping and MDS, along with corresponding image frequencies on HF.  Because the sample rate of the RTL-SDR is 28.8 MHz, ALL signals above half this frequency (e.g. 14.4 MHz) are, by definition, Nyquist images.  The far-right column is "apparent" dynamic range - see the discussion below.

Frequency (MHz)	Clipping (dBm)	MDS (dBm)	Image  (MHz)	Apparent DR (dB)
0.5	-17	-97	28.3	80
1.0	-24	-107	27.8	83
2.0	-27	-109	26.8	82
4.0	-27	-116	24.8	89
7.0	-29	-117	21.8	88
10.0	-30	-117	18.8	87
14.0	-30	-117	14.8	87
15.0	-30	-116	13.8	86
18.0	-30	-116	10.8	86
21.0	-29	-114	7.8	85
24.0	-25	-109	4.8	84
28.0	-14	-95	0.8	81
29.0	-16	-94	0.2	78
35.0	-16	-90	6.2	74
40.0	-2	-84	11.2	82

Discussion:

Table 1 tells us several things:

1. As mentioned in the documentation found at the RTL-SDR Blog web site, signals below 2 MHz - and especially below 1 MHz - are rolled off by the "Bias Tee" blocking choke which has insufficient inductance at frequencies below the AM broadcast band.  To a degree, this effect can be mitigated by removal of the RF choke which will remove the capability to inject DC onto the cable.  This table includes the effect of this choke.
   
2. The RTL-SDR has a low-pass filter to diplex the HF and VHF and above frequencies to separate signal paths, and the effects of this filter are becoming evident above 15 meters (21 MHz).  This also means that by itself, the RTL-SDR becomes "deaf as a post" on the 12 and 10 meter bands.
   
3. Note that at the very low frequencies, the sensitivity appears to be somewhat reduced.  The effects of high-pass roll-off - likely caused by coupling capacitors and the input bias inductor - appear to be evident through at least 7 MHz.
4. There is clearly no image rejection at all, considering that the sensitivity through 21 MHz is comparable to that below 14 MHz.  For example, 15 MHz WWV will also appear on a receiver program at 13.8 MHz, at the same apparent signal strength.
   
5. Low-pass effects are evident by 24 MHz, most likely a result of the diplexer used to split the HF (direct) and VHF/UHF signal paths.  This limits sensitivity on the 12 and 10 meter amateur bands.
6. With only 8 bits of quantization, additional noise will be generated due to the imprecise nature of the process.  This "noise" will show up as spurious signals and, less obviously, as a rise in the overall noise floor - depending on the nature of what is being digitized.  In short, the fewer the number of bits, the less likely it is that weak signals will coexist (and be audible) in the presence of strong signals.
7. Across most of the HF spectrum, the RTL-SDR will overload signal at about -30dBm - which is approximately equal to an S-meter reading of "40 over S-9".  While this seems like a fairly strong signal, this power level represents the total amount of RF energy - no matter the frequency!
8. Note that the power at which the A/D converter starts to overload is approximately -30dBm across much of the HF spectrum.  This represents the TOTAL amount of RF power required, at all frequencies combined, that will result in overload.
9. Across much of the HF and MF spectrum (0.5-28 MHz) the apparent dynamic range is on the order of 80 dB.  This number should be taken with a grain of salt as it was measured in the absence of any other signals -  hardly a real-world equivalent.  Practically speaking, this number is indicative of the best possible performance under ideal conditions.
   

Points 7 and 8, above, should be considered very carefully in terms of its implications:

• You cannot simply connect an RTL-SDR Dongle to even an "average" performing HF antenna and expect reasonable results as the total power from ALL signals reaching the receiver are likely to exceed the -30dBm signal level - particularly if you have any AM (mediumwave) transmitters anywhere nearby (e.g. within 20 miles/30km).
  

• Particularly on the lower bands (80, 40, 30 meters) the signal levels of amateur and especially shortwave broadcast signals can, by themselves, exceed the -30dBm overload level - particularly in Europe and the eastern U.S. On some bands, the shortwave broadcast band adjacent to the amateur band (e.g. 41 and 40 meters) are too close to effectively filter out and it may be that an RTL-SDR is simply not usable on these bands when propagation favors reception on these frequencies. 
  

• As seen in Table 1 (above) the RTL-SDR dongle becomes increasingly deaf on the higher HF bands (particularly 12 and 10 meters) making them unusable at these frequencies without additional amplification AND filtering.  What's worse is that the lower HF frequencies (e.g. below 10 MHz) are typically very noisy while the higher frequencies are quiet by comparison.  If you connect an antenna to the RTL-SDR dongle with no band-pass filtering and try to tune in, say, the 15 meter band, you will likely hear only noise (and signals) around 4.8 MHz, which will likely overwhelm any weak, 15 meter signals.
  

• If you plan to use an RTL-SDR for HF reception and expect even mediocre performance, you should precede it with a band-pass filter for the frequency band of interest.  For the highest HF bands (15, 12 and 10 meters) the typical noise floor in a quiet location is around -120 dBm (in a 500 Hz bandwidth) - which is well below the noise floor of the RTL-SDR at these frequency meaning that a preamplifier (along with a bandpass filter) will be required for reasonable performance.
  

• It's worth remembering that unlike an analog receive system, one cannot always use all 8 bits for digitization:  The signal input must be kept well below the "full scale" level (at and above which "clipping" will occur, causing distortion and signal degradation everywhere else) to accommodate for random fluctuations that are ever-present on signals input from the antenna.  What this means is that the A/D must be under-driven overall and that fewer bits are actually being used most of the time.  In order to maintain suitable margin, it's typical to drive the A/D converter at between 1/4 and 1/2 full scale, meaning that for 8 bits of A/D conversion, 2-4 bits are typically being used - and fewer, still, when the band is "dead" and signals are weak.
  

Conclusions:

• DO NOT simply connect an RTL-SDR to your HF antenna and expect it to work as well as even a low-end shortwave receiver:  If it doesn't get overloaded by local AM broadcast (mediumwave) signals, it will get overloaded by strong shortwave broadcast signals when conditions are favorable on certain bands.  This isn't to say that you won't hear anything if you do so, but know that normal signal levels present on even an "average" antenna will be enough to overload the RTL-SDR dongle.
  

• ALWAYS precede an RTL-SDR with a band-pass filter that is specific for the frequency range of interest. For example, if you are interested in 40 meter reception (7.0-7.3 MHz for ITU region 2) your filter should pass only frequencies in this range, and this filtering will prevent unwanted reception of signals at the image frequency around 21.8 MHz.   Unfortunately, the use of a band-pass filter precludes reception outside its design range, but this is necessary considering the limited capability of the RTL-SDR dongle in terms of handling both strong and weak signals at the same time and its unfettered response to unwanted images.
  

• In some cases - even with a mediocre antenna - the signals in the desired frequency range may exceed the signal handling capability of the RTL-SDR and cause overload.  As noted above, a single signal stronger than -30dBm can do this, but so could a number of signals whose total power can exceed this.  Typically, this overload is manifest as intermittent distortion across the entire receiver as signals fade in and out.  In such cases, it might be beneficial to attenuate the signals reaching the RTL-SDR as the degradation caused by overload is more "destructive" across the entire receive frequency range than too-little signal.
  

• Because of image response and roll-off, the HF port of the RTL-SDR is really not well-suited for 12 and 10 meter (24-30 MHz) reception.

Comment about HF upconverters:

Specifically to address some of these issues, there are upconverters available for the RTL-SDR that will upconvert the HF spectrum to VHF (typically in the 100-130 MHz range) to allow the use of that signal path, making use of the Raphael R820T converter.  This converter has the advantage of having a degree of band-pass filtering and the ability to use AGC (automatic gain control) on the signal path.

This method can be useful, but there are several caveats:

• If frequency stability is of importance, the addition of the upconverter introduces two additional frequency stability issues:  Drift of the upconverter itself, and the fact that any existing drift in the dongle itself will be multiplied because of its operation at the higher frequency.  This can be an issue in environments where the temperature is not stable and/or when a frequency sensitive mode like SSB or (especially) digital modes are used.
  

• If multiple bands and receivers are to be used, there may be the temptation to upconvert the output of the upconverter to VHF and distribute this signal to the receivers.  While this may work in many cases, it's worth noting that if the entire HF spectrum is converted, the total signal power level can be significant, potentially overloading the upconverter itself, any RF amplifiers that might be used at VHF, and/or the front end of the RTL-SDR dongles themselves.
• While a "fix" might normally be to filter out just the HF band(s) of interest, this can become impractical if the 40 meter band (7.0-7.3 MHz) is upconverted to, say, 137 MHz where it can become difficult to make an effective band-pass filter at that frequency.
• It is possible to filter for a specific HF band before the upconversion, but this means that each RTL-SDR would require its very own upconverter.  While effective, the cost of an upconverter for each band may be prohibitive.

* * *

Successfully using an RTL-SDR on HF:

As mentioned earlier, one must precede the RTL-SDR with a band-pass filter to obtain reasonable performance on HF - or any other frequency range where very strong and very weak signals will be simultaneously present at the antenna input:  Remember that, especially in the "direct" mode, all signals applied to the RF input - even those MHz away from where the receiver is tuned - will count "against" you in terms of the total amount of RF power that may be applied to the A/D converter before it clips/overloads.

To see some RTL-SDR based HF receive systems in operation - and to be able to directly compare them with higher-performance receivers using the 16 bit signal path, visit the Northern Utah WebSDR site at sdrutah.org (link).  This will take you to a "landing page" where you can select several receivers - specifically:

• WebSDR #3:  This server uses RTL-SDRs for both the 80 and 40 meter bands.
  

• WebSDR #1:  This server uses SDRPlay RSP1a receivers which have both an RF AGC and around 14 bits of A/D converter depth - plus external band-pass filtering.
  

Both of these systems use the same antenna for 80 and 40 meters and it is possible to directly compare signals between the two in side-by-side windows.  Generally, the 8 bit RTL-SDRs used on WebSDR #3 hold their own compared to those on WebSDR #1, and this is possible only because the WebSDRs are preceded with both band-pass filtering and AGC as described in the link below.

* * *

Additional resources:

• Band-pass filters are readily available in kit form from QRP Labs http://qrp-labs.com/bpfkit.html  
  

• An article on using band-pass filtering and AGC (Automatic Gain Control) to improve the performance of an RTL-SDR when used for amateur band service may be found here:  http://www.sdrutah.org/info/rtl_sdr_agc.html .  Several "test" receivers are currently in operation at the Northern Utah WebSDR that demonstrate the efficacy of doing this.

 This page stolen from ka7oei.blogspot.com

[End]

Quieting an insanely (RFI) noisy LED floodlight

A friend of mine recently installed some inexpensive Chinese-made floodlights to illuminate his backyard, but was dismayed to discover that when they were on, his 80, 40 and 20 (shortwave - 3.5-14.5 MHz) reception "went away", replaced with a very strong noise that was "20 over" - a degradation of apparent sensitivity of much more than 20dB.  As it turned out, almost every frequency below and above this range he checked was also affected to a similar degree.

RF noise from "grow lights" - the same phenomenon Several years ago, there was some noise (pun intended) in the Amateur press about LED power supplies being sold that caused a tremendous amount of RF interference - and many of these stories also included anecdotes of many of these interference sources having been tracked down and found to have been "grow" operations.  Later, some stories surfaced where law enforcement officers were able to locate some of these "grow ops" simply by finding the source of RF interference. The LED power supply described on this page is of the same type that was found to cause these very high levels of RF interference.

Even though these lights aren't turned on very often, he decided that their flaws went firmly against his eternal crusade against RFI-generating devices at his house.  After all, when it comes to RF interference, one should remember this cardinal rule:

Most RFI begins at home!

To be sure, there are many cases in which there are noisy power lines or a neighbors plasma TV - just two in a long list of things that can cause interference, but the worst offenders in generating interference are likely in one's own house.   The main reasons for this are simple and (for the most part) obvious:

• They are nearby.  If a noise generating device is in your house, it's very close-by - and the closer it is, the more your antenna is likely to intercept "grunge" from that device.

• They are connected to the same wiring as everything else in your house.  There's nothing like a piece of copper to convey RF all over the place with minimal loss, and if a noise generator is powered from the mains, it's likely conducting much of that noise into the same mains connections that power your radios.

• If you are like most amateurs, you probably have radiating feedlines on your HF antennas.  By their very nature, almost all HF antennas tend to radiate a bit of RF on their feedlines.  For some antennas (e.g. dipoles, yagis, loops) this is incidental - often due to inadequate balun design, but other antennas (offset-fed antennas like Windoms, end-fed antennas) this is often by nature or design.  If the feedline of your HF antenna isn't very well-balanced (often using a "current mode" choke) some of your "noise" from the devices in your house wiring is being conducted from your shack, onto the feedline and then into your antenna.  Fixing this problem certainly warrants a series of articles itself, but suffice it to say, "noisy" devices will seem worse because of this issue than they would normally be.

Figure 1:
The constant-current LED driver with added filtering.  This LED driver
is typical of what is seen in these devices:  A rather generic, potted module
of likely-questionable lineage and quality.
Click on the image for a larger version.

What are these things?


As is typical with these inexpensive LED lamps, the power supply is a constant current module that uses PWM/switching techniques to regulate the current applied to the LED array to some value.  As can be seen Figure 1, this is simply a box with two sets of wires:  The AC (main) input on one side and the DC output to the LEDs on the other.

Because these are constant current supplies, they can be used over a wide range of LED module voltages:  22 to 36 volts, according to its label of that in Figure 1.  Noting the official "50 watt" power rating, we can do the math and see that with a constant 1500mA, the power being delivered to the LED array can vary from 33 to 54 watts, depending on its actual operating voltage.  Depending on the design, these supplies may or may not have their DC outputs isolated from the mains input via an internal transformer, so it is best to assume that they are not isolated and that the DC outputs will be line-referenced and hazardous (even lethal!) to touch.

In this example, the red and black DC leads disappeared into the body of the case where it would connect to an LED module that is (presumably!) insulated from the lamp's case.  Because you can't be sure what to expect, one must always make sure that the safety ground of these lamp housings is actually connected to the case (the ground wires in these devices are often not connected to the case at the factory!) and that it is plugged into a GFCI-protected outlet.

How bad was it?

In the case of these LED floodlights, the only connection that they had to the rest of the universe was via their power connections, so it was clearly via its power leads that they were radiating their "grunge".  To determine in some quantitative way how noisy this device was, a simple test fixture was constructed to measure the energy imparted on the mains power lead, represented schematically in Figure 2, below:

Figure 2:
Test fixture to analyze the amount of RF being conducted from the LED's current supply to its mains leads..
"Ca" and "Cb" are 0.1 to 0.47 "X" class "safety" capacitors used for mains filtering and "La" is a bifilar mains choke of at least 1 milliHenry per winding, these constituting a filter to decouple noise already present on the mains from the test fixture:  One of the filters depicted in Figure 5 could have been used for this purpose.
RF coupling transformer "Ta" consists of a Mix 31 clamp-on ferrite choke with a single wire going to the "Device Under Test" as the primary and 6-8 turns of smaller wire as the secondary to couple RF from it.  The box marked "protection" is simply two back-to-back 5.1 volt Zener diodes in series to protect the analyzer from voltage transients caused by turn on/off transients.
Click on the image for a larger version.

In this circuit we see a common-mode line filter using Ca, Cb and La forming a circuit to attenuate noise that might already be on the mains.  The goal is that when we measure RF noise via coupling transformer Ta, we are (mostly) seeing the noise from the device being tested and not that which may already happen to be on the mains.

The result of this measurement can be seen in Figure 3, below, covering the range from nearly DC to 1 GHz, with the cyan trace being with the unit turned off and the yellow trace with it turned on:

Figure 3:
Noise from the power supply as seen from 0 to 1 GHz.  The blue trace is with the LED power supply powered down while the yellow trance shows it powered up.  As can be seen, it is a potent noise generator well into the UHF spectrum - but particularly at and below 100 MHz!
The various signals on the cyan trace are off-air signals, including AM, FM and TV broadcast and 800 MHz - plus some leakage from the noisy mains through the Figure 2 filter:  Ingress of these signals is the inevitable consequence of the rather simple lash-up and not conducting these tests inside an RF-screened room!
Click on the image for a larger version.

While this test fixture isn't perfect (e.g. some leakage from the mains through the filter, some couple of broadcast signals directly into the fixture over the air and the fact that the coupling coefficient is unknown because I didn't bother to determine it!) it did the job of giving a relative indication of how much "grunge" the LED's power supply put into the mains - and this same information would later be useful to get a general idea as to how much our mitigation efforts reduced this noise.  As can be seen, below 100 MHz the added noise (in a 3 MHz detection bandwidth) is nearly 50dB (100000 x) higher than the noise floor of the analyzer and the test fixture.

Refocusing on a smaller frequency range with different analyzer settings, let's take another look at how bad it is over the lower HF range:

Figure 4:
A re-done plot over the range of 0-100 MHz, this time with an 8 MHz resolution bandwidth.  The higher resolution bandwidth results in a higher reading from the QRM generator as its output is broadband noise.
Over much of this range, the base noise level (in cyan) is below the measurement sensitivity of the analyzer.
Click on the image for a larger version.

From the plot in Figure 4 we can start to get a picture of how bad the situation really is.  As can be seen at Marker #1, we measured a power level of about -3dBm - or 0.5 milliwatts within an 8 MHz bandwidth, but if we were to integrate this energy over the entire 0-100 MHz range we can see that there may be, perhaps a couple of 10s of milliwatts of noise being coupled into the mains:  We can only guess at the true amount of conducted RF owing to the comparative crudity of our test fixture and its unknown coupling coefficient across the RF spectrum, but we can be reasonably sure that what we see on this trace is but a fraction of the total energy present.

Figure 5:
Some board-mountable Shaffner mains filters from the Electronic
Goldmine, item G21844 (no longer available - sorry...)
Click on the image for a larger version.

As noted earlier, the entire purpose of these measurements was not to determine an absolute level of RF energy, but rather to have a means of repeatably measuring how bad things are - and also to be able to determine if our mitigation methods are having the desired effect.

"Fixing" the problem:

One solution to this problem (aside from not getting cheap, uncertified devices in the first place - but even then, one is never sure what one is really buying!) is to add known-to-be-effective filtering to the mains leads.

At about the time my friend brought these lamps to me, I noticed that the Electronic Goldmine had, on sale, some small, board-mount mains filters, so I suggested that he buy at least two for each of his three lights (for a total of six) - so he bought 10 of them.  These particular devices were attractive because they were relatively inexpensive, potted (helpful, because this will be mounted outdoors where moisture ingress could be a problem) and small enough to fit in the limited-space enclosure in the back of the floodlight.  Being that the lamps were only "50 watt", the 1.6 amp rating of these filters would be more than adequate.

Figure 6:
Another view of added filtering and their integration into the enclosure.
The Shaffner filters were mounted "dead bug" (leads up) and held in place
using both the ground wires and silicone (RTV) sealing compound.  The
lug at the lower-right was added to help make sure that this plate was
electrically bonded to the main body of the LED floodlight.
Click on the image for a larger version.

As can be seen in Figure 1 and Figure 6, two of these filters were installed "back to back" in the back of the lamp housing, using direct-soldered connections between the ground terminals of the filters and the metal plate itself with short pieces of heavy (8 AWG) copper wire to keep the impedance of these leads as low as possible:  Even a few inches/centimeters extra was found to significantly reduce the efficacy of these filters at VHF and higher frequencies.

You may notice something else about the layout:  The wires going in and out of the LED driver are bundled together with plastic wire ties and routed to the "far" side of the power supply, as distant from the mains filters and wires as possible - this to minimize the amount of RF energy that might be coupled from these "noisy" wires into the power cord - something that would surely "un-do" some of our hard-won efforts in minimize the amount of conducted RF noise.

The result:

The results of this effort can be seen in Figure 7, below:

Figure 7:
"Before" and "After" traces over the 0-10 MHz range.  The cyan trance is with the LED unit powered down, the yellow trance is without filtering and the magenta trace is including filtering.  Note the lower resolution bandwidth (91kHz) as compared to the other figures which will tend to reduce the apparent level of broadband noise from LED driver and accentuate those of "coherent" signals such as broadcast stations.
The strong signals at about 1.0-1.4 MHz are due to ingress of local AM broadcast stations into the lashed-up test figure, the level exceeding that of the leakage through the filter.
Click on the image for a larger version.

In Figure 7, above, we can see multiple traces - with the explanation below:

• The Cyan (blue-ish) trace is our baseline measurement with the LED driver module powered down.  The signals below about 1.5 MHz are ingress from strong, nearby AM broadcast stations, some of which are nearly as strong as the noise at specific frequencies.
• The Yellow trace is with no filtering of the LED driver module, showing the relative energy from the LED driver module over the 0-10 MHz range.
• The Magenta (purple-ish) trace is with the LED driver module powered up with the added filtering.

"Could you have just snapped ferrites on the power cable?" In reading this article, one might wonder if we could have solved the problem simply by putting snap-on ferrites on the power cord. I doubt it. Snap-on ferrite devices are very good about reducing the amount of RF conducted on wire, but with the extreme nature of the interference of these devices, it would never have been enough at HF.  The reason for this is that in order to adequately quash the QRM to the "point of undetectability" it would take at least several k-ohms of impedance on the power cable to solve the problem. While it is possible that one can do this, it would take several large-ish cores (probably mix 31) with a dozen or more turns on each just to add that much reactance - but that material and winding topology would only work to the high end of the HF spectrum, so you'd need another core or two with windings on different materials - say 43 and 61 mix. To make matters worse, you'd have to keep these chokes well-separated physically or else RF energy would be conducted around them - or even radiate directly from this rather large structure:  You certainly wouldn't have been able to easily fit it in the back side of the lamp's enclosure. Self-contained filter modules like the ones used are specifically designed to quash RF over a very wide frequency range:  Not only are bifilar inductors used, but capacitors are also used to force the interfering energy to common mode so that the inductors can best do their job, plus there are other capacitors that do an excellent job of shunting RF to the case to "completely" contain that energy. In other words:  In such an extreme case, you'd be far better off using an L/C filter like that depicted in Figure 5 without even bothering with ferrite chokes.

Interpreting these results we can see that over much of this range that the filtering reduced the amount of conducted noise to just above that of the cyan line, knocking the noise down by roughly 20dB over the range.  These filters start to lose their effectiveness below 1 MHz which is why, at very low frequencies (below 500 kHz) one starts to see more conducted energy - but these frequencies don't radiate very well, anyway so they are of generally less concern in most amateur stations.

When this plot was taken, the circuit depicted in Figure 2 was very close to the filter networks and it was believed that some energy was directly coupling into it from the LED driver module.   After the lamp was assembled (the cover put on and the power cord fitted) another test was done and no difference at all could be seen in the "on" and "off" traces - except at frequencies below about 1.5 MHz:  I somehow managed to omit capturing this trace.

Did it help?

Yes!

My friend reinstalled these lights and was happy to report that upon listening on various HF bands from 160 through 10 meters, he was unable to detect when the lights were on or off, indicating that the modification was successful.  It is possible that within a few feet/meters of these lights that some low-level direct radiation of noise could have occurred on VHF/UHF frequencies, but this energy was demonstrably not being conducted via the power cord, and emissions would not likely be detectable more than a few feet/meters away, anyway.

Would just a single  filter have done the job?

Probably - but since the lights were a bit of a pain to take down and put up again it was decided to use two of these filters just to avoid the possible hassle of having to take them down (and apart) again if just one filter hadn't been enough! 

Follow-up:

It would seem that the statement above about questionable quality was justified:  In the two years since these were installed, only one of these lights continues to work, the other having failed in fairly quick succession.

The phrase "Caveat Emptor" comes to mind!

* * * * * * * * * * *

Links to other articles about power supply noise reduction found at ka7oei.blogspot.com:

• Containing RF noise from a "pure" sine wave UPS.   Even when it is not operating your sine wave UPS may be producing a lot of HF radio interference!
• Completely containing switching power supply RFI - link.  Sometimes it can be difficult to quiet a switching power supply, so it may be necessary to put it in a box with strong filtering on all of the conductors that enter/leave.
• Minimizing VHF (and HF) RFI from electronic ballasts and fluorescent tubes - link.  Electronic light ballasts, like many switching power supplies, operate in the LF frequency range so "cleaning them up" at VLF/LF/MF frequencies can be a challenge.
• Quieting high current switching power supplies used in the shack - link.  This page describes techniques that can be used to reduce the amount of RF energy produced by switching power supplies that you may be using to power your radios.  Again, higher-inductance chokes may be required at VLF/LF/MF frequencies.
• Reducing switching supply racket - link.  This describes techniques that can be used to beef up the filtering for switching supplies in general.
• Quieting a 150 watt Samlex sine wave inverter - link.  This page talks about making an RF-noisy sine wave inverter RF-quiet.
  

This page stolen from ka7oei.blogspot.com

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