Wednesday, November 6, 2019

Characterizing the Mini-Circuits ZFSC-4-3, ZFDC-20-3, ZFSC-4-1-BNC+ and ZFSC-2-1+ well below their designed frequency range

Figure 1:
The collection of devices to be tested - plus a few 50 ohm terminators.
Click on the image for a larger version.
Rummaging through a box of RF stuff I ran across several multi-port devices made by Mini-
Circuits Labs that I'd picked up over the years - typically at amateur radio swap meets.

The "official" specs of these devices are easy to find (at least for the newer "plus" versions) but what if, like me, one was interested in using them at frequencies below their official design specs - such as the lower amateur bands, including 80, 160, 630 and 2200 meters?  How much "extra" design margin was built into these device?

Wielding my DG8SAQ Vector Network Analyzer, I decided to find out.  For these measurements I limited the measurement range to between 10 kHz and 60 MHz.  I also built several homebrew versions to see if I could, for little cost, come up with suitable versions of my own - and these will be described in a follow-up article.

All of the devices described here could also be used to combine signals from multiple sources.  Unless the signals being combined are "phase coherent" (e.g. from the same signal source) the insertion loss will be the same as that in splitter operation.  They will be referred to only as "splitters" in this article to minimize clutter.

Figure 2:
 Insertion loss of the ZFSC-4-3 from 10 kHz to 60 MHz.
Even though the "official" low-frequency specification is 10 MHz,
it should be quite usable on 160 meters (down to at least 1.8 MHz).
Click on the image for a larger version.
ZFSC-4-3 four-way splitter:

This device, equipped with BNC connectors, has an "official" frequency range of 10-300 MHz (the currently-offered "plus" version has the same ratings), splits the signal 4 ways with a theoretical insertion loss of 6 dB, but practically speaking, the actual loss is rated as being closer to 6.4 dB over the lower end of the design range.  Although this device - and others below - are billed as a splitters, they may be used to combine disparate signals from multiple sources onto a single line with the same amount of insertion loss.

Figure 2 shows the measured insertion loss over the range of 10 kHz to 60 MHz.

Figure 3:
 Isolation of the ZFSC-4-3 from 10 kHz to 60 MHz between ports 1 and 2.
Click on the image for a larger version.
This shows us that down to about 1.8 MHz (160 meters) that the insertion loss (blue trace) is only slightly (0.15dB) higher than the rated specs - and the Smith chart (red trace) is also reasonably well-behaved.  The next marker to the left (#3) is placed at 100 kHz and we see that the insertion loss is closer to 9 dB and that the impedance has dropped to around 12 ohms - getting worse at 20 and 10 kHz where the losses are 20dB or more and the measured impedance is only a few ohms.

What this tells us is that this device is likely to be useful down to about 500 kHz, below which point the insertion loss and impedance mismatch start to become significant - likely due to the fact that the intrinsic impedance of the ferrite devices within the splitter has dropped too low at these frequencies to remain "transparent".

Figure 4:
 Isolation of the ZFSC-4-3 from 10 kHz to 60 MHz between ports 1 and 3.
Click on the image for a larger version.
Figure 3 shows the port-to-port isolation between ports 1 and 2 and the scene is similar:  The insertion loss curve is pretty flat to about 500 kHz where it starts to vary and much below 100 kHz, the isolation seems to increase, but this correlates to the insertion losses.

Figure 4 shows the port-to-port isolation between ports 1 and 3.  This is different from that in Figure 3 because a 4-way splitter actually consists of three two-way splitters:  One to split two ways, and this path is then split two more ways with ports 1 and 2 on one splitter and 3 and 4 on another - and cross-coupling to the "other" splitter is apparently not as good at frequencies below the design.

It is worth noting that all of the above measurements are contingent on all ports "seeing" a 50 ohm source and load - either from the instrument itself doing the port-to-port measurements, or by terminating the "unused" ports (e.g. those not involved in the measurements) with known-good 50 ohm loads.  It is likely that real-world devices (antennas, receivers, amplifiers, filters) connected to any splitter will not have as good a return loss (effectively, VSWR) as a load and this will affect the isolation and apparent insertion loss.

Despite what the "official" ratings say, this device would be suitable down to at least 160 meters (1.8-2.0) MHz and likely usable through the entire AM broadcast band and, possibly, the 630 meter band.
Figure 5:

Insertion loss of the ZFSC-4-1 splitter between 10 kHz and 60 MHz.
Click on the image for a larger version.

ZFSC-4-1-BNC+ four-way splitter:

This device, also equipped with BNC connectors, has an "official" frequency range of 1-1000 MHz, splits the signal 4 ways with a theoretical insertion loss of 6 dB, but practically speaking, the actual loss is rated as being closer to 6.4 dB over the lower end of the range.

Figure 6:
 Typical port-to-port isolation of the ZFSC-4-3 from 10 kHz to 60 MHz.
Click on the image for a larger version.
As Figure 5 shows, this device does a much better job at the low end of things than the ZFSC-4-3:  At 100 kHz, the insertion loss is only slightly (0.2dB) higher than at 1.8 MHz and Marker #3 at this frequency on the Smith chart shows a reasonable (approx. 1.5:1) VSWR.  By the time one gets to 20 and 10 kHz, the VSWR and insertion loss have risen - but not as bad as that of the ZFSC-4-3.

Figure 6 shows the typical port-to-port isolation (ports 1 and 2 in this case) showing that down around 100 kHz, the isolation has dropped to about 20dB - still reasonable, and comparable to the isolation to be expected at the high end (published specs, near 1 GHz) of the design frequency range.

Clearly, if one has the amateur 2200 and 630 meter bands in mind - or one is splitting signals above about 100 kHz to feed several receivers - this is a much better choice than the ZFSC-4-3.
Figure 7:
 Insertion loss of the ZFSC-2-1+ two-way splitter between 10 kHz
and 60 MHz.
Click on the image for a larger version.

ZFSC-2-1+ two-way splitter:

This device, equipped with BNC connectors, has an "official" frequency range of 5-500 MHz (the "plus" version has the same ratings), splits the signal 2 ways with a theoretical insertion loss of 3 dB, but practically speaking, the actual loss is rated as being closer to 3.3 dB over the lower end of the range.

Figure 8:
Port-to-port isolation of the ZFSC-2-1+ two-way splitter between 10 kHz
and 60 MHz.
Click on the image for a larger version.
Figure 7 shows the measured insertion loss and surprisingly, it looks quite good down to 100 kHz - probably due, in part, to the fact that unlike the four-way splitters, there is likely only a single ferrite device contained within to incur losses at the low end where it "runs out" of inductance on the transformer.  Down at 20 kHz the loss has gone up by about 2dB and the impedance is in the area of 20 ohms, but this device may still be fairly usable in some applications.

Figure 8 shows the port-to-port isolation and this remains above 20dB down to about 250 kHz, quickly dropping to about 14dB at 100 kHz.

What this tells us is that this device is still likely to be usable down to 100 kHz if one is able to tolerate a couple of extra dB of loss and only mediocre isolation.

Figure 9:
Insertion loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.  Because the "Couple" port is pulling a slight amount of
energy from the through line, a small amount of insertion loss
is to be expected.
Click on the image for a larger version.
ZFDC-20-3 20dB directional coupler:

This device is not a splitter, but rather a device designed to directionally "siphon" a small amount of signal from the "through" line - but do this only do this in one direction.  This device is typically used to sample (with 20dB of attenuation) a signal on a given line, or if turned around to couple in the opposite direction it can insert a signal on this same line.  A common application of this device is to measure return loss (or VSWRm using a pair of these devices), allow non-intrusive monitoring of signals on a cable and it can be used to insert a signal on that same line - say for receiver sensitivity testing - on a cable that cannot be interrupted.  Unlike a splitter, connecting/disconnecting a device on the "Couple" port will have a very small effect on the through-signal.

Figure 10:
Forward coupling loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.
Click on the image for a larger version.

The "official" specs of the ZFDC-20-3 indicate a frequency range of 200 kHz to 250 MHz, but one can see in Figure 9 that the insertion loss is well below 1 dB down to around 20 kHz - although the VSWR at this frequency climbs to nearly 3:1:  At 50 kHz, the insertion loss is still only about 0.25dB and the VSWR is about 1.5:1 - still within the usable range for applications that can tolerate a small amount of degradation.

On the sample port we can see on Figure 10 that the coupling is ruler-flat down to at least 100 kHz and still staying within 1dB of the nominal value down to 10 kHz - but one should keep in mind the fact that the insertion loss and the varying impedance will likely affect the through-line's signals below around 50 kHz.
Figure 11:
Reverse coupling loss of the ZFDC-20-3 20 dB coupler from 10 kHz
to 60 MHz.  Because the coupling is 20dB in the forward direction,
the attenuation values depicted in the above graph should be reduced
by that amount.  The "bump" at the extreme low end is an artifact
of the configuration of the test instrument.
Click on the image for a larger version.

Figure 11 shows the "reverse" coupling loss (e.g. "directionality").  Ideally, no signal should be detectable when the "load" is a perfect, non-reflective 50 ohms but due to imperfections in the load, device, cabling and measurement will reduce this.

This shows that the absolute directionality+coupling exceeds about 60dB (about 40dB of directivity compared to the "forward" coupling) at all frequencies below 60 MHz down to about 20 kHz:  Values below about 70dB (the "floor" between 20 kHz and 10 MHz) are representative of the limits of the test instrument and its configuration so they may actually be greater than this.  Below about 15 kHz, the "bump" is mostly due to measurement artifacts - but this still indicates that the relative directionality is at least 30dB.

These measurements indicate that this device is usable down to 50 kHz with only minor degradation, and would probably work down to 25 kHz in applications where one can tolerate a bit of extra insertion and return loss.

Final comment about the Mini-Circuits devices:

In reviewing the above tests, it would appear that these Mini-Circuits four-way splitters and the directional coupler are generally useful down to about 1/10th of their "official" low frequency rating and that down to 1/5th of their low-frequency rating, they more or less meet their "official" specs.

Follow-up article to come:

I have built several homebrew versions of 2 and 4 way splitters to see if I could, for little cost, come up with suitable versions of my own - and these will be described in a follow-up article.

* * *

Stolen from


Sunday, October 27, 2019

Shunt regulation of series-connected lead-acid batteries to equalize the voltage

There are many instances where series lead-acid batteries are connected - including:
  • 12 volt systems using two 6 volt "golf cart" batteries in series
  • 24 or 48 volts in a UPS (Uninterruptible Power Supply) consisting of several 6 or 12 volt batteries
  • A UPS system that requires much more than 48 volts - more on this, later.
One issue that arises with any system in which cells/batteries are series-connected is that the voltages across them will ultimately be unequal.  For example, in a hypothetical 24 volt system where the nominal float voltage would be about 27.1 volts it is likely that the two "12 volt" batteries will be slightly different.  If the voltages are "pretty close" (within 0.1 volt or so) there is probably on reason to be worried - but as batteries age, they will inevitably drift apart:  Before you know it you might end up with one battery at, say, 14.2 volts where it will be evolving gas (or if it is a "sealed" battery like an AGM, will start to lose electrolyte) and the other battery will then be around 12.9 volts where it will be chronically under-charged, leading to loss of system capacity.

In either case - if the battery is exposed to a consistently high or low voltage - the life of the battery will be reduced - possibly dramatically!

Figure 1:
A small pile consisting of four shunt regulators.
For convenience, these are outfitted with spring-loaded
alligator clips which allows easy installation and removal -
and eliminates the need to sandwich a ring lug on the
battery terminal and potentially increase the total series
resistance of the bank by doing so.
Click on the image for a larger version.
While there are certainly other considerations, keeping the series-connected batterys' voltages in check will certainly go a ways toward maximizing system longevity.

First, a couple words of warning:
  • Batteries are high-current devices:  Shorting can cause injury/fire, so be very careful!  Remove metal jewelry - particularly any rings, or at least cover them with tape.
  • Some types of batteries - such as "flooded-cell" lead acid - can weep a bit of sulfuric acid which can cause burns on skin and damage to clothing.
  • Some UPS systems are not isolated from the mains power and pose an electrical shock hazard:  Always assume this to be the case and take precautions (e.g. power down/disconnect.)
  • Even moderate/low voltages can cause electrical shock.
  • While this information is presented here in good faith, it is up to you to do research about its validity before implement it, taking responsibility for doing so.
  • It is up to you to research and implement safe procedures when working on this - or any - electrical gear and you are solely responsible for any damage/injury that may result.

"How do you know this?"

A quick check of manufacturers' specifications and recommendations will reveal that exposing batteries to either too-high or too-low voltage will compromise longevity, so it would seem to be "pound foolish" to ignore it happening.

As a "case study" I maintained, for several decades, several multi-battery UPS systems - the largest of which was a 300kVa system considering of two banks of forty 12 volt batteries in series that were wired in parallel (e.g. 542 volts nominal) - and with a total of eighty batteries, things are going to drift around!

Having an available BMS (Battery Monitoring System) I could easily track the voltages on the batteries - and inevitably, they started to drift apart as the batteries aged, were exposed to slightly different ambient temperatures (e.g. the ones higher up will be a degree or so warmer the the lower ones) and manufacturing variances.  Knowing full-well the implications of batteries that were drifting apart, I soon devised a simple shunt regulator, described on this page:
These "shunt" regulators are very simple:  Just several diodes (including a Zener) plus an LED that allows a selectable amount of "leakage" current so that "better" batteries (e.g. those with lower self-discharge/leakage current) aren't exposed to elevated voltages by the current of the "other" batteries.  By "tweaking" things one can balance the system so that all batteries will obtain very close to equal voltages.

Despite the fact that only the current through a few lowly LEDs (20-30 milliamps, maximum) is microscopic compared to the currents seen when the UPS was "on battery" and fully-loaded (several hundred amps) this system worked well for several years - but it required frequent adjustment as the batteries aged and their own internal leakage currents started to change:  The battery monitoring system's "live" voltage readout was an invaluable aid to allow tweaking - but this task eventually got "old".

A better shunt regulator:

The "LED+Zener+diode" arrangement has a sharper voltage-current "knee" than a simple resistor, but it was not quite "sharp enough" so I decided to upgrade to a circuit that would respond much more forcefully with increased voltage - and I chose to use the venerable TL431 "programmable Zener".  This chip is ubiquitous, appearing in almost every PC power supply ever made:  It has a temperature-stable onboard voltage reference, and it can handle up to 100 milliamps - several times the current of the original circuit.

This circuit is represented in Figure 2, below:
Figure 2:
Schematic and mechanical layout of the shunt regulators depicted in Figures 1 and 3.
Click on the image for a larger version.

This circuit is very simple in its operation:
  • U1, the TL431 will turn "on" if its reference voltage exceeds 2.5 volts, drawing current.
  • R4, the potentiometer, is adjusted so that the "reference" terminal is 2.5 volts at the desired shunt voltage.  A 10 turn potentiometer is strongly recommended as the setting of the precise voltage will be both "fiddly" and easily disturbed if a single-turn pot is used.
  • When the battery voltage is below the adjusted shunt voltage, U1 is "off", the circuit drawing a few hundred microamps.  This is likely to be less than the self-discharge current of battery itself.
  • When the battery voltage is above the adjusted shunt voltage, U1 will turn on:  The LED will illuminate - the brightness roughly proportional to the shunt current - and the bulk of the current will flow through R2.
  • Because the voltage will be high enough to activate the shunt only when the battery bank is being charged, these shunts will have negligible load when the bank is actually being used (e.g. power being drawn due to a power failure.)
  • R2 (and R1/LED1) limit the maximum current that is likely to be drawn by the circuit in the event that voltage cannot be drawn down below the threshold voltage.  This can occur during bulk charging of the battery banks and it can also occur if the sum of the threshold voltages of the individual shunt regulators is lower than the float voltage - something that could happen on a system that adjusts the float voltage with temperature (discussed below).

In practice, if we were to set the shunt voltage is set to 13.55 volts, the lead resistance connecting the circuit to the battery will result in a very sharp "knee", the circuit going from "off", drawing a few hundred microamps, to "on" and drawing nearly 100 milliamps - in just a few millivolts or 10s of millivolts  - depending on the gauge of the wire used to connect to the battery.
Figure 3:
Internal and external views of the shunt regulator.
These were built "dead bug" rather than on a circuit
board, using sleeving to insulate conductors that might
otherwise touch.  Braided silicone-fiberglass tubing
was used to insulate the large resistor as it is designed
to handle a bit of heat with the remainder of the circuit -
and the top of the braided tubing - being secured with
heat-shrinkable tubing:  The end of the LED and
potentiometer protrude from the end of the tubing.
Click on the above for a larger version.

In operation the it is suggested that the voltage threshold be adjusted to just light the LED at the ideal float voltage in order to force enough current through the charging system to assure that a small amount of current is flowing through each battery - or through the shunt regulator.  In systems that keep the batteries maintained at a constant temperature the float voltage will remain constant, but in some cases - where the batteries are in an uncontrolled climate environment - the temperature and the float voltage may vary - typically being slightly increased, by many "smart chargers", at low temperatures and decreased at high temperatures.

The information in Figure 2 suggests a voltage that works out to be about 2.26 volts/cell, which is a reasonable value for a temperature range between 10C and 35C (50F-95F).  If the charge voltage rises above this value, the shunt regulators will start to conduct - but the voltages across the battery will be equalized, provided that they were adjusted to the same voltage:  The heat produced - even though it may be just a few watts - will not be of detriment in extremely cold conditions to the performance of the battery.  At high temperatures the lower voltage being produced by the charger may not "trigger" the shunt regulator at the average voltage, but it will still keep any errant cells from straying too far in voltage from the ideal:  Because self-discharge and leakage currents of batteries increases at higher temperatures, it is arguably more important that measures to be taken to keep everything equal!

The diagram in Figure 2 shows two options -  Values for a circuit to be used with a "12 volt" battery and values for a "6 volt" battery - but the operation and set-up is identical in each case as described on the drawing itself:
  • Set the wiper of R4, the 10 turn potentiometer, the wiper is at the "ground" end.
  • Set an adjustable power supply for precisely the desired shunt voltage.  For normal "room-like" temperatures with Lead Acid batteries, 13.55 and 6.775 volts is recommended for 12 and 6 volt types, respectively.
  • Connect the unit to the power supply and adjust the potentiometer so that the LED just illuminates.   Note that even a few millivolts will make a significant difference in LED brightness which means that one adjusts several in on session and if they are "approximately" the same brightness, they will be really close to each other in threshold voltage.
  • It is strongly recommended that all units in a particular battery bank be adjusted to the same voltage.

It was convenient, at the time, to construct these circuits in a "dead bug" manner (see Figure 3) with no circuit board:  Once the layout was determined - and thoroughly documented - it took only a few minutes to assemble each unit, trimming/bending/insulating/soldering the leads in assembly-line fashion.

The entire circuit was covered with insulating tubing - but to cover the main heat-generating component, R2, I obtained some high-temperature, silicone-fiberglass tubing.  This tubing extends beyond R2 and a small piece of "normal" heat-shrink tubing is used to cover these components and hold everything in place with the tip of the LED and the adjustment screw of R4 protruding.

Because of the heat being produced - which could be well over 1 watt - R2 should be placed as far away from other components - particularly U1, which also generates heat.  If you choose to replicate this circuit on a small board it is strongly suggested that R2 and U1 be separated - and that R4, the potentiometer, not be placed too-near R2, either.

The leads connecting the unit are color-coded for polarity and in this case, they were fitted with alligator clips which provide a convenient connection to the battery terminals.

Long-term observation:

During the time that the UPS was active, batteries were replaced only when they degraded as indicated by the resistance measurements of the battery monitoring system:  By the time the UPS was finally shut down after 22 years of operation some of the batteries were "new" and some were as old as 13 years and still within their specifications of internal resistance - and much of this is attributable to the fact that these shunt devices did a very good job of confining the "float" voltages of all 80 batteries to within +/- 50 millivolts - most of that variation being due to not all shunt regulator units being more-precisely adjusted than that.

To be sure, the battery monitoring system did do in-situ impedance testing and a battery was pulled and replaced when its resistance exceeded a threshold determined by observation and correlation of the "failed" battery with its actual amp-hour rating measured after the fact:  By the time the internal resistance of the battery exceeded its mark (0.005 ohms for the particular 100 amp-hour, high-current UPS batteries that were used) it had dropped below about 80 amp-hours as measured at the 20 amp rate.

This long-term observation also showed that the LEDs were a useful visual indicator:  If an LED wasn't illuminated at least dimly it meant that the particular battery's leakage current had exceeded the average of idle current (shunt regulator current plus the battery leakage current) and that its terminal voltage was dropping - something that was usually a sign that that particular battery should be watched very closely.

What about an "Equalization charge"?

It is recommended by many battery manufacturers that an "equalization charge" be applied to the batteries periodically  to raise their terminal voltage, presumably "stirring up" the internal electrolyte of flooded-cell batteries.  In such situations, the shunt regulator will try to clamp the voltage, but since it is current-limited, the batteries will still see elevated voltage:  The shunt regulator should help divide the voltage of series-connected batteries to assure that this purposely-high potential will be the same across all batteries.  (Note:  One should not equalize AGM batteries as this can lead to internal gas pressure that can be vented and cause loss of electrolyte.)


You might ask:  "Don't I need to equalize the voltage of the individual 2 volt cells within a 6 or 12 volt battery for this to work?"  The answer is:  It would be nice if this was possible, but connections to individual cells is usually no possible.  Fortunately, in a single battery, individual cells are usually pretty-well matched as they were made at the same time and typically experience (pretty much) the same temperature throughout their lives.

If you have a system that uses individual 2-volt cells, the above circuit will not work at that low voltage - but there is a version of the TL431 (e.g. the TLVH431) that will work at the "2 volts" of individual cells.  Its maximum current rating is lower than the TL431, but it should be adequate for many applications.  The circuit in Figure 2 would have to be modified slightly to accommodate these changes (e.g. adjust values of R2 and R3 - R1 and the LED would be eliminated as the voltage would probably be marginal/too low for it to work) - but that might be the subject of another article.

* * *

Another related page at

This page describes a circuit of similar function, also based on the TL431, that is used for equalizing LiFEPO4 cells.  With component changes to adjust the voltage threshold, it could be modified for 6 or 12 volt batteries.  Because it uses power transistors, it can handle much more shunt current - but note that even with the large 300 kVA UPS, 50-80 milliamps, the current capability of the TL431 itself, was sufficient to keep relatively healthy batteries equalized.

One saving grace is that like most power systems, the UPS was rarely "on battery" which meant that even at just 50 milliamps or so, even an imbalance of a few amp-hours would eventually be equalized - assuming that the batteries themselves were run "completely down".  With a low-voltage disconnect built into a system that cut off the load at the 25%-30% level, this should never happen, so a bank that consists of multiple batteries - even those with slightly different amp/hour capacities (due to manufacturing differences, age) should "track" reasonably well upon recharge.


This page stolen from

Monday, September 30, 2019

A LiIon Pack for the Yaesu FT-530 - and (possibly) other older radios.

Since the mid 1990s my "go to" handie-talking has been the venerable Yaesu FT-530.  Fairly popular at the time, this radio has endured, possessing pretty much all of the features that I would want in a handie-talkie such as dual-band operation, reasonably high power (2-5 watts), the ability to receive out-of-band (NOAA weather, for example), and decent battery life.

When I bought this radio I immediately got higher-capacity battery packs - first NiCd, then NiMH - to allow longer-duration operation over the original 600 mAH battery pack.  I also have a "shell" to allow operation from alkaline cells - the 10 cell version allowing the radio to operate in spite of the significant voltage drop when transmitting due to their internal resistance.

Switching from NiCd or NiMH to LiIon cells:

Over time, the original NiCd and NiMH cells faded and in the early 2000s, I updated the old "high capacity" NiMH battery back to LiIon cells, using four cylindrical "18650" cells, providing 2.0-2.5 rated amp-hours with 7.2 volts (assuming a 3.6 volt/cell nominal voltage) using the cell technology then available.  In 2009, these cells had also faded in their usable capacity so I "refreshed" the pack once again with "prismatic" LiIon cells - this time using foil-wrapped LiPO cells to better-utilize the volumetric capacity within the battery cases and to (slightly) reduce weight.

Flashing forward to 2019, these decade-old cells had begun to show their age, so it was time to rebuild the pack yet again - this time, documenting how the pack was put together.

Note:  That written below would apply if I'd chosen to use 18650 or similar cylindrical cells.

"Re-celling" with prismatics:

Figure 1:
The 10 year old pack.  The old cells have started to "puff up" -
a sign that they need to be replaced.  The separators between
cells for this pack were just thin cardboard.
Click on the image for a larger version
In 2009, I looked around for cells that would better-fit within the original "high capacity" battery case better than the 18650 cylindrical cells and after a bit of searching, I found the model number 703562-2C sold by (information on that cell may be found here) - a 1.5 amp-hour foil-pouch cell that is 7mm thick, 35mm wide and 65mm long and weighing only about 25 grams each (about 0.9 oz).

Taking careful measurements I determined that four of these cells - and the protection circuit - would fit in the case and wired in series-parallel, a 7.2 volt, 3 amp-hour pack could be assembled.  Amazingly enough, these cells are still being sold, available for less than $7.00 each plus shipping.

When using these cells - which are essentially foil pouches with explosive lithium compounds inside - it is imperative that one must take several precautions, including:
  • The use of a "protection" circuit.  Especially when using cells in series, you MUST include a circuit that prevents either cell from being overcharged or over=discharged:  Either state can (and will!) damage the cell, making it more prone to "rapid, explosive self-disassembly".
  • Include current limiting.  A typical "protection circuit" will usually offer overcurrent protection in the event that the terminals accidentally get shorted - and this is a good thing as these cells can produce tens of amps of short-circuit current.   The circuit I used does do this, but there is also a "thermal fuse" included as well for redundancy.  For this radio, 3-5 amps of protection is adequate and will prevent catastrophic results.
  • Allow room for expansion.  These foil-pouch cells will expand slightly with normal use and over time and they should not be packed tightly into the available space - this, to prevent the battery case from being forced apart when this happens, but also to prevent the cells from being crushed/damaged by their own expansion and posing a hazard.
In late 2018, I noticed that one of the two battery packs that I'd put together 9 years earlier had started to swell a bit indicating that they were now at the end of their useful life, so I ordered more cells - and this time, I documented the "rebuild" process.

Warnings and weasel words:

Because Lithium-Ion cells - and other cells - can be dangerous, and there is the possibility of damage or injury, I must insert a few warnings at this point:
  • If you wish to rebuild/build your own battery pack using LiIon - or other types of cells - it is up to you to take the required safety precautions when doing so, and to accept the risk should fire, explosion, damage, injury or even death result.
  • While I can offer advice on how to rebuild battery packs using these cells, I cannot control the quality of the cells, the safety and usability of the build or the way that it is used and implemented.  It is up to you to do due diligence when it comes to safely constructing/using cells and educating yourself on the best way to do - or not do it!
  • Appropriate care must be taken in the use and maintenance of this and other battery/cell types.  It is up to you to determine the most appropriate and safest way to do this. 
    Figure 2:
    The four LiIon cells to be used for the rebuild.
    Click on the image for a larger version.
  • I cannot be responsible for your actions or the results of those actions or any damage/injury that might result.  As mentioned above, make sure that you do your own research, and take due care to ensure the safety in the construction and use of a battery pack.

The rebuild:

Starting out with some "large" battery packs that I'd been using with my FT-530 (e.g. a "high capacity" pack, about twice the length of the original packs) I carefully removed the original contents - including the 2.5mm charging connector, and disconnecting/removing the metal connections for the "drop-in" charger:  Because we will NOT be able to use the original NiCd/NiMH charger, we will not be needing these.

Figure 3:
The four cells installed within the case.  2 mil Nomex sheets have been
placed between the two parallel cells with a 20 mil Nomex
sheet between the two series sets of cells.  The pairs of cells are "staggered"
to better-allow for the expected expansion of foil-pouch cells such as these.
The protection circuit may be seen in the upper-right corner of the pack.
Click on the image for a larger version.
Having on-hand the four LiIon cells required (e.g. 703562-2C sold by  - information on that cell may be found here) and a suitable 2 cell "protection circuit" capable of handling 3 amps or so (also available from I set to work.

To prevent abrasion and possible shorting, each cell must be physically insulated from each other:  We do not want to have the possibility of the cells rubbing against each other - or the possibility of unwanted electrical contact.  For this, I'd used the thin cardboard such as that from a cereal box or a box of crackers when I first constructed the battery in 2009, but this time I used various thicknesses of Nomex (tm) sheet - a nearly-indestructible paper-like insulation material:  This is probably overkill as the thin cardboard had worked fine, but I decided to use it because I'd had it on-hand from another project, having obtained it via EvilBay.  The advantage of the nomex over the cardboard is that it is a bit thinner, allowing a bit more room in the pack in the "thickness" dimension.

Figure 4:
 The completed, rebuilt LiIon battery pack, held
together with some polyimide tape - which I
had on-hand.

Click on the image for a larger version.
As it turned out, the width and length of the cells was narrower and quite a bit shorter than the inside of the battery case:  If I could have found cells that were a better "fit" and no thicker I could have managed more than a 3 amp-hour (nominal) capacity for the battery pack.  These cells' thickness by themselves doesn't give much margin for the internal space within the case itself, but by staggering the centers of these cells a bit there is more room for expansion of the cells in their normal change in size as they are cycled and as they age.  The FT-530 case has several millimeters of additional clearance in the dimension of cell thickness to allow for such expansion.

Clear RTV (silicone) sealant/adhesive is used to hold everything together:  A thin layer is used between the two parallel cells to hold the Nomex sheet in place - and to hold the two cells together and more RTV is used to hold the two sets of cells at an offset.  The use of RTV is suggested as it is flexible and has some "give" - something that is absolutely necessary with these types of cells to avoid damage during their normal use and lifetime.


One cannot use the original NiCd or NiMH charger for the "new" battery - instead, a constant-voltage, current-limited supply is used.  While this sounds complicated, I simply used an LM317 adjustable voltage regulator - the circuit being taken from the standard data sheet - depicted below in Figure 6 - with the exception that a 5k, 10-turn potentiometer is used to set the voltage, and there is an LED (with series resistor) placed across the output as an indication of applied voltage.

To charge, the regulated supply is set (nominally) to 8.2 volts and allowed to charge for 6-8 hours when powered from a 1-2 amp power supply, after which time the battery should be removed from the voltage source:  In the interest of longevity one should NOT apply charging voltage continuously as maintaining a battery at "full charge voltage" accelerates chemical degradation of any lithium-based rechargeable cell.  The precise amount of time to leave the battery "on charge" isn't critical:  It should be long enough to adequately charge the battery, but it should not be ignored and simply left for long periods.

It is certainly possible to find a cast-off or surplus "8.4 volt" LiIon charger intended for two cells in series and this device could be adapted.  The caveat to this is that this charger must have been designed for a "protected" battery pack - the clue to this being that the charger will have exactly two connections to the pack to be charged.  Of course, it will be up to you to come up with a way to make a connection to the original battery pack - perhaps in a manner similar to that depicted in Figure 5, below.

Making LiIon cells last longer:|
Figure 5:
Homebrew charger for the LiIon pack.  In the lower left is a connector,
made from circuit board material and hobby brass, that slides over the
top of the battery pack to make contact.  The circuit itself (upper right)
is built onto a heat sink and contains an LM-317 voltage regulator
circuit, an LED, and a 5k, 10 turn potentiometer to set the voltage.
The LED, connected on the output side of the regulator, illuminates
when voltage comes from the battery (e.g. to make sure that a connection
is made to a charger) or the 12 volt (nominal) power source for charging.
Click on the image for a larger version.

To further-promote longevity of the battery it has been suggested by battery manufacturers and other "experts" 1 - footnote that the charge voltage be reduced from the nominal 4.2 volt/cell value - a value of 4.05 volts/cell (or 8.1 volts charging) to double the lifetime (in terms of charge cycles) and slow the inevitable degradation over time:  This is likely one of the reasons why I got about 10 years out of the original set when a typical LiIon cell/battery will last about 5 years.

What about capacity loss from not "fully charging" the battery?  Charging to 4.05 volts/cell will yield about 80-85% of capacity as compared to a full charge of a new cell, but it is not uncommon for a new LiIon cell - charged consistently to the "full" 4.2 volts/cell - to lose 15-20% after the first year due to degradation.  What this means is that in the long run, the net loss is mitigated, anyway, also offset by the slower degradation of the cell over time by this same reduction in charging voltage.

Actual use:
Figure 6:
Circuit diagram of the LiIon charger - just a "datasheet standard" circuit
to regulate to the desired charge voltage.  The LED may be any color and
is used to detect when the battery is properly connected to the charger
and to indicate the presence of charge voltage.  U1, the LM317, should be
mounted to a heat sink.  Potentiometer R1 is adjusted to set the output
to the desired full-charge voltage.
Click on the image for a larger version.

Having used LiIon cells with my FT-530 for about 20 years now I would not go back to the NiCd/NiMH types again - except, perhaps, unless they were put in the 10-cell AA battery pack shell that I also have.  For many years my FT-530 has been programmed to turn itself on (using its built-in clock) in the morning to monitor a local repeater - and it will turn itself off after 30 minutes unless I hit a button or transmit:  If I only listen, the radio will go 2-3 months between charges when used this way.

In "heavy" use - such as a public service event where frequent transmission is required - I can use the radio at least "all day" at full power (2-2.5 watts output) without running it down.  When the cells are depleted, I get reasonable warning - particularly since the FT-530 has an on-screen voltmeter than may be enabled.

If I see the battery voltage drop below 6.9-7.0 volts during receive I know that I should consider finding the spare pack - and the radio will continue to work until the point at which the display starts to flash at about 5.5 volts.  Interestingly, the radio will quit (turn itself off) before the "protection circuit" kicks in due to low cell voltage - but little actual battery capacity is left anyway below 6.0 volts, so it appears that a "2S" (e.g. two series) Lithium-Ion cells are a pretty good match for the FT-530.

Comment:  The FT-530 is capable of operating from up to 15 volts, so a "3S" (3 cell in series) pack is possible - but the limited space in this battery pack case means that, at least with these types of cells, that six of these cells cannot be put in series-parallel:  Three of these cells in series would yield a "10.8 volt" 1500 mAH pack rather than the nominal "7.2 volts" at 3000 mAH of the pack described here.

Use of LiIon Packs with other, older radios.

Until, perhaps, the early 2000s, many amateur handheld transceivers were shipped with NiCd or NiMH battery packs - and this shows that with proper care and attention to detail that it should be possible to retrofit one of these battery packs with modern LiIon cells, giving the original radio - which is still likely to be very usable these days - much more operational capacity out in the field.

Before "converting" such a radio, there are several things to consider:
  • The voltage range of the radio.  Keeping in mind that the voltage of a LiIon cell can vary from 4.2 volts, freshly off the charger, down about 3.0 volts meaning that a hypothetical 2-cell battery such as this can produce between 8.4 and 6.0 volts, spending most of its operating time in the 6.8-7.2 volt range.  If, in this 2-cell example, your radio can happily operate in this voltage range, your radio might be a good candidate.
    • Other examples:
      • 3 cells:  9.0-12.6 volts, spending most of its time in the 10.2-10.8 volt range.  May be usable for radios that are capable of operating directly from a 12 volt supply.
      • 4 cells:  12.0-16.4 volts, spending most of its time in the 13.6-14.4 volt range.  Many "12 volt" capable radios have an upper voltage limit of 15.0 volts, so this may be too high for safe operation with a freshly-charged battery. 
  • A means of charging.  Make sure that you can charge any battery that you might put together.  In my case, I built a very simple charger (see figures 5 and 6) but I could have adapted an old LiIon charger - or even used a bench-top power supply that was properly adjusted for the charge voltage.  If your radio has some means of built-in charging, it probably cannot be safely used.
  •  A usable battery pack case.  In many instances one can re-use the original battery pack's case and stuff into it appropriately-sized cells as was done here.  If one does this it will be necessary to carefully measure the internal dimensions and find the prismatic cells that will fit inside - remembering to allow room for the "protection circuit".
* * * * * *


1 - "How to Prolong Lithium-Based batteries" -

This page stolen from


Wednesday, August 14, 2019

Revisiting the limited attenuation high-pass filter for the KiwiSDR

In the June 18, 2018 entry of this blog (see that page here) I described a device that reduced the lower-frequency (below approximately 10 MHz) by approximately 12dB while leaving higher-frequency signals (pretty much) untouched.

Why the need?

As it turns out, the KiwiSDR is "sort of" deaf.  Using a variety of measurement techniques, the absolute sensitivity of the KiwiSDR sitting on my workbench at 28.25 MHz was determined to be approximately -155dBm for a 0 dB signal-noise ratio in a 1 Hz bandwidth.

While this may sound impressive, it isn't quite enough to allow the receiver to "hear" the theoretical noise floor of -160dBm (1 Hz bandwidth) at 30 MHz according to ITU-R P.372.7 as depicted in the chart below in Figure 1:
Figure 1:  "Typical" noise floor for various radio environments.  Because the above chart is based on a 500 Hz bandwidth, one would subtract 17dB from the power level to scale to a 1 Hz bandwidth.
While it is likely that most RF environments - typically urban environments - are above the "Quiet Rural" line depicted in Figure 1, it does show that if you happen to place the KiwiSDR in a particularly quiet location, it will not "hear" the signals that are right at the predicted noise levels.  If there are other losses in the system - such as those caused by the cabling or splitters (e.g. for multiple receivers) the situation could get even worse.

The obvious answer is to add an amplifier:  Assuming no other losses, about 10dB is more than enough to overcome the KiwiSDR's noise floor - plus "a bit extra" to minimize the dilution by the receiver's noise.

There is a problem with doing this is hinted at the nature of the graph itself.  As one can see, the noise at 5 MHz is nearly 20dB higher than that at 30 MHz.  While this means that the intrinsic sensitivity of the receiver is more than adequate at these (lower) frequencies, there's another problem:  Signals at these lower HF frequencies will also be very much stronger.

It was observed that the KiwiSDR would exhibit an A/D converter overload (at 28.25 MHz) at -15dBm - and while this is a much higher level than the signal levels depicted in the chart above - because Figure 1 just depicts the noise level - the fact that the receiver itself is inherently broadband, much more noise is intercepted.  For example, if we were to re-scale the above power levels for a 5 MHz bandwidth, the noise power alone would be increased by 40-ish dB.

This does not take into account that the frequency range below 10 MHz is replete with strong signals in most parts of the world - particularly at night, some of which have been measured to be stronger than -30dBm - and there are multiple signals of this sort that are present, the total power of which can be cumulative.  What make things worse is that on these frequencies there are very often strong static crashes - particularly in the summer - that may be equal or stronger than the signals present in their "S-meter" reading, but these crashes are inherently broadband, which means that the receiver is intercepting much more signal than the signal meter will indicate.

The "solution" to this is to put the (overall) signal gain where it is needed:  Amplify the high-frequency (e.g. above approximately 10 MHz) signals more than the low-frequency signals - and one way to do this is to construct a filter that leaves attenuates the lower-frequency signals without bothering the higher-frequency signals.

But the previous filter already does this!

The original filter that does this has been in service for months, now - and it has been working very well, but when I installed it, I overlooked something:  The gain of the antenna being used drops off precipitously at MF and LF frequencies.  What this meant was that with the 12 dB or so drop in signal level by the time one gets to 7-8 MHz persisting down to DC, the signals on the 630 meter amateur band (and lower, for that matter) are also attenuated by the same 12dB - but these same signals - from the antenna - are already dropping off, putting these lower-frequency signals (again) below the KiwiSDR's noise floor.

Reworking the filter:

To that end, I re-worked the filter.  Previously, it was simply a 3rd-order high-pass filter with some "bypass" so that a limited amount of the lower-frequency energy would be allowed through and this meant that from the cut-off frequency down to (essentially) DC, there would be 12-ish dB loss.  What I needed, instead, was to affect the lower HF frequencies, but leave the very low frequencies alone.

There was a complication:  The signal path for the KiwiSDR already includes an effective filter for the AM broadcast ("mediumwave") frequencies (described in the 15 February, 2018 blog entry - "Managing HF signal dynamics and preventing overload with the RTL (and KiwiSDR) receivers" - see that page here) and to have both sets of filtering in series - as it is now - would mean that the KiwiSDR would have difficulty hearing weaker signals on the broadcast band - as it does now.  This meant that I needed to reject frequencies between approximately 1.7 MHz to 10 MHz, but leave the signals outside that range alone.

For this, the free "ELSIE" program came to the rescue:  A 3rd-order Butterworth filter, centered on 4.2 MHz with a 10 MHz band-pass was determined to provide the necessary rejection at the boundaries and like the previous filter, it, too, would have a controlled amount of bypass to allow some signal to pass through it as the diagram in Figure 2, below, shows:

Figure 2:
The response plot of the limited-attenuation band-stop filter.
Click on the image for a larger version

The schematic of this device of may be seen here:
Figure 3:
Diagram of the limited-attenuation band-stop filter.
Click on the image for a slightly larger version.

If you have visited the 15 February, 2018 page, you will notice very distinct similarities between its main filter element and this circuit, right down to the application of signal "bypass" to set a maximum amount of attenuation that can occur.

In this filter, L1/C1, L2/C2 and L3/C3 are resonated to 3.7 MHz with the values selected to provide the desired attenuation at the frequencies at which the cut-off is to begin.  In this case, this filter is a slightly-tweaked version of a 3-pole Butterworth filter designed for a 50 ohm termination and has a theoretical 3dB passband of 11 MHz centered at 3.7 MHz.  The theoretical -6dB points of basic filter - ignoring R1/R2/L4 - is approximately 1.6 and 8.5 MHz with the -1dB points occurring at around 1.1 and 11.8 MHz.

Components R1/R2/L4 provide a degree of "bypassing" that leaks a controlled amount of signal around this filter:  Without these components, the attenuation could be in excess of 60dB near 3.7 MHz, but as can be seen, the actual attenuation is around 14dB, +/- 1dB or so.  While a simple resistor could have been used to accomplish this, the L4 slightly reduces the attenuation at the high end of the HF spectrum while R2 suppresses some of the asymmetry seen in the bottom of the attenuation curve that is caused by L4.

Figure 4:
As-built limited-attenuation band-stop filter.  This circuit - later put in an enclosure - is built "Manhattan" style using a combination of molded and toroidal chokes.  The BNC connectors visible were temporary, used only on the workbench for testing and characterization.
Click on the image for a larger version.

  • As can be seen from the Smith chart in Figure 2, this filter provides a 50 ohm match only at frequencies removed from the portion where the attenuation is occurring.  For this reason it is recommended that this filter be placed fairly close to the receiver (or splitter, if several receivers are being used) - this, to prevent impedance transformation on the line.  Similarly, it is recommended that this filter be preceded by some sort of amplification to source the filter with something near-ish 50 ohms.
  • If amplification is used for the receiver, it is suggested that the amplifier be placed immediately after this filter:  The attenuation at the lower frequencies will reduce the probability of amplifier overloaded by the often-strong signals at these frequencies as well as the summer static.  The impact of the filter on the system noise figure at low frequencies is offset by the typically-high noise level while the low loss of the filter at higher frequencies means that there will be little degradation at the high end of the HF spectrum.

This page stolen from


Sunday, June 9, 2019

Using the same feedline for HF/6 meters and 2 meters/70cm (with a diplexer)

Yesterday, a work party went to the remote HF station of the Utah Amateur radio club to install a new antenna for the lower HF bands.  At the site was an existing G5RV-type antenna which provided coverage on 80, 40, 20 and 10 meters (more or less) - but it didn't provide 160 meter coverage (mostly useful during winter months, after sunset) and this G5RV was not taking full advantage of the 65 foot tower on site, being anchored near the top, but sloping down toward ground that was also rising to the west.
Figure 1:
The exterior of the completed diplexer designed to allow
HF+6 meters to co-habitate with 2 meters and 70cm on the same feedline.
Click on the image for a larger version.

Taking this antenna design on as a project, former Utahn/club member Mike, WA7ARK, decided to take advantage of his recent research, simulating, and real-world testing of multi-band end-fed half-wave antennas 1 and suggested a 160 meter end-fed half-wave wire:  If it worked as expected, it would provide useful coverage over the lower half of the 160 meter band, much (if not all) of 80/75 meters, 60 meters and 40 meters:  It may also be useful on 20 and 10 meters as well.

In short, the 1/2 half-wave antenna consists of approximately 260 feet of wire fed at roughly the 0.45 wavelength point by a 1:7 broadband transformer to provide a 49-fold impedance transformation.  When done properly, this combination can provide a reasonably good 50 ohm match on the first 3 or 4 half-wave multiples, (e.g. 160 meters=1/2 wave, 80 meters=2/2 wave, 60 meters=3/2 wave, 40 meters=4/2 wave, 20 meters=8/2 wave, 10 meters=16/2 wave.)  Much above 40 meters, the antenna was expected to lose effectiveness - at least in part due to the higher-order multiples, but also with the transformer "running out of steam" at the higher frequencies.

But I digress...

No antenna is useful without some sort of feedline.  At the top of this tower was already-mounted a VHF/UHF fiberglass vertical fed with 1/2 Heliax (tm) hardline and rather than going through the trouble of running yet another feedline - which would be possible, but probably take more time than we'd have in just one day of our work party - I decided to construct a device that would allow this one feedline - which would be lower-loss than, say, RG-8 style coaxial cable:  Because we'd be "force feeding" the antenna at impedances other than 50 ohms to attain greater operational bandwidth on the covered bands there was incentive to use the lowest-loss transmission line possible.

A "high/low" diplexer:

The solution to this problem is to employ a diplexer - a device that will take the signals from a common port and send high frequencies to the second port and low frequencies to the third port.  In this case, I started out with the basic design goals:
  • Use an inductor-input low-pass filter set comfortably above the 6 meter band - say at 65 MHz.
  • A capacitor-input high-pass filter set comfortably below the 2 meter band - perhaps 130 MHz.
Using the (free!) Elsie filter design program I plugged those values in and did some tweaking, ultimately deciding that an "N=5" Chebychev filter with 0.01dB ripple seemed to be appropriate.  I had to model these filters independently of each other because the free version of the program did not allow them to be bridged together at a common point - but taking the values given by the program and plugging them into "LTSpice" (by Linear Technologies - now Analog Devices) to simulate the combined circuit permitted a more analysis and "virtual" tweaking.

As expected, bridging the two filters at a common point "messes up" the response a bit - but for a circuit as simple as this, some experiment tweaking of values is all that is really needed.  Once I was satisfied with the result, I constructed one of these devices and analyzed it with my (relatively newly-acquired) DG8SAQ VNA to assess both the insertion loss and the matching.

The response for "6 meters and below" is as follows:

Figure 2:
The "low side" response of the diplexer with the VHF+ port terminated.  As can be seen, the insertion loss is below 0.5dB with a "reasonable" match at all frequencies 6 meters and below.  The isolation of this filter at 2 meters and above is well over 45dB.  The spurious response at the top end of this sweep (at around 725 MHz) is likely due to a resonance of the enclosure and has no bearing on its intended use.
Click on the image for a larger version.

At this point I should mention that the need for this filter arose rather suddenly:  About two weeks ago, we had taken inventory of what feedlines were available on the site and knew that we would be "short" a feedline - and in the likely event that we (probably) would not have the time to run a new one, I designed and constructed this diplexer - and a duplicate (one of these is required for each "end" of the cable!) - over the course of two evenings.  Had I more time I'm sure that I could have tweaked values a bit and reduced the insertion loss even more.

Moving the VNA to the VHF/UHF port and putting the load on the HF+6 meter port, I ran another sweep which looks like this:

Figure 3:
The "high side" response of the diplexer with the HF+6 meter port terminated.  The insertion loss here is actually lower than that on the HF port - at least on the 2 and 70cm bands.    The isolation at 6 meters is a bit over 30 dB, increasing to over 60dB at 10 meters - more than adequate for our purposes.
Click on the image for a larger version.
As with the the "low" side, I'm sure that a bit of extra tweaking would have helped things a bit, but for its intended goal - providing an RF path to VHF/UHF vertical - its performance was plenty good - on the order of a commercially-available device.

Figure 4:
Inside the diplexer.  The diplexer was constructed in a box
that had previously been used for some satellite equipment.  The circuit
itself was built "dead bug" style on a piece of glass-epoxy circuit board.
The "common" HF-UHF in/out port is in the upper-left corner, the VHF-
UHF port in the upper right and the HF+6 meter port in the lower-left.
Note that the leads in the VHF/UHF path are kept as short as possible
with the components laying against the ground plane.
Click on the image for a larger version.
Figure 4 shows the interior of the constructed diplexer, built inside a Hammond 1590D die-cast aluminum case from a discarded piece of satellite equipment.  As it happens, the N-type connectors with attached lengths of UT-141 50 ohm hardline had been part of this same equipment and were put to use for the three input/output lines.  The use of the "N" type connectors were ultimately helpful as they are better-suited for outdoor use as they are designed to be weather-resistant on their own:  The connectors on the outdoor box were sealed with tape and wrap, anyway!

The capacitors used are NP0/C0G type ceramic disk, each rated for at least 1kV (and hi-pot tested to 3 kVAC - a bit over 4kV pk) and the inductors themselves are wound using tin-plated 12 AWG copper wire.  It is expected that this device should be able to handle at least several hundred watts on HF and 6 meters over a wide variety of mismatch conditions and 100 watts on 2 meters and 70cm.

Again, had I more time to build this device - and were it absolutely necessary to reduce the insertion loss even more - I would have done more tweaking of the capacitor and inductor values 2.  While the insertion loss on the VHF/UHF port is gratifyingly low, it would have no doubt been even lower if surface-mount capacitors and 50 ohm strip-line had been employed.  Finally, this device could have been constructed in an enclosure of about 1/3rd this size - particularly if a circuit board had been made along with a bit of clever arrangement of the components - but I used what I had, in the time that I had.


The schematic diagram of the as-built filter along with some component information is depicted in Figure 5, below:
Figure 5:
The schematic diagram of the as-built filter along with information about the parts used. The "half turn" specified is simply due to the fact that when you wind a coil so that both wire ends point in the same direction, an extra half-turn naturally exists.  The major modification required when the input of the high-pass section was bridged with the low pass section was to change the value of C3 from 33pF to 18pF.
The "nominal" predicted values of the inductors are:  L1=L3=93nH;  L2=192nH;  L4=L5=47nH. In construction, extraneous factors will require a bit of tweaking of these values to achieve best performance.
Click on the image for a larger version.
All of the inductors are wound using 12 AWG tin-plated solid copper wire on a 3/8" (9.5mm) O.D. drill bit as a former.  12 AWG wire is probably overkill, but tin-plated wire was readily available in this size:  As small as 16 AWG wire could have been used - particularly on the "High" (2M/70cm) side with little impact on insertion loss, but since HF energy actually flows through L1-L3, I would not recommend using wire smaller than that:  The use of smaller wire would somewhat increase inductance for the same number of turns so coils using it would have to be modified accordingly.

All of the capacitors are of the low-loss NP0 (a.k.a. C0G) type and no other type of ceramic capacitor should be used.  If desired, Mica capacitors may also be used:  These may offer slightly lower losses - particularly on VHF and UHF - but they are typically much more expensive.  If only low powers are expected (less than 25 watts) then 50 volt capacitors will suffice, but if a full 100 watts is anticipated on HF - particularly if a radio's built-in antenna tuner is used - that C1, C2 and also C3 (which "sees" the HF energy) be rated for at least 500 volts with 1kV being preferred.  For VHF/UHF power levels up through 50 watts or so, 100 volt capacitors are adequate for C4 and C5.

It should go without saying that the junction between J1, C3 and the end of L1 be kept as short and compact as possible as poor construction practices here - and with the remaining components in the VHF/UHF path - will result in higher losses.  It is recommended that when constructing this circuit on a ground plane such as a piece of un-etched circuit board material (as seen in Figure 4) that all of the VHF/UHF branch components be laid flat, as close to the ground substrate as possible:  If one wishes to etch a circuit board, a 50 ohm strip line  along with surface-mount capacitors of appropriate voltage rating would be preferred.

* * *

In use:

A pair of these devices were installed yesterday at the remote radio site:  One located atop the tower near the feedpoints of the VHF/UHF and the HF antennas and the other indoors, where the feedline enters the building, where the cable from the HF/6M port goes to the HF-6M transceiver and the other goes to the 2 meter/70cm dual band transceiver.

At least with moderate VSWR levels (e.g. up to 4:1 or so) the components within this diplexer should have minimal effect on matching - at least on the lower HF bands.  Because the "new" antenna was designed mainly for operation from 160 through 40 meters, the diplexer should be more or less "invisible" over this VSWR range - which is also about as much range as radios' built-in tuners can typically handle, anyway.  Although no scientific testing was done, there was no discernible effect on the performance of the VHF/UHF radio system.

For the diplexer located atop the tower, three small drain holes were drilled in the side facing downwards and in addition to the connectors (all N-type, which are ostensibly weatherproof) being sealed with appropriate methods, a bead of silicone was run along the seam of the die-cast box that was facing upwards.  These precautions should prevent moisture from accumulating in the box and based on past experience, it should experience a long service life.

* * *


1 - Mike's recent experimentation has been with end-fed half-wave antenna.  While the traditional end-fed antenna (e.g. a "Zepp") has been used for about a century, this differs from the traditional implementation by using a broadband matching transformer to match to the high impedance "near" the end of the 1/2 wave section so that, unlike a traditional Zepp, it will work on multiple bands' half-wave harmonic multiples - and the use of this transformer has the added advantage of providing a DC ground for the wire to eliminate static accumulation.

The strategic placement of this transformer, along with a short "tail" of cable connected to the transformer and appropriate feedline decoupling, multi-band operation without inducing currents along the feedline itself, yielding performance very similar to that of a 1/2 wave dipole at its fundamental frequency while allowing operation on more harmonic frequencies than is typically possible with a standard center-fed dipole.   With antenna lengths appropriate for 160 and 75 meter phone, the 40 meter "resonance" turns out to be just above the 40 meter band, but a small inductor was placed in a 40 meter current node near the far end of the antenna and this moved the resonance near the middle of the band without having an effect 160 or 75 meters.  The transformer that was used was supplied by "" and was designed for the 160-40 meter frequency range.

The use of an end-fed half wave can be advantageous over a dipole since, unlike a dipole, there are only two "ends" that may need to be supported which may be more convenient in instances where there is a single elevated structure for attachment while avoiding the weight and sag of a feedpoint (which may include a balun) that might be located mid-span.  Its relatively high feedpoint impedance can, if the antenna is properly installed, reduce the amount of current flowing in other structures (e.g. the tower itself, the feedline) as compared to an antenna like a "sloper" and both reduce the probability of RFI being caused by transmitted energy as well as interference conducted from the ham shack's electrical system, onto the feedline and antenna - provided that the proper precautions are taken.

2 -  A VNA (Vector Network Analyzer) is pretty much the ideal tool for characterizing a device like this, but satisfactory results could have been obtained using simpler devices.  Insertion loss could be measured using a pair of known-accurate watt/VSWR meters with one on the input and another on the output of the "leg" of the filter, terminating the meter on the output side with a known-good 50 ohm load.  Spot-checks would be done on frequencies - such as on the band nearest the edge (e.g. 6 meters and 2 meters for the low and high sides, respectively) as well as the "next" bands over (10 meters and 70cm.)  Iterative measurements of best VSWR and lowest measured losses should get one "pretty darn close" to optimum.

Similarly, one could use the meter on the input to monitor for changes in the VSWR, iteratively tweaking the inductors as necessary to minimize both VSWR and insertion loss.

* * *

This page stolen from


Friday, May 31, 2019

Revisiting the 20 meter "helical resonator" band-pass/notch filters

In my June 30, 2014 blog entry (linked here) I described simple band-pass and band-stop filters that could offer a degree of isolation between two 20 meter stations.

Why is this important?  Having more than one station on, say, 20 meters is advantageous because (with Field Day, at least) a separate 20 meter SSB and a 20 meter CW (or digital) station will count as two stations on separate "bands".  Having two transmitters on the same amateur band, in close proximity, offers a significant challenge - requiring as much antenna separation as possible and the careful selection radios that will "play nice" in the presence of each other.

These filters would be unique compared other "HF band" filters (e.g. "Dunestar") in that not only would they be offer some rejection from another station on the same band, but you would also be able to transmit through it with acceptable losses.

An added challenge is that some modern software-defined radios use "direct sampling" RF front ends and these are arguably less-able to deal with very strong, nearby transmitters than their analog counterparts - particularly if the other transmitter is operating on a frequency within its band-pass filtering:  In other words, with a radio like an Icom IC-7300, you probably can't co-exist with another station on the same band - Even the "Digi-Sel" in your IC-7610 might not be able to save you!

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As a bit of a challenge to myself, I decided to make these filter elements as cheap as possible, choosing to construct them to fit within a metal 1 gallon paint can.  As noted in the article linked above, the volume of these cans are actually slightly too small for optimal use at 20 meters, but I decided to see what I could do.

Details of the construction of these filters is described in the article linked above.  I will reiterate:  There are certainly better ways to implement many of the aspects of these filters than the way that I did it - but it is a starting point, if nothing else.

How they are used:
Figure 1:
The 20 meter notch (left) and band-pass filters.  These filters, constructed
using gallon paint cans, can be used to provide a degree of isolation between
two stations operating on the 20 meter band - such as an SSB and a
CW/digital station.
Click on the image for a larger version.

The idea is simple - with two separate components:
  • A band-pass filter for use on the CW (or digital) station that would pass the (comparatively) narrow range of frequencies likely to be used for operation in that narrow portion of the 20 meter amateur band.  These would be tuned to pass energy in the general area of 14.02 to 14.08 MHz.
  • A band-stop (a.k.a. "notch") filter that would be used on the SSB station to reduce the amount of impinging energy from the CW (or digital) station, tuned to the same frequency range as being used for transmitting by that station.
The above strategy can work because the CW (or digital) station will operate over a rather narrow frequency range - a few 10s of kHz at most - which means that retuning is not required for the notch filter used on the SSB station and only minor "touch up" tuning might ever be required for the CW/digital station.

When I originally constructed these filters I had available a few pieces of test equipment:
  • Signal generator/transmitter
  • Power meter/watt meter
  • A broadband noise generator
  • A calibrated spectrum analyzer
For measuring insertion loss and checking general matching, one's own HF transciever along with a dummy load and wattmeter is more than adequate:  It will give a real-world indication of the insertion loss of the filter - and a VSWR meter will provide a general indication of the state of the match at the various frequencies.  If the radio being used has a built-in antenna tuner, slight mismatching cause by the filter may be "removed" from the point of view of the radio to assure maximum power transmission through the device.

For determining the rejection of the notch or loss of the band-pass filter the task is a bit trickier, so I used what was available:  A broad-band noise source and a spectrum analyzer - this combination being used in lieu of a spectrum analyzer with a tracking generator.  Practically speaking, the use of a tracking generator versus a noise source amounts to different measurement techniques as they can, in this instance, give equivalent results - but a detailed discussion of this could be the entire topic of a different article!

Re-testing the "paint can" filters with a VNA:

I recently added another piece of equipment to my workbench:  A DG6SAQ "VNWA" (read about that device here -  This is a reasonably-priced (in the $550-$700 range, depending on options, exchange rates, etc.) piece of equipment, usable from a few kHz up to about 1.3 GHz, that can do a "proper" job of analyzing an RF device, being able to measure insertion loss, return loss (which can be used to calculate VSWR) and complex impedance - just to name a few.  For the purposes of building, testing and analyzing RF filter circuits like this - and RF amplifiers - it is a more useful tool than a spectrum analyzer+tracking generator.

While the equipment originally used (transmitter, wattmeter, analyzer) can be used to analyze the critical properties of the filter, the use of a Vector Network Analyzer can simultaneously give several parameters about the nature of the filters' parameters.  It should go without saying that this allows comparatively easy tweaking of the tuning and coupling of the resonators - as well as trying different configurations - and observing the results real-time.

Bandpass filter analysis:

Armed with this gear, I set the bandpass resonator on the workbench and used the VNA to measure its properties.

Figure 2:
VNA plots of the 20 meter helical bandpass filter showing the insertion loss (the blue line near the top) and the impedance (the red circle overlaying the series of green circles) at at various frequencies.
Click on the image for a larger version.

As can be seen, the insertion loss points are as follows:
  • 14.07 MHz:  1.3dB.  At this frequency the match is quite good - well inside the inner-most green circle, which denotes a VSWR of 1.5:1 or better.
  • 14.15 MHz:  3dB
  • 14.25 MHz:  >6dB
  • 14.35 MHz:  9dB
These figures aren't spectacular when it comes to off-frequency rejection and this is pretty much a limit of the loaded "Q" of the filter.  Unfortunately, there isn't too much that can be done about this without dramatically changing the physical design of the bandpass filter - notably decreasing the losses associated with the resonating inductor, but aside from silver-plating it (which would help only "somewhat") it would take some combination of a conductor with much larger surface area and the use of a larger enclosure - or adding a second such filter in series with the first.

(At this time, I do not have a second band-pass filter on-hand or else I would have done an analysis with two connected in series.)

Having said this, reducing the power of an offending signal by just 6 dB (e.g. the attenuation at 14.25 MHz when the peak is set to 14.07 MHz) can have a significant effect on the reduction of the symptoms of front-end overload - particularly if one ascribes the idea that a 6 dB reduction correlates to a 18 dB reduction of intermodulation products according to the "1:3" IMD rule in this situation.

Out-of-band rejection of the 20 meter band-pass filter:

Knowing the efficacy of a single bandpass filter element within the 20 meter band, the question arose:  How well does this filter remove out-of-band frequencies?  The VNA provided an easy answer:

Figure 3:
Plot showing attenuation on the 40, 30, 17, 15, 12 and 10 meter amateur bands through the single 20 meter resonator.  The attenuation figures for various bands can be read in the upper left-hand corner of the plot.
Click on the image for a larger version.
Looking at Figure 3 we note that on the bands below 20 meters, the single band-pass filter has excellent rejection - exceeding 39 dB on 30 meters and over 50 dB on 40 meters and below.  As expected with a capacitively-coupled resonator, the attenuation above the design frequency isn't as good, but it exceeds 25 dB on all HF bands.  By itself, this filter would do an excellent job in reducing overload from adjacent-band signals - particularly on bands lower than 20 meters.

Band-stop (notch) filter analysis:

With a UHF "tee" connector right at the notch resonator (connected as depicted in Figure 5) one gets the results shown in Figure 4, below:

Figure 4:
The "notch" response with no stub between the Tee and the notch resonator.
Click on the image for a larger version.
As seen in Figure 4, the notch has relatively little effect in the upper end of the band (above about 14.2 MHz) - which is as it should be - but the notch depth is only around 6 dB in the CW/digital segment - again, a result of the resonator's limited "Q" and other factors that reduce it from the ideal.
Figure 5:
Notch with no "stub" - the result being shown in the plot
of Figure 4, above.
Click on the image for a larger version.

Fortunately, there's something that we can do about this:  Change the impedance at the resonator in our favor.

If one takes a look at the red circle, we can see that the resistance at marker #1 is in the area of 15 ohms.  If we apply a length of transmission line, we can use it to transform the impedance at the "Tee" connector to do a better job of shunting RF at a particular frequency.

Here's what happens if a 1/4 wavelength stub is connected between the "Tee" and the notch filter, connected in the manner depicted in Figure 7:

Figure 6:
The "notch" with a 1/4 wavelength stub inserted between the Tee and the resonator.  As we can see, the notch depth is greatly increased - around 25dB - but the response isn't particularly useful!
Click on the image for a larger version.
Adding the 1/4 wavelength stub of RG-8 type cable greatly increased the notch depth - around 25dB - and this is evidenced by the fact that the red marker (#4) shows, in the circle, an impedance of just a few ohms.  Unfortunately, this has two undesired effects:
Figure 7:
A stub between the tee and notch element.  As can be seen
in Figures 6 and 7, the length of this stub can change
the depth of the notch and the "shape" of the notch and
the attenuation of the nearby frequencies that are not
to be notched out.
Click on the image for a larger version.
  • The frequency "distance" between the notch and the peak is now too wide - greater than the width of the 20 meter band.  This means that this configuration is not really usable for our intended purpose.
  • There is now an asymmetry in the response:  The "peak" (minimum attenuation) is lower in frequency than the notch.
Also note that an upward frequency shift of the notch occurred due to various factors - mostly the reduced "loading" of the resonator - but because I was interested in the shape of the response rather than the actual frequency,  I did not retune the notch to the target frequency.
What about using something shorter than a 1/4 wave to do a less-dramatic transformation?

Figure 8:
The same arrangement as in Figure 4, but with a shorter cable - one that is approximately 0.15 wavelength.  The VSWR is acceptable with radios with built-in tuners - but careful adjustment of coupling and stub length could bring more of the red circle to the 50 ohm "center" of the green circles of the Smith plot.
Click on the image for a larger version.

In figure 8, a shorter cable was used - one that was about 0.15 wavelength of RG-8 type cable.  As figure 8 shows, the attenuation isn't as great - around 10 dB - but we we see that there is minimal effect on frequencies above 14.2 MHz and we see something else that is interesting:  An asymmetrical response that works in our favor:  There is less attenuation above the notch - where our SSB operation is to occur - than below it.  This "trick" is frequently used in notch filters - such as those used for repeater duplexers - to increase both the notch depth and provide an asymmetry that favors the frequency at which one wishes to have the least amount of attenuation.

Its worth noting that even though there was a frequency shift with this stub, it was much less than that depicted in Figure 6, so I was able to easily re-tune it.

Two notch filters cascaded:

As it happens, I do have two devices that may be used as a notch resonator - the second one being the band-pass resonator - if I use only one of its ports.  The results are as follows:

Figure 9:
The result of cascading two notch filters.  The insertion loss is only slightly higher, but the notch depth is now over 20dB.
In reality, the second "notch" filter was just the band-pass filter with only one of the two ports connected.  A bit more tweaking would likely have reduced the insertion loss and brought the "pass" frequencies' matches closer to 50 ohms.
Click on the image for a larger version.

In this test I placed two notch filters in succession as depicted in Figure 10 - using 0.15 wavelength stubs between the two "tee" connectors (and notch resonators) and between the tees and the resonators themselves:  For some reason, I seem to have a bunch of RG-8 jumpers of about that length - around 0.15 wavelengths at 20 meters, which works out to be in the area of 7-8 feet (a bit over 2 meters) physical length, so that is what I used.

Figure 10:
Two notch elements connected with stubs to improve performance.
With properly selection of stub length, not only can the notch depth be "greater than the sum of the parts", but the actual shape of the response can be adjusted.  For Figure 9, all of the stubs had an electrical length of approximately 0.15 wavelength - but no additional testing was done with other lengths at this time.
Click on the image for a larger version.

As Figure 9 shows, the notch depth is a bit over 20 dB and there is only a slight increase in insertion loss to the 20 meter SSB frequencies:  Clearly, this configuration is definitely having a significant effect on the undesired CW/SSB signal and would probably solve most in-band overload issues!

Final comments:

No doubt different (better!) results could have been obtained with different stub lengths and configurations, but I have only so many random chunks of cable on hand and, more importantly, until(?) I construct another notch and pass resonator, I won't be able to use two cavities in the field.

One thing that is difficult to predict before-hand is the effect of source and load impedances other than ideal 50 ohm resistive that would be present with the output (transmit) and input (receive) impedances of HF transceivers, which are "nominally" 50 ohms:  In reality, these impedances can vary quite a bit (even between receive and transmit on the same radio!) and these differences can have an effect on the precise amount of attenuation that will be seen.  Practically speaking, it will be the transmit loss that will be considered - and this will also be affected by the radio's built-in antenna tuner, if it is inline.  Yet another factor that will affect the performance is the match of the connected antenna - particularly as seen at the far ("radio") end of the feedline - especially if a tuner is used there, too.

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What would I do different if I construct more of these things?  As noted in the original June 30, 2014 blog entry (linked here) a major weak point of these resonators is the physical construction of the capacitive coupling probes:  They are not easy to adjust, and they are quite fragile - a good "jar" of the filter (pun intended!) can knock them out of position and cause detuning.  Having a means of being able to adjust the coupling without disassembly - perhaps the use of plastic screws accessible from outside the filter - would be very useful, allowing both critical coupling and tuning to be better-achieved while greatly improving ruggedness.

Perhaps, that will be a future project.

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