Modifying the MFJ-5008 parabolic ultrasonic receiver for better sensitivity and wider frequency response

Figure 1:
Front view MFJ-5008 parabolic dish with
integrated microphone and receiver
(located on the back side).
Click on the image for a larger version

The MFJ-5008 Parabolic Ultrasonic receiver

Note:

Since the MFJ-5008 is no longer being sold it can be found only on the "used" market.  A future posting in this blog will show how to construct a similar unit using readily-available kits and parts.

The MFJ-5008 was marketed primarily for detecting arcing on failing power line hardware, but there are other reasons why you might use such a device:

• Listen to Bats' echolocation.  The "clicks" emitted by bats are well above human hearing.
• Listen to other animals and insects.  Other animals and insects also emit ultrasonic sounds - both for echolocation and communication.
• Find high pressure leaks.  Leaks in high-pressure systems (water, gas, engines, compressors) often make a lot of noise at these frequencies.
• Locate switching power supplies.  These devices often make noise due to magnetostriction of devices (transformers, coils.)

As I find this topic to be interesting, I've written about the detection of ultrasonic signals on two previous occasions in this blog:

• Improving my ultrasonic sniffer for finding power line arcing by using MEMs microphones - Link
• An ultrasonic superheterodyne receive converter (e.g. "Bat Listener") - Link
  

* * * * *

While there are several devices out there that you can buy to enable listening at these frequencies, the landscape has changed in the past few years when it comes to how one might do this on a budget:

• In years past, the MFJ-5008 was available - its primary purpose being to locate and identify arcing on power lines and related infrastructure.  As MFJ is no longer in business, this devices is available only on the used market.
• Some "bat listeners" have used electret microphones.  These inexpensive capsule microphones - while having good response across the human hearing range - lose sensitivity rapidly above this, limiting their usefulness above 20-30 kHz.  In doing A/B testing with a MEMS and an Electret ("capsule") microphone, the MEMS appears to be superior in every way when it comes to ultrasonic response.
  
• Many "bat listeners" have used ceramic transducers.  Most often found for the 40 kHz range (and some were made at lower frequencies) these can be fairly sensitive.  Their frequency range is quite limited and they are only usable within a few kHz above and below their design frequency at best.  As different types of ultrasonic noise sources tend to occur at various frequencies, being able to detect such energy at various points across the spectrum can improve the usability of the device as MEMS microphones have a wide response.
  
• MEMs-based microphones have become cheap and available.  These devices - based on microscopic elements - can operate over a frequency range from a few 10s of Hz to over 100 kHz making the excellent replacements for the (increasingly hard-to-find) ceramic transducers.  Having a wide frequency range allows the user to tune to the peak frequency of the noise source rather than being limited to the immediate vicinity of 40 kHz.
  

* * * * *

How the MFJ-5008 works

Made by (the now defunct) MFJ Enterprises, this includes a 18" (46cm) diameter vacuum-formed plastic parabolic dish with a 40 kHz ceramic transducer at its focus.  Mounted on the back of the dish is a direct-conversion receiver centered at about 40 kHz that converts energy around this frequency to the audible range.  As can be seen in Figure 1 there is a bar across the front in which the ceramic transducer is mounted - but it also has holes that - along with one located behind it in the plastic dish - form a crude sighting system that works quite well to determine from where detected noises might be emanating.

If one disassembles the electronics of the MFJ-5008 they will discover a small circuit board with rather common components - namely a 555 timer used as the oscillator, an LM386 audio amplifier to drive the headphones and a few common transistors to amplify and convert the ultrasonic signals to audible.  There is a "tuning" control on board consisting of a 10k trimmer potentiometer, but it is not accessible from the outside - and it has a range of about 38-48 kHz:  A slight modification will be necessary to allow us to take advantage of the wider frequency response of the MEMS microphone.

Consider the schematic of the MFJ-5008, below: 

Figure 2:
Schematic of the MFJ-5008 ultrasonic receiver.  The circuitry is straightforward - a simple, run-of-the-mill direct-conversion design that is very similar to the one described in the April, 2006 QST article.  Changes to C2/C8 and the added inductor are noted on the diagram.
Click on the image for a larger version.

 

If we compare the above schematic with that from the April, 2006 QST article, A Home-made Ultrasonic Power Line Arc Detector - link) we see some very striking similarities:  Both use a 555 timer for the local oscillator, both use a series of bipolar transistors for signal amplification, and both use a single JFET for the frequency conversion mixer.  There are some differences, but these are pretty much superficial when you consider that the same goal is accomplished with the same types of components.

A cursory analysis of the above diagram shows that the first two amplifier stages are coupled with 1uF capacitors allowing the full audio frequency range to pass:  This mystified me at first, but in looking at the circuit board and noting some unpopulated parts locations I realized that there may have been plans to allow this circuit to be used at audio frequencies - and, perhaps, have a switch to select audible or ultrasonic ranges as well.

For the original 40 kHz ceramic transducer, this wide frequency range isn't a problem, but for a MEMS microphone - which can hear equally well over a 100Hz through at least 60 kHz, this would be:  As the mixer (Q3) is just single-ended, it will happily amplify the original input as well as do a frequency conversion meaning that you are likely to hear audio-frequency "bleedthrough" on the audio output - and indeed, when I retrofitted it with a MEMS microphone (to be described shortly) I did.

Figure 3:
Picture of the MFJ-5008 with location of the various
various components and board locations involved in the
modifications annotated.
Click on the image for a larger version.

Adding "proper" high-pass filtering to the MFJ-5008

The only sort of "high-pass" filtering present are capacitors C10 and C11 which are conspicuous by their being in series:  Why use two capacitors (1000pF and 220pF) rather than just a single 180pF capacitor?  The answer lies on the circuit board where there are unpopulated locations marked "L1" and "L2" (see Figure 2) which correspond with an (uninstalled) pair of inductors between the junction of C10 and C11 and ground.

To make the unit much less sensitive to audio frequency - and to make it more compatible with a MEMs microphone, several changes should be made:

• Change C2 and C8 to 0.01uF (e.g. 10nF) capacitors.  This will prevent the first two amplifier stages from being overloaded by audio frequencies and go a long ways in prevent "bleedthrough".
• Install inductance at the positions of L1 and L2.  I suspect that two inductors were in mind when they designed the board as high-inductance, surface-mount devices are comparatively rare and expensive, so they could use a pair of lower-value coils to get the desired value.  See the footnotes on the bottom of this blog for suggested inductors.
  

• 

  Figure 4:
  Apparently designed to be used in several ways, the MFJ-
  5008's board has several unused parts locations, including
  positions for an inductor that could be used for improved
  high-pass filtering as shown here.
  Click on the image for a larger version.
  

  Connect a 4.7k resistor between the center pin of the RCA connector (to the microphone) and the "V+" pad near the un-populated switch.  This inserts a current-limited 9 volt supply on the microphone lead.
  

The amount of inductance to install at L1 and L2 isn't too critical, but finding such components may be awkward - but the total amount of inductance to use may be anything between 27mH (that's milliHenries!) and 68mH with 47mH being optimal - a relatively huge amount for an SMD device.  In perusing my collection of inductors, I found a through-hole 27mH inductor that I tacked into place, securing it with glue:  Note that it gets soldered across the two pads of L1 and L2 closest to the socketed 555 IC as Figure 4 depicts.

When modifying the MFJ-5008, the MEMS microphone was fitted first and it became clear that audio-frequency energy sailed right through the system, significantly reducing its efficacy at the detection of ultrasonic energy.  It is my opinion that both the changing of C2 and C8 to 0.01uF capacitors and the addition of the inductor are necessary modifications for good performance. 

Note:

If you don't have a suitable inductor for the above modification, the receiver will still work, but you will hear a bit of audible frequency bleedthrough:  In a location with high ambient noise, this may be a problem, but in an otherwise quiet location, it probably won't be an issue:  Changing C2 and C8 do a reasonable job of reducing audio-frequency response and should be considered to be mandatory if you use a MEMS microphone.

In other words, if you don't install the inductor, don't let that stop you from making the modification to the MFJ-5008 and using it with a MEMS microphone - just be aware of the audio frequency "bleedthrough" issue.

Extending the tuning range of the MFJ-5008

Figure 5:
Potentiometer R10 - originally 10k - was replaced with a
50k miniature potentiometer to allow tuning.  A 4.7k resistor
paralleling R22 can be seen in this fuzzy photo.
Click on the image for a larger version.

As the tuning control isn't readily accessible - unless you drill a hole in the box and use a screwdriver - a modification is required to both make the tuning accessible and increase the range.  To do this, I found a small 50k potentiometer and soldered it into place where the original 10k trimmer (R20) was:  Note that two of the potentiometer's leads are connected together, so the "new" device would go between Pin 7 of the 555 in the schematic and resistor R22.  While doing this, R22 should be changed from its original value of 10k to 3.3k (or you could tack a 4.7 or 5.1k resistor in parallel with it).  Increasing the value of R20 from 10k to 50k allows the frequency to be tuned down to 20-22 kHz while lower the value of R22 allows it to be tuned above 50k, the range where noisy, arcing connections (and bats!) is likely to be found.

Figure 6:
The modified MFJ-5008 with the (barely visible)
tuning knob sticking out on the left.  The blue
label indicates the approximate tuning frequency.
Click on the image for a larger version.

While I was able to cram the (very small) potentiometer onto the board (Figure 4), you may need to be creative - possibly mounting the potentiometer on the cover or side of the box using (very short!) flying leads:  If you use a metal potentiometer, I suggest connecting is body to the "ground" of the circuit (e.g. the outside shell of the microphone's phono plug) to prevent pick-up of nearby electric fields that might affect tuning.

The final result of the modification can be seen in Figure 5:  The cut-off shaft of the potentiometer protrudes slightly out of the left side of the enclosure and there is a label depicting the approximate frequency of the oscillator (and the center of the converted range) with respect to the adjustment of the potentiometer and its white paint mark.

What potentiometer to use?

To fit in the location of the original 10k trimmer, one needs to use a small potentiometer:  A suitably small potentiometer is the Bourns 3310C-001-503L which is available from DigiKey HERE and from Mouser Electronics HERE.  With a bit of care, it can be mounted to the board and the case modified to allow the shaft to protrude out the side - but it would be a good idea to use something (e.g. "hot melt" glue) to make it more rigid and prevent fatiguing/breaking the potentiometer's leads.  If you are creative, a larger potentiometer might be usable, attached with flying leads, but if it's metal, be sure to connect its body to the V- (battery negative - the shell of the phono plug will work) to minimize noise pick-up.

Note:

If you don't make the (highly recommended!) "tuning" modification, the MEMS microphone is still useful in that its sensitivity extends over a wide frequency range:  You may be able to adjust the original potentiometer (which can be adjusted between 35 and 48 kHz) to a frequency that is more suited for the types of noises that you are seeking.

Using a MEMS microphone

Note:  

In this section, I refer to a "homebrew" MEMS microphone carrier board - but there are "breakout" boards available that are already assembled:  This next section describes how either a "breakout" board or a homebrew board like this may be mounted in the focus of the dish.

Figure 7:
The original 40 kHz ceramic transducer and
carrier board (top) and the homebrew version with
the MEMS microphone (bottom) both mounted using
the pairs of screws on stand-offs in the front bracket.
Click on the image for a larger version.

Farther down this page you will find a description of a commercially-available MEMS break-out board (from SparkFun) and how it may be used, should you be unwilling to assemble your own!

                    * * * * *

For the specific MFJ-5008 depicted in this article I used an already-prepared MEMS microphone module:  This was described in a previous article linked HERE.  This circuit was designed to accept a wide range of voltages (3.5-10) to be imposed onto the same conductor as the audio, making it easy to interface on a single cable as we did here.

In the MFJ-5008, there is an aluminum "U" channel across the front in which the ceramic transducer is mounted and its location places it at the focus of the parabolic dish.  What this means is that when we replace this device with something else - a MEMS microphone in this case - it must not only be located at the same axial position (left, right, up, down) as the original, but the sensing element must also be at the same distance from the surface of the dish.

Behind the nesting cover (accessible via the removal of four screws - two at each end) there is a circuit board mounted on two stand-offs and the focus of this dish is precisely midway between the two.  Removing this and peering inside the original ceramic transducer, you can see the element located inside, recessed slightly from the front grille:  The distance of that element from the circuit board is that which should be replicated with the replacement microphone.

Figure 8:
Homebrew carrier board with MEMS
microphone installed, facing the surface of the
dish.  The microphone's "sound hole" - facing
from the camera in this photo - is located
precisely between the two mounting screws.
Click on the image for a larger version.

As can be seen in Figures 7 and 8, I mounted the homebrew MEMS modules on the "front" side of a piece of PCB prototype board, taking care of placing the center of the microphone (not visible in the photo) on the center line between the two screws and equidistant between them.  Once this was done, the "new" microphone was mounted back in the "U" channel and the wires soldered as seen in Figure 8.  As it turned out, the thickness of the homebrew board placed the MEMS element at the same distance from the dish as the original element - a fact later verified by noting that the "sharpness" and accuracy of the pointing with the new element seemed to be the same as before.

Using a Sparkfun MEMS microphone "breakout" board

Soldering a tiny microphone module successfully to a circuit board requires a bit of skill - but there are "breakout" boards that already have the microphone and some of the needed components already on them - and one of these is available from SparkFun (the "BOB-19389") for about US$9.00 at the time of writing. While it is possible to order from SparkFun directly, I ordered it via Amazon for the same price - plus shipping was "free".  Detailed information on this board may be found here:

https://www.sparkfun.com/sparkfun-analog-mems-microphone-breakout-sph8878lr5h-1.html

This breakout board contains both a microphone and an operational amplifier and here are their respective data sheets:

• Microphone element data sheet - LINK
• Op Amp data sheet - LINK
  

As originally designed the SparkFun board "sort of" works for ultrasonic detection, but there are a few circuit elements that require attention before we use it.  Consider the schematic, below:

Figure 9:
Diagram of the SparkFun BOB-19389 MEMS breakout board.  As can be seen,
there's nothing special about this design:  A microphone coupled to a single op-amp section - but
but there's a problem with this circuit in our application:  The gain set by R4 is unnecessarily high
for our needs and this - along with C3 - reduce the useful frequency response to less than 30-sh kHz.
Click on the image for a larger version.

The implementation of this breakout board is nothing special - and it's worth noting that even without the gain of the op-amp, the MEMS microphone itself would have a suitable amount of drive for the MFJ-5008.

As part of our circuit analysis, I will call the reader's attention to R4 and C3 (300k and 27pF, respectively) which form a simple low-pass filter - but these components, along with the unity-gain bandwidth product of this op amp being 1 MHz - conspire to cause the frequency response to roll off rather dramatically above 15-20 kHz or so:  It will still detect lower-frequency ultrasonic signals, but sensitivity is reduced while the signals that we don't want (e.g. audio-range frequencies) are not attenuated - and even if the frequency response was flat into the ultrasonic range, it would have way too much gain for our application, anyway!

Figure 10:
A close-up of the SparkFun BOB-19389 MEMS microphone
break-out board.  The location of C3 - now replaced by a
resistor.  Not also that the "sound hole" of the microphone
is on the bottom of the board, facing down in this photo.
Click on the image for a larger version.

The "fix" is to replace C3 with a resistor.  For the MFJ-5008 I would suggest using a 10k resistor in this location and by lowering the gain, the op amp's bandwidth product isn't going to get in the way of the needed frequency response.  While it doesn't really matter if one removes the capacitor or not when using a 10k resistor (the -3dB point for a 10k resistor and 27pF capacitor is somewhere north of 500 kHz) it's pretty easy to remove just the capacitor and replace it with the resistor if you have SMD parts on hand.  If you have only through-hole parts, it should be possible to tack a 1/4 or 1/8 watt 10k resistor across them.  (Note:  I used the MEMS board in Figure 10 for a different project which is why there's a 47k resistor at the position of C3:  A 10k is appropriate for the MFJ-5008.)

The other issue is that of the voltage range of the breakout board's components.  In testing, the board worked "OK" at just 1.8 volts - below the "official" specifications of the the Op Amp - but it worked "better" in the specified 2.3-3.6 volt range.  In the modification for the MFJ-5008 described above, the addition of the 4.7k resistor across the "audio in" phono plug put the full 9 volts battery voltage (minus resistive drop) on this line so we need to do two things to make this work:

• Limit the voltage to the 2.3-3.6 volt range.
• Combine split the audio signal from the voltage at the microphone breakout board.

Fortunately, this is quite easy, requiring just a small number of components and the following diagram shows:

Figure 11:
Powering the SparkFun MEMS break-out board from the audio cable with DC bias on it as depicted in the MFJ-5008 modifications, above.  Capacitor C1 blocks the DC from the Op Amp,
resistor R1 isolates the audio and DC lines while LED1 is used as a voltage shunt to limit the
voltage to somewhere between 2.3 and 3.6 volts:  An ordinary white or blue LED is perfect for this
as they are readily available and provide a voltage in the middle of this range.
Click on the image for a larger version.

Note:  I could have simply run a separate DC line from the circuit board to the detector, but this would have still required regulating the voltage down to the voltage needed for the MEMS device:  Putting DC on the signal line is easy to do and it requires only a few, inexpensive components.

Capacitor C1 has two functions:  Block the DC from the "Audio Out" terminal and to offer a bit of a high-pass frequency response to filter audio-range energy.  Resistor R1 extracts the voltage from the "DC + Audio" line and sends it to the "VCC" terminal on the breakout board and across this, the LED acts as a voltage limiter.  As noted in the diagram above, one can use a blue or white LED as the voltage limiter:  These will "turn on" at between 2.8 and 3.2 volts which is right in the range that we need.  Alternatively, if you have some "old fashioned" red LEDs that operate from about 1.7-1.8 volts, two of these in series will do the job.

Figure 12:
The SparkFun MEMS microphone break-out board with
the circuitry in Figure 11.  These components could be
"dead bug" mounted like shown in the photo or they
could be incorporated on the "carrier" board used to hold
it at the focus of the dish - either method works!  The
"sound hole" can be seen in the lower-right portion of the
board, just above the letter "H".  Note that it is not
centered on the board - something to note when mounting.
Click on the image for a larger version.

It is recommended that you use the "diode test" function of an volt-ohm meter to verify the turn-on voltage of your LEDs and to make sure that they are connected correctly.  If you have a variable-voltage bench power supply, connect it across the two leads and, starting out at less than 2 volts, slowly increase it while measuring the voltage across the "GND" and "VCC" connections:  The voltage should limit in the 2.3-3.6 volt range and you should see the LED(s) dimly illuminate.  In testing I haven't found that light falling on the LED causes any effects in the audio, but if you are, for some reason, worried about that, feel free to cover the LED with black paint, put it in some black heat-shrink tube or shield it from light in some other way.  (Note that in the MFJ-5008, the carrier board is contained within the "C" channel aluminum pieces and mostly shielded from light, anyway.)

These three components may be mounted either as shown in Figure 12 with the components' "flying leads" holding things together, or on a piece of prototype board to function as the "carrier" board of the same type shown in Figures 7 and 8.  Note that the "sound hole" on the breakout board is on the "back" (non-component) side of the circuit board (visible in Figure 12) and that it is NOT in the center of the board and take this into account when you are mounting it to the "carrier" board.

Final words on the MFJ-5008 modifications

The above modifications should allow the MFJ-5008 to work over a wider variety of frequencies to allow optimum detection of energy from electric arcs, high-pressure gas leaks, bats, insects, switch-mode power supplies and many other things.

Prior to modification, a "test range" was set up in my back yard:  A 40 kHz transducer was driven with a sweep/function generator (an old Wavetek Model 180) and the output set at its lowest-possible setting.  From about 33 feet (10 meters) away the "warble" from the swept output was easily audible - but not particularly strong.

After the modification, the subjective impression was that the sensitivity was equal or better than the original 40 kHz ceramic transducer - but a quick walk around the house revealed the ringing presence of several switch-mode power supplies, each producing low-level noises of their own due to magnetostriction of components within - something that was totally inaudible prior to the modification, made possible only by the broad-range response of the MEMS microphone and the added ability to tune the center frequency.

* * * * *

Footnote:

• Here are a few suggested parts for the inductor in the modification of the MFJ-5008 - all 47mH:
• https://www.mouser.com/ProductDetail/Murata-Power-Solutions/17476C?qs=5CKLVr1iF0nvNdEM16T%2F2A%3D%3D
• https://www.mouser.com/ProductDetail/EPCOS-TDK/B82144A2476J?qs=v4Mlc8l4PHmthTExsnwGmg%3D%3D
• https://www.digikey.com/en/products/detail/bourns-inc/RLB1014-473KL/2561378
• https://www.digikey.com/en/products/detail/murata-power-solutions-inc/22R476C/1924732
• https://www.digikey.com/en/products/detail/central-technologies/CTS4HTF-473J/16048522

 * * * * *

This page stolen from ka7oei.blogspot.com

[END]

Impedance matching (auto) transformer and common-mode choke for the JPC-7 dipole and other electrically-short (loaded) dipoles and verticals

Figure 1:
The JPC-7 loaded dipole out in the wild!
Click on the image for a larger version.

Loading coils and "electrically-short" antennas

It is well-known that you can make a "short" wire (e.g. one that is significantly shorter than 1/4 wavelength at the operating frequency) resonant by putting in series with it a coil.  There is no "magic" in this as the inductance of the coil, appropriately chosen, can completely cancel out the capacitance of the electrically-short wire, result being that at "resonance" we are left only with a pure resistance.

In an ideal situation, what we would be left with would be just the radiation resistance of this antenna and for such an antenna, this would mean that the feedpoint resistance would be less than 50 Ohms - probably much less!  In reality, the feedpoint resistance would really a combination of "ground" (counterpoise) losses, conductor losses of the antenna, and losses of the coil itself.

What this means is that if you have an electrically short antenna such as a loaded dipole or vertical with only a series loading coil tuned to resonance at the frequency of interest and no other matching scheme, its feedpoint impedance should be well under 50 Ohms on some bands if it is operating efficiently.

This is often not the case with portable antennas!

Figure 2:
The original stainless steel coil (rear) for the
JPC-7 (and JPC-12) with the coil rewound with
silver-plated "jewelry" wire in the front.
Click on the image for a larger version.

The JPC-7

Some time ago I wrote extensively about the JPC-7 (See the article, "Observations, analysis and field use of the JPC-7 portable "dipole" antenna" - LINK) where I discussed the bits and pieces comprising it:  I have used it in the field a number of times, finding it to work as advertised.

In short, this is a loaded dipole - at least on the lower amateur bands (especially 40 and 30 meters) that is intended for portable use:  On these bands (including 20 and 17 meters) it is physically shorter than 1/2 wavelength and it requires the adjustment of its series inductors to resonate.  On the higher bands (15 and above) its overall length approaches and exceeds a half wavelength meaning that it's a full-sized dipole and is (generally) tuned by adjusting the length of the telescoping sections.

Lossy coils!

There is a down-side:  As sold, it has loading coils that are wound with stainless steel:  As noted in the original article, these coils are very lossy, with MOST of the RF power being dissipated as heat on the lower bands (40 and 30 meters in particular - roughly an "S" unit of signal loss) where a fair bit of inductance is required.

Figure 3:
An example of heating of a stainless steel
loading coil on a short vertical - here, made by
Wolf River.  On 40 meters the temperature of
the coil rose by more than 30F (17C) with
60 watts of RF applied for 60 seconds.
Click on the image for a larger version.

The reason for this is that an electrically-short antenna (one that is physically short compared to the wavelength.)  The total length of the telescoping sections alone put together is about 198" (5 meters) - which is about 12.5% of a wavelength at 40 meters implies that the feedpoint resistance would, were there no loss at all, be around 8-10 Ohms, resulting in a VSWR of more than 4:1.

Calculations and measurements indicate that the approximate Ohmic loss of the original stainless-steel loading coil - if we optimistically presume it to have a Q of 47 - would be about 19 Ohms per coil (remember that there are two coils!) and the sum of the two coils would push feedpoint resistance near-ish 50 Ohms.  The result is that roughly 1 "S-unit" (about 6dB) is lost in the coils alone:  Contacts would still be made, but running a "compromised" antenna (e.g. physically small) that already would be less-efficient than its full-sized counterpart and adding another S-unit of loss doesn't sound like an optimal solution!

Using silver-plated copper "Jewelry Wire" (found on Amazon) to rewind the original loading coils dramatically improved the "Q" (approximately 200) and lowering the Ohmic loss to around 4 Ohms.  The result of this is that rather than something in the 40-50 Ohms for the feedpoint resistance, it dropped to "about 15" Ohms on 40 meters - a VSWR of around 3:1 - and even lower impedance than that (higher VSWR) when I reconfigured the antenna for 60 meters (e.g. added extra screw-together sections, moved the coils next to the feedpoint and added extra "drooping" wires to the ends of the dipole).  At the higher bands (20 meters and up) the feedpoint impedance is close enough to 50 Ohms that one can probably forego the auto transformer at all.

For more information about the "Silver-plated versus Stainless Steel" topic, see the blog entry "Rewinding the Stainless Steel coils with Silver-Plated copper wire on the JPC-7 and JPC-12 antennas" - link.

When a worse VSWR is a good thing!

The first thought when being faced with a higher VSWR on an antenna might be that it was made to be worse - but here is a instance where this is not the case.  As noted earlier, an electrically short antenna like a dipole or vertical can be made to be "longer" (from an RF standpoint) with the addition of a "loading" coil - but the job of the coil is to cancel out the capacitance of the, leaving only the resistive portion of the antenna's feedpoint impedance.

For a full-sized dipole or vertical, this resistance is "close enough" to 50 Ohms (perhaps 35-70 Ohms, depending on the antenna and its environment) to provide a decent load to a modern radio - even one without tuner.  But a very small antenna - where a lot more "coil" is required - will have a lower feedpoint resistance unless your coil is very lossy, as was the case with the stainless steel coils on the JPC-7.  With the lower-loss silver plated coil we (mostly) eliminate it as a lossy component - but end up with a different problem.

With a feedpoint resistance of 13-15 Ohms on 40 meters with the JPC-7 and silver plated coil and its resulting 3-ish:1 VSWR one can "fix" this with an antenna tuner to make the radio happy - and I have done this many times, placing the tuner (an LDG Z-11 Pro) right at the antenna (only a few feet/a meter of coax) but almost all common antenna tuners have quite high losses at these low impedances.

Testing with the cover of the tuner removed, I have noted that one ore more of its toroids in particular will run very warm with just 100 watts of power - Figure 4 shows the inside of this tuner showing one of its toroids discolored because of this.  Fortunately, iron-powder toroids are very forgiving of heating with very high Curie temperatures and other than cosmetic (e.g. discoloring the paint) moderate heating won't have any lasting effects as long as it remains intact (e.g. not cracked) and there aren't problems with (possibly-degraded) insulation between turns of the windings.

The other issue is that the balun originally supplied with the JPC-7 - intended for 50 Ohm operation - also got very warm, and after a bit more than a minute of continuous 100 watts at 40 meters the VSWR would start to rise due to its ferrite reaching the Curie temperature, causing the permeability to drop like a rock:  Essentially, the ferrite would "go away" when it got hot - likely not a problem on SSB or CW, but it might be on "key down" digital modes at full power.  This heating seemed to be more severe at the low impedances (below 20 Ohms) than at 50 Ohms.

Eliminating the tuner

Figure 4:
Inside the LDG Z-11 antenna tuner.  The center
toroid shows evidence of have been heated,
apparently due to matching very low "R".
Click on the image for a larger version.

By definition, we can remove the reactive component of the short antenna with the loading coil:  Its inductance will cancel out the capacitance of the antenna at resonance (which is the very definition of resonance) leaving only a pure resistance.  While an antenna tuner is able to cancel out capacitive and inductive reactance - or just pure resistance - we have a situation where, with a properly-tuned loading coil - we have only resistance and for that we don't need a tuner and we can use just a transformer, to change the impedance from whatever it is to 50 Ohms.

An easy way to do this is with an autotransformer.  This is a device with just one winding and in this case - where we are trying to tune to a feedpoint resistance lower than 50 Ohms - we can feed our power across the ends of the entire coil and tap it at various points along the winding to get our desired (lower) impedance.  For my application, having several taps between about 10 and 40 Ohms (plus the natural 50 Ohm feed impedance) would assure the ability to attain a VSWR of better than 1.5:1 for any purely resistive impedance between 7 and 75 Ohms.

The tyranny of the "electrically small antenna" and efficiency

It's worth noting several things about electrically-small low-band HF antennas - which includes portable antennas like the JPC-7, JPC-12 as well as mobile antennas - and how they interact with common antenna tuners (which an autotransformer is not):

• Any efficient, electrically-small vertical antenna will have a very low impedance once it is resonated:  For example, a "perfect", loss-less 1.5 meter (4.9 foot) long vertical antenna system on 40 meters would have a radiation resistance of about half an Ohm.
• Without losses due to the coils and stainless-steel telescoping rods, etc., the feedpoint resistance of the JPC-7 would, at 40 meters, be in the vicinity of 3-5 Ohms, depending on how many screw-together sections are used (e.g. the longer, the higher).
  
• Any automatic (or manual) antenna tuner that you are likely to ever use for portable operation will have rather poor efficiency when trying to match at lower than 20 Ohms or so - which translates to heat as demonstrated in Figure 4.

These facts - among others - conspire against having a small, efficient mobile antenna for the lower HF bands (e.g. 80-40 meters).  In the real world, losses (coil, antenna wire, ground) will conspire to make the feedpoint impedance much higher than the "less than an Ohm" that the would theoretically be - and any difference between the feedpoint resistance at resonance and the predicted radiation resistance is where most of the power in such an antenna system is lost:  In a typical antenna of this sort, the vast majority of transmitted power is lost in heat rather than radiated.

With significant efforts, it may be practical to get the losses of such an antenna system (which includes not just the antenna, but the series matching coil and ground losses an other factors) down to about 10 Ohms - still far above the 0.5-5 Ohms of our "perfect" antennas in the examples above - but as we know, physics conspires against us as trying to force-feed such an antenna with a tuner will probably put it into the impedance range where it is very inefficient.

It's worth noting that many simple and inexpensive mobile antennas achieve at least part of their "matching" to 50 Ohms simply by being lossy:  Most of the power is simply burned up in the coil.  This method is convenient in that it simplifies the problem with matching and is often accompanied by much wider tuning bandwidth (reducing the need to frequently re-tune when one changes frequency) than with our hypothetical "high efficiency" antenna, but the trade-off is poor efficiency.

Auto transformer for impedance matching

Another way to handle this is to simply transform (pun intended!) the impedance downwards from 50 Ohms - and one way that this could be done is with a transformer of some type - and the simplest of these is one with a single winding, called an autotransformer:  Such transformers are commonly used to match a random wire (9:1 matching to about 450 Ohms) and for end-fed half-wave antennas (49:1 matching to about 2450 Ohms) - but we can also efficiently transform the impedance downwards.  By designing appropriately, this transformer can be made to be very efficient.

It would seem that the use of an auto transformer for matching a low-impedance antenna - such as a low-band mobile antenna on a vehicle - used to be more common decades ago, but has fallen out of favor, possibly due to the easy and cheap availability of automatic antenna tuners:  Devices that do this function include the Atlas MT-1 (see Figure 5) and the Swan MMBX, both of which have a number of low-impedance taps. 

Figure 5:
The Atlas MT-1 autotransformer,  The variety of
taps available provide the possibility of achieving a 1.5:1
match to any resistive loads between 9 and 75 Ohms.
Click on the image for a larger version.

My initial thought was to use a ferrite toroid as the core for the auto transformer.  As a general rule of thumb, a transformer should ideally have an inductive reactance of about ten times that of the operating impedance at the lowest frequency (e.g. 500 Ohms for a 50 Ohm system) but, in a pinch, just three times the operating impedance (e.g. 150 Ohms for a 50 Ohm system) was "OK".  With this in mind I wound 7 turns on an FT140-43 toroid with multiple taps.  The inductance of this arrangement was about 45uH which correlates with about 1900 Ohms at 7 MHz - well above the target inductive reactance but it would have been difficult to achieve the multiple taps needed to attain the impedance steps with fewer turns.

This transformer - wound on ferrite - did not work well at all!  When testing it on the antenna, I could not achieve a sensible match and I quickly realized that the problem was due to leakage inductance of the transformer itself.  An ideal transformer would simply transform the voltage according to the tap's turns ratio, but any practical transformer will place some amount of inductance in series with the supposedly ideal tap, and it was likely this spurious series inductance (which needed only to be a few uH to make it "un-matchable") was totally messing up the attempt to tune the antenna, departing far from the ideal transformer at RF.

Measuring the self-inductance of the Atlas MT-1 confirmed this:  Its end-to-end inductance was about 2uH and the inductances between the taps and ground - the results of these measurements made using my HP-4275A LCR Meter (at 4 and 10 MHz - interpolated at 7 MHz) are as follows:

Tap marking (Ohms)	@ 4 MHz Inductance uH (XL Ohms)	@ 7 MHz (Interpolated) Inductance uH (XL Ohms)	@ 10 MHz Inductance uH (XL Ohms)
52	1.87uH (46.6)	1.8uH (79)	1.74uH (116)
23	0.95uH (24.4)	0.95uH (41.8)	0.95uH (61)
18	0.77uH (18.1)	0.75uH (33)	0.72uH (47)
13	0.61uH (14.3)	0.57uH (25)	0.53uH (35.8)

Figure 6:
The impedances (X_L ) of the taps on the Atlas MT-1 auto transformer versus frequency.

While "about 2uH" of inductance at 40 meters (7 MHz) doesn't fit the "3x reactance" rule-of-thumb (e.g. 79 Ohms X_L in a 50 Ohm system) it will still work OK, acting as a parallel inductance across the antenna - but the important part is that there will be a fraction of the leakage inductance compared to the version with the ferrite core mentioned above:  A small amount of this inductance would lower the resonance frequency slightly, but not disastrously so.

Figure 7:
The auto-transformer, wound on a T157-2 iron-powder
toroid with taps terminated with 2.5mm banana plugs.
Click on the image for a larger version.

Replicating the auto transformer

Rather than reinventing the wheel, I decided to (more or less) replicate the electrical properties of the MT-1 (and similar devices) and for this I chose a T157-2 Iron-powder toroid.  With a target inductance of "about" 2uH I wound 13 turns of 16AWG silver-plated PTFE (Teflon) insulated wire which should, in theory yield about 2.4uH - but when compressed together on the core it yielded about 3.6uH which correlates with about 158 Ohm at 7 MHz -  almost exactly 3x the 50 Ohm system impedance.

As can be seen in Figure 7, taps were placed at 6, 7, 8, 9 and 11 turns (from ground) by scraping the insulation off the side if the wire and tack-soldering wires to it providing impedance taps of approximately 11, 14, 19, 24, 36 Ohms - plus another wire across the 50 Ohm feed for the higher bands:  These impedances resulted from where the turns landed and it was convenient to attach taps rather than from any attempts to obtain specific or precise impedances:  After construction, I labeled the leads with the approximate impedances - for obvious reasons!

I used five taps to allow a selection of an impedance to be able to obtain about 1.25:1 VSWR or better, but if I were happy with just 1.5:1, I could have chosen fewer taps in the manner of the Atlas MT-1 discussed, above.

As the impedance of a tap is related to square relation of the number of turns (e.g. twice the number of turns results in 4x the impedance) there's a pretty simple formula to follow to calculate the impedance of a tap:

Z_tap = (Z_sys) / ((Turns_total/Turns_tap)²)

Where:

Z_tap = Impedance of the autotransformer tap

Z_sys = System impedance (typically 50 Ohms)

Turns_total = Total number of turns on the autotransformer (13 turns in our example)

Turns_tap = Number of turns from the bottom (ground) end of the autotransformer to the tap

In other words:

 Z_tap = (50) / ((Turns_total/Turns_tap)²)

Taking our 13 turn autotransformer as an example, we can calculate the impedance at any turn.  Taking the 8th turn as an example:

Z_tap = (50) / ((13/8)²)  therefore,

Z_tap = 18.9 Ohms 

Or, if you know the desired target impedance and want to calculate the turn on which to make that tap, here's the above formula rewritten to solve for it:

Turns_tap = Turns_total / √(Z_sys /Z_tap)

I also included a "50 Ohm" tap (which is connected at the "top" of the transformer, across all of the windings) so that I could still use the common-mode choke (described below) even when operating on the higher bands (20 meters and above) where the natural impedance was close enough to 50 Ohms that I probably wouldn't have needed the autotransformer for impedance transformation, anyway.

At the end of the flying leads are 2.5mm "banana" plugs - which plug in to the feedpoint of the JPC-7.  These allow the selection of taps on the auto transformer which permits the VSWR to be minimized for those bands for which the feedpoint impedance is significantly lower than 50 Ohms:  A bit of care is required to prevent the "floating" banana plugs from touching each other (or anything else metal) but this isn't actually much of a problem.

Initial testing using a kludge of clip leads, I verified with my NanoVNA that the auto transformer worked as it should (e.g. I was able to attain less than 1.5:1 VSWR on 60, 40 and 30 meters) and almost as important, the tuning with the auto transformer was only slightly different from that using the original balun indicating that the leakage inductance of the auto transformer was not much different than that of the originally-supplied balun.

Adding a common-mode choke

Feeding a dipole (which is a balanced antenna) with coaxial cable has the inherent hazard of RF appearing on the coaxial cable feedline due to the symmetry of the antenna.  Excessive RF on the feedline can result in a "hot" rig - that is, RF energy appearing on the chassis of the radio as well which can result in distortion (RF getting into the microphone) and/or malfunction of peripherals (outboard keyer malfunctioning, USB interfaces crashing, interference to the sound card) and out "in the field" where one may not have an elaborate ground system already, this may be more likely than at home.

Figure 8:
The auto transformer (left) plus a common-mode coaxial
choke (right).  The choke is wound on an FT140-43 ferrite
toroid.  Both toroids are in the foreground for comparison.
Click on the image for a larger version.

The "input" to the auto transformer is simply the opposite ends of its 13 turn winding which would normally be soldered to an RF connector.  Rather than doing that, I soldered it to a 36" (91cm) piece of RG-316 PTFE coaxial cable - the shield going to the "bottom" (ground) side of the auto transformer, insulating the connections with adhesive-lined heat-shrink tubing.  The rest of this RG-316 was wound on an FT140-43 toroid yielding 13 turns using the "cross-over" technique where about half of the turns are wound on the opposite side of the toroid:  This method is said to (slightly) increase the series choking impedance at higher frequencies (e.g. 15 meters and up).

Not having a UHF connector designed for RG-316 on hand, I used a crimp-type PL-259 intended for RG-58.  I stripped more than usual of the jacket from the end of the coax, folding the shield over the outer sheath.  Using some PTFE tubing and part of the jacket stripped from the coax itself I was able to increase the effective diameter of the inner dielectric.  Assembling the cable - remembering to include the ferrule and pieces of adhesive-lined heat shrink - I was able to fold the outer shield over the ferrule after a bit of tugging on it to increase its inner diameter.  At that point, I was able to crimp the ferrule into place, securing the coaxial cable firmly.

Figure 9:
The auto transformer with the common
mode choke on the JPC-7's feed.
Click on the image for a larger version.

Since RG-316 is fairly small (it's the same size as RG-174) - and because the weight of the connecting coaxial cable and the common-mode choke itself would be hanging from the cable - I protected the connector with several pieces of adhesive-lined shrink tubing - using a smaller piece just behind the connector to increase its outside diameter and then a larger piece over the ferrule, onto the previous piece of tubing.

 

Not content with this, I wound several turns of "miniature" paracord (1.15mm diameter) onto the ferrule and tied it securely, feeding both free ends underneath yet another piece of heat-shrink tubing that was then installed over where I'd tied the paracord - taking careful care not to damage the cord when applying heat to shrink it.

These two strands of mini-paracord were then counter-wound over the RG-316 as can be seen in Figures 8 and 9 and were tied to the ferrite core of the common-mode choke such that when hanging, the weight of the connector was on the cord and not the coaxial cable:  I did a similar thing between the core of the auto transformer and the balun to prevent the cable itself from being pulled.

Putting it on the antenna

Figures 8 and 9 shows the combination auto transformer and common-mode choke at the feedpoint of the JPC-7 loaded vertical.  As noted earlier, testing showed only a slight difference in tuning between the lowest VSWR achieved with the original 1:1 balun and the transformer-choke combination indicating that its effect was minimal:  As figure 10 shows, transmitting 100 watts on 40 meters also resulted in only very slight heating of the auto transformer - certainly a much lower amount of signal loss than that which resulted in the heating and discoloring of the toroid in the antenna tuner pictured in Figure 4.

Figure 10:
Thermal infrared view of the autotransformer
(top) and common-mode choke (bottom)
after 60 seconds key-down with 100 watts
on 40 meters.  The temperature of the
autotransformer increased only by about 2F
(1C) while the common-mode choke got about
10F (6C) warmer.
Click for a slightly larger version.

Testing the common-mode choke

 

The efficacy of the common-mode coaxial choke was also verified:  Without it, grasping the shield of the coaxial cable with one's hand would result in slight detuning of the antenna, but with it, there was no detectable effect - and there was no detectable amount of "hot rig" due to the presence of common-mode currents flowing beyond the choke and onto the radio's chassis - even without the use of a counterpoise/ground wire.

 

The presence or lack of effect of the change of antenna tuning when body capacitance is introduced is a simple - but effective - means of determining the presence of RF current on the feedline at the point where it is grasped.  Figure 10 shows that this core heated only minimally - also indicative of low loss.

Does it work?

I have put this configuration pictured in Figure 9 on the air several times since assembling it on 60 through 15 meters.  As expected, the best match on 60 meters (<1.5:1) required the 11 Ohm tap while 40 meters seemed fine with either the 11 or 14 Ohm tap.  20 meters, on the other hand, found the best match using the 36 Ohm tap while 15 meters worked well with either this or the 50 Ohm tap.  Again, the heating of the autotransformer at 100 watts was also minimal on any band - even on the 60 and 40 meters where the losses would probably have been the highest.

Conclusion

The use of an autotransformer rather than an L/C antenna tuner is a time-honored means of matching an "electrically-short" antenna, so what has been presented is nothing new - but it may be "new" to some of the readers.  For a portable antenna such as this, its size and relative simplicity can't be beat as it's far smaller than any antenna tuner that could handle 100 watts at the low impedances that may be presented - and it's certainly lower loss as well!

The only "complication" is that which is already intrinsic to this type of antenna:  As this is a dipole, there are two elements - each with its own coil and telescoping rod making it a bit "fiddly" to tune, something best done with a VNA or antenna analyzer.  With this antenna I keep a card that is marked with the physical locations of the tap positions of the two coils for the various bands:  These are held up to the coil and the sliders adjusted, quickly getting "close" to a match with the analyzer used to do any final touch-ups on the tuning.

* * * * * * *

Related pages:

• For more information about the "Silver-plated versus Stainless Steel" topic, see the blog entry "Rewinding the Stainless Steel coils with Silver-Plated copper wire on the JPC-7 and JPC-12 antennas" - link.
• Analysis, observations about the JPC-7 loaded dipole - link.
• Analysis, observations about the JPC-7 portable vertical - link.
• Reducing RF susceptibility for the HamGadgets "Ultra Pico Keyer" - and mimizing RF issues on portable HF stations in general - link.

* * * * * * *

This page stolen from ka7oei.com

[END]

 

How I prevented QRM to HF reception from my solar and AC inverter at Quartzfest

Figure 1:
White board from Quartzfest!
Click on the image for a larger version

As it happens, I found myself at QuartzFest in Arizona in the latter half of January, 2026 where we set up some banners proclaiming the existence of the Northern Utah WebSDR (link) - but I also scribbled on a small white board the words "QRM-Free Solar is possible - Ask How!".

Between the SDR, this message and the diverse portable HF antennas erected, I have had a lot of conversations over the past several days about these and many other topics, meeting new people and re-acquainting myself with others that I've seen on and off over the past several years of my attending QuartzFest (this is year #4 for me.)

RFI-less solar IS possible 

During the "Solar Walkabout" - an on-foot tour to look at how others camping have deployed their solar panels - I volunteered to have folks look at what I'd set up:  It's nothing obviously special - a glass-panel 200 watt Renogy folding array and another Renogy "flexible" solar array - but there is one major difference:  It does NOT produce HF QRM, meaning that I can plant my portable antennas near my panels and not get any interference on HF.

As I've done some previous articles on this, what I'll present here is mostly a set of links to those articles with a quick overview, but this effectively puts that information in one, handy place.

Let's start with quieting the Renogy solar charge controllers:

Reducing QRM (interference) from a Renogy 200 watt (or any other!) portable solar panel system- Link

Figure 2:
My humble, RF-quiet solar array at 2026 Quartzfest
Click on the image for a larger version.

The main issue with Solar charge controllers is that you have a "dipole + transmitter" situation:  The panels themselves do NOT cause RFI, but the charge controller is effectively a transmitter - especially if it's a PWM and/or MPPT-type - and the legs of the "dipole" are the solar panels (possibly long wires connected to large, rectangular pieces of metal) and another set of wires going to the battery - which also find their way around your RV/campsite via the inverters, DC wires, etc.:  It is no surprise at all that RF finds its way out of these things!  By adding filtering, we are effectively "shorting out" the the RF at the feedpoint of this hypothetical dipole and preventing it from radiating.

To quiet these panels, I added bifilar-wound ferrite toroids - but also bypass capacitors:  The toroids (ferrite) alone will probably knock down the QRM by 2-3 "S" Units, but if you are getting S-9+ interference from your solar, simply knocking it down to S-6 or S-7 when you are in the boondocks - where the natural noise floor is closer to S-1 or S-1 - is still pretty bad!

The key here is adding capacitors in addition to the ferrites and this method is perfectly capable of quieting even the noisiest of solar chargers.  It is also vitally important to put this filtering physically close to the noisy device and use good-quality bypass capacitors. 

Figure 3:
Filtering on the bottom of the Renogy controller
making it RF-quiet.
Click on the image for a larger version.

While the above blog entry showed a modest (200 watt) system, the above can be scaled up for higher-power systems:  Larger wire will handle more current and larger toroids will accommodate it!

RF Quieting a Samlex 150 watt Sine Wave inverter - Link 

Another component of RV/camping with power is the inverter to run mains-voltage devices, and these can be terrible noise sources.  The article above shows how it's possible to make one of these devices completely quiet.  For the older Samlex inverter - which was terribly noisy out-of-the-box, it is now quiet enough that I can power LED Christmas lights from it that are strong from the same mast as the antenna and I get NO RFI (the LED Christmas themselves don't produce QRM).

I was fortunate that there was enough room in the Samlex's case to be able to add this filtering, but it may be added externally as well, provided that the leads are kept short.

What follows below are some methods for quieting UPSs (Uninterruptable Power Supplies).  These are very much like the inverters in an RV in that they produce mains voltage from battery power - and the same problems with RFI occur:

A high-current DC (and AC) noise filter for UPS or RV use - Link

This shows a rather extreme example (an 8kVA UPS) where high currents are involved:  Such would be the case with a kilowatt-class DC-AC inverter or even a large PV system.

Containing RF noise from a sine wave UPS - Link

This article shows the techniques involved in quieting a lower-power UPS, but it also introduces some other components:  Rather than winding your own filter using toroids and wire, you can get "Line Filter" modules from electronic parts supplies (e.g. Digi-Key, Mouser) with brand names like "Corcom" or "Delta" (among many others.)  These are self-contained modules with the components built-in - available in a wide variety of voltage and current ratings - that can do an excellent job of filtering.

 

Completely containing switching power supply RFI - Link

This is an extreme example, but it shows how one might be able to make even the noisiest switching power supply quiet - and this might be important to someone who is trying to get every device in their ham shack - whether it be at home or on the road - quiet.  This method is foolproof in its effectiveness, but it is also likely overkill for many applications, but it discusses the "how and why" these techniques can work.

* * * * * * 

I hope that this helps those who venture out in the wild with their RVs, solar power and battery system and still be able to operate HF.

This page stolen from ka7oei.blogspot.com

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