A 15 (and 10) meter high-pass filter for Field Day

QRM from a transmitter to receivers on lower bands

A friend of mine belongs to a club in a town north of me and he was describing an issue that they've been having for the past several years during ARRL Field Day:  A station on an upper band (e.g. 15 or 10 meters) degrading reception on 20 or even 40 meters when transmitting.  What was needed was something that could be used on both 15 and 10 meters and protect the lower bands (e.g. 20, 40 and 80) meters - and this protection would go the other way, preventing the 15/10 meter station's receiver from being overloaded by transmissions on the lower bands.

Figure 1:
Exterior of the 15 Meter high-pass filter,
built into a die-cast aluminum box with single-hole
UHF connectors on the sides.
Click on the image for a larger version.

First, a bit of background.

The ARRL Field Day event is held on the fourth (but not last) weekend every June.  During this event thousands of clubs and individuals go forth into the wilds to set up and operate an event where they attempt to contact as many stations as they can in a 24 (or 27) hour period.  In the case of club stations - or where multiple individuals are involved - it's very common to have more than one transmitter at a given site.

As the Field Day rules stipulate that all antennas/radios be within a 1000 foot (305 meter) circle it isn't possible to provide much geographical separation between different transmitters.  This separation is important because a transmitter produces a very strong signal and the received signals are very weak by comparison:  Receivers can be easily overloaded by these nearby strong signal sources and transmitters can produce low-level signals on frequencies other than those on which they are operation - ones that are too weak to cause problems under normal situations but when placed in close proximity to a receiver these weak emissions can block out/interfere with other receivers - even on different frequency bands.

Other-band signals can cause problems

The degree to which a transmitter radiates these low-level spurious signals - and that to which a receiver is able to tolerate a very strong signal - depends considerably on the transmitter/receiver itself.  Some high end makes of radios (e.g. Elecraft, Flex ^{Footnote 1}) can be very clean in terms of transmitted spectra and higher-end receivers of all makes may be capable of tolerating a very strong signals - perhaps even in the same band - and this strategy works as long as the potentially-interfering transmitter itself is clean:  If that "other" transmitter is producing noise at any frequency of reception, there's nothing that can be done at that receiver to fix the problem other than to quiet or clean up the errant transmitter.  Meanwhile, even a "good" radio - such as an Icom IC-706MK2G or IC-7300 ^{Footnote 2} or a Yaesu FT-757 - which works well by itself - may not "play nice with others" when immersed in an environment with multiple transmitters and receivers in very close quarters for reasons largely related to their design architecture.

What this means is that if one uses directional antennas (e.g. Yagis or beams) they are often placed north-south of each other and pointed parallel  ^{Footnote 3} so that there is some isolation off to the sides of these antennas - and it goes without saying that antennas of any sort are separated as far as the rules - or the operating space (e.g. park, yard, forest clearing) allows.

Sometimes, this isn't enough:  Interference can result despite the precautions (e.g. transmit/receiver separation) so additional filtering may be necessary.

The use of Band-Pass filters

Barring the ability to separate antennas or place them in each others' nulls, there are other options:  From a number of manufacturers ^{Footnote 4} there are available band-pass filters that - as the name implies - are designed to pass one specific amateur HF band with low attenuation (loss) while offering significant rejection of other bands above and below.  By placing one of these filters inline with the radio and the antenna, it not only will reduce the probability that a very strong signal from another band might overload the receiver (a particular problem with a radio like the Icom IC-7300 and certain models of other radios from other manufacturers) but it also attenuates the broad-band noise ^{Footnote 5} that almost all HF radios produce that can encompass frequencies other than the band on which they are operating.

This low-level interference - often in the form of a white noise (or hiss) is produced by the amplifier stages in the transmitter itself.  Most modern HF transceivers - while equipped with low-pass filters that attenuate harmonics at multiples of the transmitted signal and generally prevent this noise from being emitted on the next-higher non-WARC band - do NOT have an equivalent high-pass filter in them that prevents low-level spurious signals or broadband noise from being output to the antenna on frequencies below that on which it is operating.  What this means is that a transmitter operating on, say, 15 meters, can produce a "hiss" that may degrade reception on 20, 40 or even 80 meters whenever it is keyed up - in this example, depending on how well that 15 meter antenna can radiate such signals and how close the two antennas are to each other.  (I discussed this very problem in an early article of this blog:  Getting the rigs ready for Field Day - Link).

For this reason it is often preferable to use a band-pass filter on every transmitter that is used, for the specific band on which it will be operated:  This will not only protect that receiver from the other bands' signals but also prevent the low-level energy from being emitted on bands other than that on which it is being used.

As an aside:  In some cases simply enabling the radio's built-in antenna tuner - or using an external tuner - may significantly reduce the amount of out-of-band energy that the transmitter emits as well as adding to the attenuation from "other-band" signals during receive. ^{Footnote 6} 

A high-pass filter

While band-specific filters are preferred, my friend presented a case where a high-pass filter (one that blocks signals below a certain frequency) may be appropriate.  In his Field Day environment there has always been a station operating on 20 meters and usually another operating on 40 meters as well - but a third station was available to operate on 15 or 10 meters - depending on propagation conditions.  The problem was that when this third station transmitted, 20 and 40 meters were often degraded - likely by the broadband noise mentioned earlier.

While it would be possible to obtain separate 15 and 10 meter band-pass filters at some expense, I decided on a different approach:  A 15 meter high-pass filter.  This filter - which could be made to strongly attenuate frequencies on the non-WARC amateur bands below 15 meters (e.g. 20, 40 and 80 meters) - it would have the advantage of also being usable on both 15 and 10 meters.  Since it was unlikely that they would have stations on both 15 and 10 meters this strategy seemed sound for their application.

Using the ELSIE program (from Tonne software - link), I first calculated an "N=5" pro-forma high-pass filter using the "Elliptical" (e.g. "Cauer") circuit topology observing that I could get low attenuation at 15 meters and above while achieving more than 40dB on 20 meters and below.  Using the ability of the ELSIE program to do Monte-Carlo type optimizations, I then tweaked the filter topology from a pure Elliptical filter to a hybrid one and this resulted in even better attenuation at 40 meters than the original:  The schematic diagram of this filter is shown below:

Figure 2:
The 15 meter high-pass filter as iterated by the ELSIE program.  The circuit topology - originally
Cauer (Ecliptic) was modified by using simple inductors in sections 1 and 5 and a capacitor at
position 6 and then re-iterated to optimize performance.
Click on the image for a larger version.

As can be seen from the diagram, there are three capacitors in series with the signal path with two inductors directly to ground:  The center inductor is in series with another capacitor, forming one of the "notches" typical of the Elliptical filter topology - and it so-happens that it's possible to tweak the filter so that this notch just happens to land in the middle of the 20 meter band to maximize attenuation there.

Rummaging around in my junk box I found several 500 volt silver-mica capacitors:  For some reason I have a lot of 160pF units, so that was placed at section #2 and three of them were put in parallel for the series capacitor in section #3.  I found a 200pF capacitor for section #6 and I paralleled a 120pF silver mica and an NP0 disc ceramic for that in section #4.

Many people doing homebrew construction seem to intensely dislike toroidal inductors - but while they would be more compact, there is no need to use them here, so large-ish air-core inductors were used.  As the inductors are all "about" the same value (in the 225-300nH range) I wound 7 turns of 17 AWG (but anything 14-18 AWG would do) wire on a 13/32" drill bit for each of them, the precise value being unimportant as their turns would be stretched/compressed while using a VNA to "dial in" the filter response.  As mentioned earlier, I'd added one more inductor from the initial design because the inductors were the cheapest of all of the components (they are just wire!) and they are very adjustable - simply by compressing/spreading the turns which meant that by picking capacitor values that were just "pretty close" to those called out by ELSIE, the coils could be used to tweak the filter's response.  Note in Figure 3 that the inductors that are close-ish to each other are placed at right-angles, or in parallel with each other:  Avoid placing two adjacent coils "end to end" with each other to minimize coupling between them.

Figure 3:
Inside the 15 meter high-pass filter.  Copper-clad PC board
material is used as the backplane (ground) with small pieces
used as "islands" for connection and support points.  The
added capacitor and inductor are those on the right-hand side.
Click on the image for a larger version.

The filter was built on a piece of copper-clad PC board material as a back-plane and ground and small pieces of that circuit board material were cut out to form "islands" - the so-called "Manhattan" construction:  These islands would allow the junctions of the various circuit components to be connected together and with these islands glued to the back-plane and mechanically support the components soldered to them.

The piece of circuit board used as the backplane was sized to fit in the bottom of a die-case aluminum box that I had handy (about 6" x 3.25" x 2" or approx. 15 x 8.3 x 5 cm - but it could have been a bit smaller) onto which I'd installed two chassis-mount UHF connectors:  These connectors were placed rather close to the bottom of the box so that their ground lugs could be soldered to the backplane, providing both the "ground" connection to the copper clad and for mechanical support

Using a VNA, I first adjusted the inductor in section 3 to provide a notch at about 14.24 MHz and then iteratively tweaked the inductors in sections 1 and 5 to provide the lowest insertion loss and lowest VSWR at 15 and 10 meters.  When I was done, the insertion loss was just fine - less than 0.5dB - but the VSWR was about 1.45:1 at 15 and 10 meter so I added two more components (the 30pF capacitor in section 7 and the inductor in section 8 of the diagram) to act as a bit of a "tuner" to improve the match:  In the figure above you can see an inductor (in section 8) that goes to the right-hand UHF connector (6 turns of the same wire as the other coils on a 13/32" drill bit) and a 30pF disk-ceramic capacitor (section 7) between the PC board "island" to which it connects and ground:  With a bit more adjustment of all four inductors this brought the VSWR at 15 and (most of) 10 meters down to about 1.25:1 or better - plenty good enough!  The response of this filter is shown below:

Figure 4:
Insertion loss and VSWR plot of the 15 MHz high-pass filter as plotted by a VNA.
As can be seen, attenuation at 40, 30 and 20 meters is well over 45dB with less than 0.5dB
at 15, 12 and 10 meters:  The VSWR is also acceptably low on these band as well.
Click on the image for a larger version.

Not shown in Figure 3, I later used RTV (silicone) adhesive to stabilize the coils and add support - after tuning, of course:  This reduces the probability of the coils being detuned by the filter being jarred or dropped.  RTV is fairly low loss (at least at HF) and far superior to "hot melt glue" in this case (it's lighter - and it won't melt!) and unlike hot glue or cyanoacrylate (e.g. "Super") glue, it can withstand mechanical shock without breaking loose - even when cold.

This filter should easily handle 100 watts - and the low loss is largely due to the use of silver-mica capacitors:  After all, 500 volt silver mica capacitors - such as those used here - may be found in wide-range antenna tuners made by LDG and the like where they would be exposed to more stress than in the filter.  If you are wondering about the use of the small, disc-ceramic capacitors, they are used in "low stress" parts of the circuit - to "trim" the capacitance to the needed value (e.g. a NP0 ceramic in parallel with a 120pF silver mica to get about 130pF) or used to "tune" the filter as in the case of the 30pF capacitor on the output.  While it might seem risky to use these tiny ceramic capacitors at 100 watts, a quick look at almost any Japanese-made amateur HF transceiver - particularly those made up until fairly recently - you'll find them sprinkled with these capacitors in the low-pass filters and even for matching in the final amplifiers - both at HF and VHF/UHF - for matching:  If it works for them, I'll not worry about using them here in the right places.

Don't forget to change the filter when you change bands!

One hazard with outboard filters of any type:  Be sure that the filter is removed if you attempt to transmit on a frequency for which it is not designed!  After nearly every Field Day I hear/read reports where someone - say, originally on 20 meters - then tries to QSY to another band with the 20 meter filter still inline using the radio's built in antenna tuner or an outboard tuner:  The result is is often that the filter is damaged!  ^{Footnote 7} 

Final comments

As can be seen from the response plot of Figure 4 this filter will attenuate signals on the bands 20 meters and below by more than 45dB and this should be enough to quash to inaudibility any low-level noise produced by the transceiver at these lower frequencies that might degrade reception on these bands.  Similarly, energy from transmissions on 20 meters and lower from other stations will be at a much lower level prior to reaching the front end of the radio using this filter, further reducing the probability that they could overload/cause noise.

One thing that has not been discussed thus far is the fact that harmonically-related frequencies (e.g. a transmitter on 7.05 MHz would have harmonics at 14.10 and 21.15 MHz) are likely to be audible on other receivers, despite heroic attempts to fully-filter them.  The reason for this is that these harmonically-related signals will be fairly strong compared to the noise floor of the amateur bands and, unlike the low-level noise discussed earlier, would have their energy concentrated into a small bandwidth.  

Such signals are also likely to be radiated not only from the antenna ports, but from other cables connected to the radios themselves - namely the power cables, audio connections, data and PTT lines which means that a filter on the output won't suppress those other leakage sources.  Other than wide-spaced separation (e.g. not placing radios in the same location and moving them as far apart as possible) there's no way to completely prevent harmonically-related QRM other than to coordinate efforts and simply avoid operations that could result in harmonically-related interference.

As it is not yet Field Day, I don't know if this filter will "fix" the problem that my friend was reporting, but  should help, and it was quick, cheap and easy to throw together. ^{Footnote 8}

* * * * *

This page stolen from ka7oei.com

[END]

Footnotes:

1. Unlike most radios, Flex radios do include filtering to prevent low-level noise from being emitted on bands lower than the one on which it's being operated:  Specific models of other manufacturers may also include this - although most do not.
   
2. Direct-sampling receivers such as that of the IC-7300 have "different" problems in the presence of very strong signals compared to more conventional superheterodyne receivers:  Any signal that hits the analog-to-digital converter can cause overload, no matter the frequency.  While a conventional receiver can have a very "strong" mixer and some "roofing" filters in its IF (Intermediate Frequency) stages, this is not possible on a direct-sampling receiver.  Instead, it must rely on a rather large number of individual, overlapping band-pass filters to cover its intended frequency range and the ultimate attenuation of these filters may not be "strong" enough to prevent a nearby transmitter on another band from adding to the already-strong mele' of signals on the crowded bands during Field Day and causing overload - or, at least, significant de-sensing (e.g. reduction in sensitivity).  This property is also what almost certainly makes them very poor candidates for being able to tolerate another local transmitter on the same band (e.g. a 20 phone and a 20 CW/digital station at the same Field Day site).  There are strategies that can improve the probability of two stations co-habitating on the same band - mostly having to do with picking the "right" radios (e.g. Elecraft K3S or the K4HD are known to work in this environment as are a few others) - as can the use of parallel-pointed Yagi antennas (see the next section, below) - or very "sharp" band-pass and notch filters can be constructed as described in two articles on this blog, namely:  A 100 watt "Helical" resonator bandpass/notch filters to increase isolation of 20 meter stations during Field Day (link) and Revisiting the 20 meter "helical resonator" band-pass/notch filters (link).
   
3. Being able to point beams parallel to each other is at least partly a matter of geography.  A station on the east coast is likely pointing their antennas west while the situation would be reversed on the west cost:  A station in the middle of the country - with signals coming from potentially all directions - would be less-likely to be able to use this tactic, at least not without a degree of coordination among the individual transmitters/stations.
4. A number of different manufacturers make band-specific filters for HF.  Depending on the design, these can offer modest (>=30dB) adjacent-band suppression - which is usually enough to solve most interference problems - or much higher degrees of filtering, even more than 50dB.  In addition, individual-band "Notch" filters are available from some suppliers that reject a specific band of frequencies which can be used several ways - on a transmitter to suppress any low-level noise that it might be generated on a specific band, or on another station to reduce the levels from a transmitter on that other band to prevent overload - and it can also be used to further-improve performance of a band-pass filter and increase attenuation on that specific band.  One of the companies that supplies such filters is Morgan Manufacturing (link):  Full disclosure - I know the person that runs this company and am quite familiar with the products.  Other manufacturers also make similar, excellent products as well.
   
5. This "hiss" can usually be detected without any sort of special equipment.  To do this, one would set up two transceivers in a relatively RF-quiet location (perhaps NOT a suburban home), each on its own antenna spaced within a few hundred feet/meters of each other.  On the radio doing the transmitting turn down the microphone gain all of the way after verifying that the RF power output would otherwise yield 100 watts peak when talking.  On the receiver, tune in the next band lower than the transmitter and note the noise floor with and without the transmitter keyed up.  In many cases, a "hiss" that can mask weak signals can be observed - particularly if using a resonant antenna on the transmitter without an antenna tuner.  If you couple carefully into the transmitter (using attenuators or directional couplers) this noise floor can be measured directly with a spectrum analyzer - even the $50-ish "TinySA" is up to the task!
   
6. Testing to determine the efficacy of the built-in tuner as a band-pass filters was done using a Kenwood TS-450SAT, a radio from the 1990s.  When the tuner was switched in and "tuned" - even if the load was already matched - it functioned as a low-Q band-pass filter that reduced the broadband noise and adjacent band signals by at least 8dB - and typically 20dB or so.  Whether or not this strategy is likely to work on specific radios (e.g. some may switch out the tuner if there is already a good match) would require testing as described above.
7. The most likely components to be damaged when trying to "force feed" RF on the "wrong" band are the capacitors, followed by toroidal inductors being somewhat less-likely - and this will often happen when transmitting is attempted at full power (100 or more watts) rather than at the low power level used for tuning.  Usually, the operator realizes the mistake after the tuner fails to find a match, or it does find a match but signals are weak or absent.  For this reason, if you are using a filter with a radio and an external tuner it's strongly recommended that you place the filter between the radio and the tuner:  This will prevent damage to the filter as the radio will protect itself if it's used on the wrong band, presumably alerting the operator to the problem!
8. It took far longer to put together this article than it did to design, gather parts, assemble and tune the filter!

 

The construction of a hybrid ring combiner: Using the same duplexer, feedline and antenna for two repeaters/links/transmitters

Two transmitters, one antenna

There are occasions when it is desirable to combine two transmitters - and receivers - onto a single antenna - perhaps at a busy repeater site where more than one link or repeater is required.

The most obvious answer to this would be to have a separate antenna - and duplexer - for each of the links - but there are potential problems with this:

• Antennas are expensive
• Feedline is expensive
  
• Duplexers are even more expensive
• If you are on a shared site, you may have difficulty installing - or even getting permission to install - another antenna - and if you are renting tower space, this "other" antenna will be an ongoing expense.
  

If you have the luxury of planning ahead and you have a bit if extra link margin ^{Footnote 1} (e.g. you have "excess" signal and can afford a bit of loss), by placing two repeaters and/or link radios on adjacent - or nearly adjacent - frequencies, you can probably use a single antenna, feedline and duplexer - potentially saving money and hassle in the long run.  Let's take as an example as a hypothetical pair of full-duplex links at a single site in the U.S. on the 70cm amateur band:

• Transmit frequencies:  421.000 and 421.100 MHz
• Receiver frequencies:   434.500 and 434.600 MHz

As these pairs of frequencies are only 100 kHz apart from each other, they will handily fall into each others' notches in their duplexers meaning that neither transmitter is likely to bother either receiver.  What this means is that you could use a single duplexer along with a bit of extra gear (more on that later) to put both radios on the same antenna.  This same technique could also be applied if, for some reason, you had two repeaters at the same site (say an analog and a digital) as will be mentioned a bit later.

Note:  While this article describes usage on the 70cm band, there's no reason why it could not be applied to other bands - higher and lower - as well.

First, let's briefly cover using a single antenna to feed more than one receiver.

Receive:  Splitting the receive signal path

Receiving is pretty easy:  Just use a splitter.  If you are using a 2-way splitter this will result in 3.0-3.5 dB loss on receive, and there is - in most cases - likely to be enough link margin so that this won't be a problem, but if not, placing a preamplifier of modest gain (say, 10dB) in front of the splitter will overcome these losses:  Avoid the temptation to use a higher-gain amplifier than this to minimize the probability that it might make the problem even worse due to overloading (desensing) and producing intermodulation products that will cause other interference issues.

In a pinch, a 2-way "Cable TV" type of splitter - with appropriate adapters - will work fine for 2 meter and 70cm even though they are designed for 75 ohm system:  They will have reasonably low loss (less than 4dB for a good-quality unit) and a port-to-port isolation of 20dB or better.  For more information see the article on this blog:  Using TV (F-connector) 75 ohm splitters and taps in 50 ohm systems on the amateur HF, VHF and UHF bands (link).

Methods of combining closely-spaced verses more-distantly spaced transmitters   When combining transmitters, there are two general situations that arise:  Those that are very closely-spaced in terms of frequency, and those that are spaced farther away from each other. For two transmitters whose frequencies are separated by a reasonable distance (say, 5-10% of the frequency) it is usually practical to use a combination of notch cavities and phasing lines to separate them from each other - not unlike the way a duplexer allows both transmit and receive at the same time on a given antenna:  There's no reason why one couldn't have two transmitters on a duplexer.  If done carefully, such combination can incur additional losses of only a couple of dB. The problem changes if the two transmitters are quite close in frequency:  Eventually, the separation is so small that it is not possible to use resonant cavities to separate them from each other:  For 2 meters, this spacing is about 400kHz or narrower while on 70cm a spacing of less than 1.5-2 MHz starts to become more difficult.  As the spacing gets narrower and narrower, losses go up and in the case of combining just two transmitters, if the losses exceed about 3 dB, it may be better to simply use a hybrid combiner like those described here rather than potentially large, expensive notch cavities.  If there are more than just two transmitters to be combined, everything gets even more complicated and careful system design and frequency planning are of paramount importance to minimize losses.

You can also find on the surplus market (e.g. EvilBay) splitters - typically made by Mini-Circuits - that are native 50 ohm.  It's worth noting that Mini-Circuits tends to rate their products very conservatively and even a splitter/combiner rated for "only" 200 or 250 MHz at the top end will work just fine at 450 MHz.  One device frequently found is the Mini-Circuits ZFSC-2-1(+) which is rated for 5-500 MHz and is suitable for all repeater bands 10 meters through 70cm.

Transmit:  Combining multiple transmitters onto a single feedline

This is a bit trickier.  

If your frequencies are very near each other (within 1-2 MHz at 70cm, closer than 500 kHz at 2 meters) it's not likely to be practical to use complicated band-pass/notch schemes to isolate the two transmitters with less than about 3 dB of loss.  Again, we are presuming that your link budget will tolerate an additional 3dB of loss - and in most cases in the "real" world, this is likely to be true.

The "easiest" way to combine two transmitters - whether the frequencies are close to each other or not - is with a hybrid combiner.  If the frequencies are quite close together - as in our example above - then the same duplexer, feedline and antenna can be used.

Note:  On 2 meters, frequencies closer than 100 kHz can likely be combined and mutually filtered with a single duplexer/band-pass cavity while on 70cm you may be able to get away with up to 500 kHz of separation - and both may require a bit of careful tuning of the duplexers/band-pass cavities.  Clearly, the closer the better!

Wilkinson Power Combiner

There are two common types of combiners that would be suitable for this - the simplest being the Wilkinson type, depicted in Figure 1.  This circuit is frequently used in power amplifiers when there are multiple gain stages in parallel that are phased identically, being fed from the same source and combined to get more output.  If you are combining different transmitters (at different frequencies) you will theoretically lose half the power - but there's really no way around this.

Figure 1:
  Wilkinson combiner diagram (from Wikipedia)
In a 50 ohm system, this uses 72 ohm line and a
100 ohm resistor between P2 and P3.

How it works

Ports P2 and P3 represent the connections to the two transmitters and P1 is the combined output.  As can be seen, there is 1/2λ (1/2 wavelength) between P2 and P3:  Any signal from P2 will arrive at P3 180 degrees out of phase and vice-versa, the ultimate result being that the signal from P3 is cancelled out at P2 and the signal from P2 is cancelled out at P3, effectively isolating the two transmitters from each other - which is important, as we'll later see.

Note that there a resistor between P2 and P3 which has a resistance of twice the system impedance - or in the case of a 50 system ohms, it should be 100 ohms.  Additionally, half of the TOTAL transmit power will be dissipated by this resistor if the signal sources are dissimilar (e.g. NOT being used to combine two parallel stages in an RF amplifier).  Finding a 100 ohm resistor suitable for 70cm that is also capable of handling 10s of watts of RF with a good return loss is tricky - and then placing it across two terminals that are "hot" with RF - makes the Wilkinson less attractive in this application.

Note that this device requires transmission line that is √2 * Zo - or in the case of a 50 ohm system, it's 50 * 1.414 = 70.7 (e.g. 71 ohms).  For our purposes, "75 ohm" cable would be fine if we were to construct one.

Hybrid ring combiner

Another type combiner that will fit the bill is the so-called "Hybrid Ring" (a.k.a. "Rat Race") combiner, depicted in Figure 2.

Figure 2: 
Hybrid Ring Combiner (from Wikipedia)
a.k.a. "Rat Race" combiner.  Like the Wilkinson,
this uses 72 ohm feedline in a 50 ohm system, but
the terminator (usually at P4) is 50 ohms.

Referring to Figure 2, our two transmitters are connected to ports P1 and P3 while output P2 contains the sum of the P1 and P3 signals and P4 contains the difference of the P1 and P3 signals.  All ports must be sourced/terminated at 50 ohms and we would typically connect 50 ohm transmitters at P1 and P3 and 50 ohm loads (antennas or dummy loads) at P2 and P4 as appropriate.  

If we look carefully, we can see that between P1 and P3 (via P2) there is 1/2λ of feedline between the two if we go clockwise around the circle - but if we go the other way around the circle we have 1/4λ plus 3/4λ - or a total of 1λ - which which means that there is 1/2λ difference in feedline length between them - thus a 180 degree phase difference - any RF being input to P3 are cancelled out at P1 and vice-versa.  With P2 being halfway between P1 and P3 - and going the other way we have 1.5λ of feedline between the same points and no cancellation we get the sum of the two transmitters..

If we were combining two in-phase RF amplifiers from the same transmitter - as we might if we had two identically-phased amplifiers that we wanted to combine for more power, we would get zero power at P4 and the combination of the two amplifiers would appear at P2:  A dummy load - connected at P4 - would see little or no power if the two two amplifiers were operating with equal power and phase.

In our case, we are combining two different transmitters - on different frequencies - and this means that half of our power will end up each on P2 and P4, so we would typically connect our dummy load at P4 and our output to our antenna (via the duplexer) at P2.

Figure 3:
Andrews FSJ1-75 75 ohm Heliax - unfortnately
no longer being made, but other types of 75 Ohm
cable may be used as well as described in the text.
Click on the image for a larger version.

Like the Wilkinson power combiner, the ring combiner requires transmission line that is √2 * Zo (e.g. 71 ohms) - and again, 75 ohm cable is fine for most purposes in a 50 ohm system.  A distinct advantage of the ring combiner is that unlike the Wilkinson, the dummy load in the ring combiner is ground-referenced rather than having a resistor across two "hot" RF terminals - simplifying design.  Additionally, because the load that we would connect to P4 is the same as our system impedance - 50 ohms - finding a suitable device on the new or surplus market is quite easy.

Either the hybrid-ring or the Wilkinson combiners should yield well over 25dB of isolation between ports when properly constructed over a wide frequency range (8-10% or more) - and values of over 50dB are attainable on the workbench with known-good loads and sources across narrower frequency ranges.  Having good isolation between ports is vital when combining transmitters:  Not only does good isolation imply lower loss  (e.g. closer to the theoretical 3dB) but in the case of two different transmitters, minimizing the amount of energy that transmitter "A" gets from transmitter "B" reduces "wasted" power due to this cross-coupling, but more importantly it reduces the likelihood of the generation of IMD (Intermodulation Distortion) products that could cause receive performance degradation and interference - more on ways to prevent this will be discussed later.

Choice of transmission line

As noted, the impedance of 75 ohm feedline is "good enough" for these two types of combiners on a 50 ohm system in most cases and this sort of coaxial cable is readily available:  RG-59 and RG-11 types are suitable - as are some RG-6 cables such as Belden 1694A that have tinned-copper shields and solid copper center conductors to which we can easily solder - but note that most RG-6 cables use aluminum shields and copper-covered steel center conductors:  In theory these would work, but making a good connection to the shield is a complication.

NOTE:  There are PTFE (Teflon) 75 ohm coaxial cables available - see Footnote 2 at the bottom of this article for more information.

Figure 4: 
A hybrid-ring combiner, constructed as a "ring" using
FSJ1-75 Heliax and "N" type connectors on pigtails with bits
of brass tubing at the joints.
Click on the image for a larger version.

While I had all three of the typical 75 ohm cable types available to me, I found - while rummaging around - some Andrews FSJ1-75 1/4" Heliax ^tm coax.  Unfortunately, this cable hasn't been made since 2011, but it may still be found on the surplus market occasionally - and the data sheet for it is still online (link).  This cable is lower loss than any of the other options mentioned - but this is probably not important as such short lengths of it are used:  Reactance losses due to the construction itself are likely to be greater than the cable losses, anyway - and I would have gotten very similar results with the other choices.  The use of the smaller cable also implies a smaller bending radius - which will be important as we'll soon see.

In theory, one could construct a hybrid ring like this using RG-11-type coaxial cable, N-type connectors and known-good ^{See Footnote 3} N-type "tee" connectors - but you will have to carefully calculate the added lengths of the connectors and adapters when doing so.

Form factor

There are several ways that this ring combiner could be built.  The most obvious is in the form of an actual ring as depicted in Figure 4 - a combiner that I built over 25 years ago, also using the same Andrews FSJ1-75.  This uses short pieces of brass tubing to connect the segments together and holes in the sides of the tubing allow solder connections to be made between the segments and to the short "pigtails" with "N" connectors on them:  It measures better than 35 dB TX port isolation at its intended frequency.   This particular combiner had been used to combine a two UHF repeaters - an FM voice repeater and a 9600 baud packet repeater, on frequencies just 25 kHz apart - onto the same duplexer/antenna for several years with excellent results using the methods described below. 

Figure 5: 
For the ring combiner on this page, it needed to be a bit more
compact and rugged, housed in a Hamnmond 1590D box.
Click on the image for a larger version.

For the combiner that I recently built there was the need for something more "compact":  The "ring" described above (in Figure 4) is about 10" (25cm) in diameter and the connectors are awkwardly spaced, on flexible leads - and the entire thing is a bit fragile.

 

For 70cm, the feedline lengths are short enough that the sections can be put into a large-ish die-cast box - particularly if the cable used has a fairly tight allowed bending radius, which would rule out RG-11.

 

In this case I had on hand a Hammond 1590D which is 7.4" (18.8cm) x 4.7" (11.95cm) x 2.2" (5.6cm).  I also happened (as one does) a number of chassis mount "N" connectors with short lengths of UT-141 (hardline) already connected to them which would allow (literally!) some flexibility as to how the internal phasing sections of feedline could be oriented and connected.  It's worth nothing that if RG-179 were used (see Footnote #2, a smaller-still metal box could be used - even for a 2 meter combiner!

 

Running the numbers

First, calculate the wavelength at the desired center frequency.  In our case, we need 421.0 MHz:

300 / F (MHz) = Wavelength

300 / 421.0 = 0.713 Meters - The wavelength at 421.0 MHz.

Figure 6:
Preparing and measuring the cable pieces.  For the ring
combiner there is one 3/4λ and three 1/4λ pieces.  They are
intentionally cut slightly long to allow stripping and soldering.
Click on the image for a larger version.

Now, we need to find the velocity factor of our particular cable, a number which indicates the tendency for electricity to move slower than light when it is conveyed through conductors in the presence of a dielectric.  For best results, find the manufacturer of the specific cable that you are using as this varies - particularly between solid and foam dielectric cables and the precise type of dielectric.  For our FSJ1-75, the stated velocity factor is 0.78 (78%).  Knowing this, we can calculate the length of one electrical wavelength of our cable:

Vf * Length = Electrical length

0.713 * 0.78 = 0.556 Meters

Figure 7: 
The stripped/prepped end of a cable segment as described in
the text.  The calculated lengths are measured between the
ends of the dielectric, where the center conductor protrudes.
Click on the image for a larger version.

As we can see from the diagram of Figure 2, we need four pieces of cable:  One that is 3/4λ (0.75λ) and three more that are 1/4λ (0.25λ), so we multiply the above to get those:

0.556 * 0.75 = 0.417 Meters for 3/4λ section  - approx. 16-3/8"

0.556 * 0.25 = 0.139 Meters for the 1/4λ sections  - approx. 5-1/2"

Prepping the cable

As we need a bit of extra cable to expose the center conductor to which we solder we need to make the sections slightly longer - about 1/2" (13mm) at each end - or about 1" (25mm) longer overall for each piece.  After cutting the pieces of cable, put a mark at the center of each and measure about 1/2" (1cm) less than half the length that we calculated above to the end strip back the outer jacket:  This will leave a section of bare shield to which we can later solder.  Now remove the shield at a point about 1/4" (0.5cm) less than half the length that we calculated above and remove the dielectric, but leaving a couple of mm (about 1/16") out from the end of the shield:  And example may be seen in Figure 7.

Note that when we calculate the length of the cable - 0.139 meters for a 1/4λ section in our example - we are measuring at the points where the center conductor protrudes from dielectric.  As can be seen in Figure 8, the center conductors are then bent over level with the top of the dielectric to be soldered to to the other cable segments and the distances are measured from the points where they are bent over.

Figure 8:
The junction of the three cables, with the center conductors
bent over and soldered - and with the shields firmly soldered
together as well, using 26 AWG bare wire during assembly.
Click on the image for a larger version.

At this point it would be a good idea to consult the specifications of the cable that you are using and determine the minimum bending radius.  For FSJ1-75, this is specified as being 1.25" (31.75mm) and for Belden 1694A this is about 2.75" (70mm):  Try to bend it less than this if you can:  Cut a piece of cardboard or paper with a circle of this radius as a comparison.

To get an idea as to how everything should be routed inside the box, the connectors+cables were first installed and tightened and the four pieces were laid out and moved about to determine what made sense.  As can be seen from Figures 8-11, each of the cables were bent at a gentle right angle so that the center conductors all came together at one point and the shields in parallel with each other - all without bending the cables at too-tight a radius.

Once the orientation of a cable end was determined, the center conductors were bent over - leaving a gap of about 1/16" (2mm) between it and the shield and the connections tacked together.  After this, some tinned 26 AWG wire was wound tightly around the shield and then soldered, making a both a very short-length electrical connection and providing mechanical rigidity as can be seen in Figure 8.

Figure 9: 
A better view of the cables being assembled in the box,
showing how the 26 AWG bare wire first used to tie the
springy cables together and then flooded with solder.
Click on the image for a larger version.

A very hot soldering iron is a must here as it allows connections to be made very quickly and prevent melting of the dielectric.

 

If you have it, use tin-lead solder (e.g. 60/40 or 63/37) as it melts at a lower temperature than "lead free" solder and is less likely to melt and damage the cable's dielectric. Additionally, a few drops of either "no-clean" or rosin flux will help make good, clean joints with minimal heat.

As it turns out, the FSJ1-75 is relatively forgiving in terms of heat if you use a hot iron and work quickly - but if you use something like 1694A or other flexible coaxial cable, you may want to tin the braid prior to assembly - starting with a few bits of "practice" coaxial cable to avoid melting pieces that you have already cut to length and prepared.

Figure 10:
The cable segments were originally fitted in the box and
tacked together as seen in Figure 9, but were carefully removed
so that the connections could be fully soldered on both sides.
Click on the image for a larger version.

Testing

A NanoVNA is a good tool to test the combiner - but you will also need TWO known-good 50 ohm terminators (dummy loads).  For this, you probably won't want to use anything but good-quality units - which are available surplus - and you will want to use an ohmmeter to verify that they are in the range of 50-52 ohms - and don't forget to take into account the resistance of your ohmmeter leads!

Before proceeding, be sure to do an "SOLT" (Short-Open-Load-Through) calibration of your NanoVNA for the frequency range of interest.  In our case, it's 400-500 MHz.   As you will be measuring "through" loss, I suggest that for measuring this device that you use the "receive" (second) port of your NanoVNA instead of the termination when doing the SOLT calibration as it will likely not be quite as good as the load that came with the NanoVNA and would otherwise make VSWR/return loss measurements "appear" to be worse than they are:  If you are interested in single-port (S11) measurements only, use your SOLT load for calibration and either it or a known-good load when testing..

Referring again to Figure 2, note which of the four connectors correlate with P1, P2, P3 and P4 (mark them with a pen or label) and connect the dummy loads to P2 and P4.  Then, connect the "In" and "Out" leads of the NanoVNA (or similar) to P1 and P3 - it doesn't matter which way.

Figure 11: 
Initial testing of the combiner, showing the NanoVNA
connected to ports P1 and P3 and the dummy loads on ports
 P2 and P4.
Click on the image for a larger version.

If all goes well, the VSWR should be below 1.25:1 and the "through loss" (which is the isolation between the two transmitters to which P1 and P3 will be connected) should be well above 25 dB.  At some frequency - probably a bit above the intended design frequency you may see the through loss dramatically increase (a "dip") in port-to-port isolation between P1 and P3.  If you do not see significantly more than 25 dB of isolation between P1 and P3, re-check your connections:  If you see way under 20dB you have probably misidentified your connections and should check again - but if you think that you have properly identified everything, re-check your math from when you calculated the cable lengths - and don't forget to include the velocity factor.

 

As we'll see later, a device like this should yield in excess of 30dB transmit-transmit port isolation over a very wide frequency range - even without any "adjustments".  In Figure 11 you can see the initial testing of the ring combiner and in the background on the NanoVNA - on the blue trace - the "dip" showing the frequency of the best port-to-port isolation - which, before tuning, was near 435 MHz.

Having verified that the isolation between P1 and P3 is good, connect one of the NanoVNA leads to P2 and move the dummy load onto the to which the NanoVNA had been connected:  If the unit is working properly, the insertion loss will be about 3dB, which is exactly correct.

Figure 12: 
A "tuning strip" used to make slight adjustments.  The
"grounded" strip is moved closer to the exposed center
conductors to increase capacitance.
Click on the image for a larger version.

Making adjustments

While the "ring" depicted in Figure 4 is elegantly simple, it can't be adjusted:  Scrunching it into the box as shown in Figures 10 and 11 allow a bit of tweaking to optimize both isolation and matching by virtue of the exposed center conductors.

When I built the combiner pictured I noted that the best insertion loss between P1 and P3 appeared as a "dip" around 435 MHz with the lowest VSWR occurring around 440 MHz.  I observed that very lightly touching my finger at the junction of P4 caused this "dip" to shift down in frequency, indicating that a very small amount of capacitance might better things at our 421.0 MHz design frequency.

Rather than connect a small variable capacitor - which would need to be of very low capacitance values (possibly less than 1pF) I did something simpler:  I cut a small strip of brass sheet and soldered one end to the shield of the coax at the junction of the cables as depicted in Figure 12.  This strip was then bent around the exposed junction - but kept at 1/8-1/16" (2-3mm) away from the center conductors to avoid shorting:  This added a bit of capacitance at that point and shifted the "dip" down in frequency, and by adjusting this brass strip closer/farther away from the center conductors of the cables, I was able to "dial" it in at 421 MHz.  If you can't get the strip close enough to the center conductor to bring the frequency down, solder a smaller strip of metal to the center conductor to form a larger capacitor plate, allowing more capacitance with greater distance.

Figure 13:
Inside the combiner - tuning strips installed on P1 and P4
which are used to slightly tweak the performance at the
intended operating frequency.
Click on the image for a larger version.

Similarly, I noted that lightly touching the connection at P1 reduced the VSWR slightly (which had started at around 1.2:1) and a similar brass strip was installed there.  The two interacted slightly with each other and I was able to get both a lower VSWR (under 1.1:1) and very good isolation (more than 50dB) at 421 MHz.  Note that when connected to the "real world" (e.g. actual transmitters and an antenna) which will probably have some reflected power and will NOT likely have precisely 50 ohms of impedance, the isolation will likely be less than what we measure on the workbench.  The lack of a really low VSWR (e.g. higher return loss) anywhere across the frequency range can likely be attributed to the fact that we used nominally 75 ohm feedline for the construction of the combiner rather than "70.7 Ohm" feedline - which is difficult to find in the form of coaxial cable!

With a bit of finessing I was able to get the port-to-port isolation and VSWR shown in the plot below:

Figure 14:
Port-to-port isolation (blue line) and VSWR (green) with a Smith chart in the background
and a legend in the upper-right corner showing the readings at the marked frequencies on the plots.
There is well over 50dB of isolation between P1 and P3 at the intended frequency.
Click on the image for a larger version.

As we can see from Figure 14, the port-to-port isolation at our target frequency is quite good - in excess of 50 dB - while the VSWR is around 1.1:1 as noted in the upper-left corner.  It's worth noting that the port-to-port isolation is better than 30dB between 400 and 445 MHz - a span of about 10% - and it's better than 20 dB from somewhere below 400 MHz to 491 MHz - and no-where on this plot does the VSWR exceed 1.4:1.  What this means is that if your goal is 30dB isolation, the design is quite forgiving.  As can be seen from the Smith chart in the center, the matching is pretty well-behaved.

What about insertion loss?  This plot gives us the answer:

Figure 15: 
TX port to antenna insertion loss (blue) and VSWR (green) - again with a Smith chart in the
background and the legend in the upper-left corner.
This plot shows just 0.05dB above the theoretical at the 421 MHz design frequency and a
good VSWR as well.
Click on the image for a larger version.

This graph shows the measured insertion loss and at our target frequency of 421 MHz, it is around 3.05 dB - very close to the theoretical 3dB loss and getting very near the uncertainty of our measurement.  Over the same "30dB port-to-port isolation" frequency range that we measured above (400-445 MHz) we see that the insertion loss is lower than 3.2 dB and that the VSWR over this range never exceeds 1.2:1.  Even up at 491 MHz where we had only about 20dB port-to-port isolation the insertion loss is a bit over 4 dB with a VSWR of 1.4:1 - not great, but still usable in a pinch.

Putting it into practice

Figure 16:
A two-stage UHF isolator with loads - a Ma-COM 7R011,
owned by the author, designed to work at UHF ^{Footnote 5}.
Although the two loads are only rated for 12 watts, the
 unit itself can handle 125 watts if the load on the left
(nearest the "output") were sized accordingly.
Click on the image for a larger version.

Being able to bash two transmitters on "nearby" frequencies together is a good thing - but we do need more than just the combiner to do so effectively and "cleanly".  (Note:  If the transmitters' frequencies were spaced farther apart, a different scheme would be required - see Footnote #4). 

 

The entire point of having a ring combiner with good port-to-port isolation is to minimize losses - here, we limit them to about 3dB - but it also prevents the RF output power of one transmitter from getting into another where mixing can occur, producing low-level IMD (intermodulation distortion) products that could cause interference to your own gear and - more importantly - other users on site.  As the port-to-port isolation is likely to diminish with "imperfect" sources and loads, we need to do more to prevent these mixing products from occurring than just the combiner.

To minimize the probability of this occurring it is also a good idea to install a ferrite isolator on the output of each transmitter, prior to the ring combiner as depicted in Figure 18.  These devices - one of which is pictured in Figure 16 - can be through of as a sort of "diode" for RF:  They will let transmit power go through them, toward the antenna, but any power that comes the other way - reflected due to VSWR or RF energy from another transmitter - will end up in its dummy load(s) - effectively preventing RF from getting back into either of the final amplifiers:  Figure 17 shows what a properly-functioning isolator does to block RF coming back from the "load" port - in this case it reduces that energy by around 60dB at the frequency to which it is adjusted.

Figure 17:
The reverse (isolation) plot of the 2-stage isolator of Figure
16.   At the frequency to which it is tuned (450 MHz)
it has over 60 dB of reverse isolation. In the other direction
(not shown in this plot) its loss is only about 1dB.
Note that +/-25 MHz from center, the isolation drops below
30dB which is why a pass cavity is suggested!
Click on the image for a larger version.

These isolators (which are "circulators" with included "dump" loads) generally come in two flavors:  Single stage and double stage - the latter having two devices in the same package like the one in Figure 16.  Typically yielding 20-30 dB of reverse isolation per stage, a double stage device typically has between 40 and 60 db reverse isolation as shown in Figure 17.  This plot also shows something else:  While the reverse isolation is very good at its tuned frequency, it decreases as you move away:  If you have other users on site - and you are not using a pass cavity between the isolator and antenna - these "off-frequency" signals will not be as well-attenuated by the isolator as at the design frequency and its efficacy will be reduced while the probability of IMD being generated in the transmitter's output will increase.

 

While isolators can be quite expensive when new, they are frequently found on the surplus market.  Most isolators are tunable, and this will need to be done to optimize performance on your operating frequency - but this can easily be done with a NanoVNA and instructions as to how to do this may be found on the Repeater Builder web site (link) as well as on videos on YouTube.  Even the extra expense of isolators and a band-pass cavity may well be less expensive than and preferable to the installation of two separate antennas, feedline and duplexers - particularly if another antenna were to incur recurring costs such as maintenance and tower rent!

 

If your combiner had a rather "average" 30dB of port-to-port isolation and you were using a two-stage isolator on each of your transmitters this would mean the RF energy from one transmitter getting into the other would be down between 70 and 90dB - and this nearly guarantees that IMD products will be extremely low and likely undetectable.

To further reduce the probably of harmful IMD product escaping your system, a band-pass cavity should be inserted between the output of the combiner and the "transmit" port of the duplexer as depicted in Figure 18:  As noted in another article on this blog (See the article:  When Band-Pass/Band-Reject (Bp/Br) duplexers really aren't band-pass - link) almost all duplexers used in amateur repeater service will NOT offer much filtering once you move a few MHz away from their tuned frequency. 

Figure 18: 
A typical application using a ring combiner, along with isolators and a band-pass cavity.
The isolator on TX #1 - and the ring combiner - minimize the amount of energy that it "sees" from
TX #2 and vice-versa while the band-pass cavity limits off-frequency energy that can enter the system.
Click on the image for a larger version.

 

By including a band-pass cavity, energy that is away from the operating frequency will be attenuated significantly.  This is important as many isolators have a rather limited frequency range over which they are most effective as seen in the figure above.  Additionally, if there are very low-level IMD products produced by mixing within your two transmitters even with isolators and the port-to-port isolation of the combiner, these will be significantly quashed by the band-pass cavity, practically eliminating even the possibility of self-generated interference.

 

Figure 19:
The completed UHF ring combiner.
The labeling on the sides of the box
identifying the ports is not visible in this photo.
Click on the image for a larger version.

Conclusion:

With a bit of planning and foresight is is possible to combine multiple transmitters - or even full duplex link radios or repeaters - onto a single antenna.  The techniques described here are most useful if you are able to do frequency planning by placing the transmit/receive frequency pairs quite close to each other, permitting the use of a common duplexer and single band-pass cavity in the TX and another in the RX leg.  Even if the frequencies are separated a bit and a common duplexer is not possible, these techniques can still be adapted.

 

In most cases, the extra 3dB of loss on receive and transmit can be ignored as such a reduction in sensitivity radiated power is likely to be unnoticed - and it well may be worth the trade-off in terms of minimizing infrastructure and even cost.

 

* * * * *

Footnotes:

1. Link margin refers to the amount of signal between a transmitter and receiver - and specifically, the degree to which it could degrade and still produce acceptable results.  Consider the following:  Let us suppose that your goal is to have a link with 20dB SINAD or better and your receiver was capable of producing a signal of this quality with -110dBm at its input terminals.  If the strength of your receive signal from the other end of the link was -90dBm, it could be reduced - via fading or other path degradation - by 20dB before it would be considered to be "faded".  If - in the process of putting two radio systems on the same antenna you were to reduce either the transmitter at the far end or the receiver at the near end by about 3dB, in our example we would still be left with about 17dB of "fade margin".
   
2. RG-179 and RG-302 - both being 75 ohm available with PTFE (Teflon) and similarly-resistant jacket - would be excellent choices in the construction of a ring combiner owing to their flexibility and heat resistance.  RG-179 is available from  both Mouser Electronics - link -and Digi-Key Electronics link at the time of writing and while rather expensive, it doesn't take a lot of cable to construct a ring combiner.  While the PTFE is preferred, it's also available with other types of dielectric/jacket which will work, but care would be required to avoid melting it during soldering.  The PTFE cables' velocity factors are typically around 69% while the polyethylene versions are around 79% - but always check the manufacturer's data sheet!
   
3. If you ever use "Tee" type adapters, be extremely careful what you get - particularly if you are using type "N".  Many "foreign-made" N-type connectors are very poorly built - both mechanically and in terms of RF - and some of them having been found to use springs to make contact (highly inductive - VERY BAD!) and/or compression-type connections that tend to oxidize.  If you can afford it, such "Tee" connectors from Pasternak (link) will likely be good - as are genuine old mil-spec devices from reputable manufacturers (Amphenol, Kings, etc.).  If you don't know their quality, be prepared to measure them carefully using a VNA - and better yet, buy an extra so that you can cut it apart and see for yourself if it resembles anything like a "constant impedance" device with solid, reliable construction.
4. Note that since in our example the two transmitters' frequencies are very close together (only a few hundred kHz at 70cm) there would be little point in putting a band-pass cavity between the output of the isolator and ring combiner, but if the transmitters were several MHz apart - a spacing at which the cavities would offer reasonable rejection - it might make sense to do so.  If one did have several MHz spacing, it's less likely that the band-pass cavity in Figure 18 would be appropriate and that one could get away with using a single duplexer, either. 
5. The two-stage UHF isolator in Figure 16 was bought by the author "as is" - with the pair of 12 watt loads - at a swap meet for about $20 - the low price being due to the fact that it had clearly been submerged in water for a while and was showing some corrosion.  It was very carefully being disassembled and thoroughly cleaned - which included washing the trimmer capacitors with denatured alcohol to remove any contaminants - and resoldering the internal connections as it was reassembled, it once again worked, more than meeting the manufacturer's specifications for both reverse isolation and forward insertion loss.
   

* * * * *

This page stolen from ka7oei.blogspot.com

[END]

Refurbishing a CIR Astro 200 HF amateur band transceiver

The Astro 200

The CIR Industries Astro 200 is a compact, synthesized 100 watt HF transceiver from the mid-late 1970s that covers the 80, 40, 20, 15 and 10 meter bands.  Intended for both home and mobile use, it is quite small - 9.75" wide, 12.5" deep and 3" tall (24.8 x 31.8 x 7.6cm) - including the rear heat sink.  Back in 1977 - when this unit was made - it seems to have cost around $995 for the version without the CW filter - about $5000 in 2025 dollars!

Figure 1:
The front panel of the CIR Astro 200.  While advanced for
its day, the radio is pretty simple by today's standard.
The lack of a tuning knob seems a bit odd.
Click on the image for a larger version.

I don't know too much about CIR Industries, except that it was around only for a few years, apparently absorbed by Cubic-Swan in about 1978 where it was rebadged with the name of the new company and - with very minor changes - became the "200A".  The history of Cubic-Swan becomes a bit muddy after the early 1980s and appears to have fizzled entirely by the mid-late 1990s.  Much of the design of the Astro 200 - and other Cubic-Swan radios - was apparently done by Don Stoner, W6TNS (who was also the "S" in SGC).

The later version of this radio, the Astro 200A, sported a 6 pin round microphone connector, black knobs, slightly different switches, a lighted meter and very slightly modified scales on the meter itself:  I suspect that the electrical differences - some of which are noted below - may have evolved during the production of the original Astro 200.

The radio's history

This unit was purchased new in 1977, with the extra-cost CW filter option, and owned by a friend of mine, having first resided in his International Scout II - and then his Jeep CJ-7 - until about 2020 (when it was removed during vehicle maintenance) seeing many hours and miles bouncing around rough, 4WD roads.  Despite having banged around for about 40 years in a vehicle, it's in remarkably good physical shape, the case having only a few minor scratches.  Unfortunately, my friend became a silent key in 2022 and the radio ended up in my hands.

A "unique" radio

The advertisements for this radio tout it as being the very first completely synthesized amateur transceiver:  Whether or not it's actually the "first", I can't be sure, and this can vary depending on what you mean by "synthesized" - but in this case the local local oscillators are referenced from a single crystal while the BFOs were independent - a common practice even into the early 2000s.  Being an early synthesized radio, it does have a few interesting quirks:

• There's no tuning knob.  Tuning is accomplished by a pair of "up/down" momentary toggle switches.  At first, this seems awkward, but one can quickly become adept to tuning a radio this way.  My friend (the one who'd owned this radio) noted that this tuning method was more convenient when bouncing about on a bumpy Jeep road than trying to use a conventional knob.
  
• Operating the "fast" switch moves the frequency up/down by about 20 kHz/second after a brief pause.
• Operating the "slow" switch moves the frequency up/down about 400 Hz/second after a brief pause.
• A brief up or down push-and-release of either switch moves the frequency by 100 Hz.
  

• 100 Hz tuning steps + Fine Tuning.  The radio tunes in 100 Hz steps, but it has a "Fine" tuning knob that moves the frequency up/down by a bit more than +/-65 Hz to allow one to get the frequency as close as you wish.  With the tendency for most amateurs these days to set their radios to an integer number of kHz (and occasionally to "0.5", 100 Hz steps are just fine and this control can be left centered most of the time.
  

• The synthesizers are a bit slow to lock.  As one tunes the radio - particularly in the "fast" mode - the synthesizers may take a second or so to catch up as it "swoops" in onto the correct frequency.  This also means that after power-up, the radio is unusable for about 30 seconds, or for up to 15 seconds after changing bands.  As the synthesizers "land" within about a second during normal tuning with the up/down switches, the radio is on frequency by the time normal human reaction time has "locked in" to what is on frequency.
  

• The "WWV" mode.  You'll note that the mode switch includes a "WWV" position.  This is actually a completely separate, direct-conversion receiver - with no AGC - that is tuned only to 10 MHz. Since it uses the (doubled) 5 MHz reference as its local oscillator, it provides an easy way to check/set the radio precisely on-frequency.
  

Despite having a digital readout and a synthesizer, it does not have a computer of any sort.  "Programming" is done using PROMs (Programmable Read-Only Memory)  to look up the synthesizer tuning information and "74LS" type logic as counters for the frequency dividers and tuning - but this also means that when it's first powered up, it always defaults to the bottom edge of the band to which it is tuned.  This is a bit of an inconvenience - but in the mid 1970's, prior to inexpensive single-chip microcontrollers with onboard program memory along with affordable development tools there was no real way around this without adding significantly to complexity and cost.  I'm looking into a simple way for the radio to "remember" the last-tuned frequency on each band - perhaps the topic of a later article.

About this radio

Figure 2:
The radio's tag - Serial #8, apparently!
Click on the image for a larger version.

This radio is apparently a very early production unit - somewhat different from that depicted in the manual:

• The Microphone connector is a standard 1/4" TRS (headphone) jack rather than a 6-pin round connector apparently used later in the production run and in a later revision, the 200A.  The additional pins on the 6 pin connector provide up/down tuning and 11 volts, allowing one to do tuning via the microphone.
  

• It was lacking the "ANL Board".  This is a very simple circuit circuit (two pairs of back-to-back diodes and an electrolytic capacitor) that reduces, according to the manual, "excessive popping or AGC pumping".  As this circuit is very simple, it was trivial to add to this radio and this somewhat reduced the tendency for the receiver to be momentarily deafened when changing modes or bands.
  

• Upon inspection of the PA (Power Amplifier) module I noted that the driver transistors were Motorola, marked with "604/438 Sample" which further implies an early production radio.  The PA transistors themselves - which are shown as being of type MRF454 in the service manual - were CD3435 made by CTC. 
  

• There are a number of doubly-balanced diode-ring mixers used throughout.  Based on the manual and photos of other units, these seem to be implemented with some sort of mixer module.  On this unit the, modules are not used as the corresponding areas on the PC boards are populated with a pair of trifilar transformers and individual diodes comprising the mixer.
  

• The serial number of this radio is "706008".  Based on other photos that I've seen online, this is apparently serial number 8 - likely having been made in June of 1977.  The date codes on internal components are consistent with the possible June 1977 assembly date.
  

Figure 3:
The inner synthesizer board - a bunch of counters.  LS-TTL
circuitry is used extensively, along with a few diode-type
PROMs for frequency/display lookup and counter set-up.
Click on the image for a larger version.

Evaluation

As I had other projects in the queue, it was only recently that I pulled this radio off the shelf.   Prior to setting it on my workbench, I blew the dust off it and carefully cleaned the front panel and around controls, throwing the knobs into an ultrasonic cleaner.

Powering it up, the unit worked - sort of:  I could hear noise, but it seemed a bit deaf - but the sensitivity changed wildly with a bit of thumping on the case, an indication of a dirty transmit/receive relay.  Even with a massively strong signal into the antenna connector - which produced a deafeningly-loud tone in the (external) speaker - I got no S-meter reading.  Many years ago, my friend and I used this radio (when it was still in his Jeep) and noticed this same problem and that it was also mitigated by a "percussive repair" and/or clicking the PTT several times, indicating that the Transmit/Receive relay may have problems.

Figure 4:
The main RF/AF board, post repair.  The layout is a bit
crowded, but pretty clean on a two-sided, glass-epoxy board.
This radio includes the optional 400 Hz CW filter.
Click on the image for a larger version.

Popping the top cover I could see that I had some work to do.  While it was remarkably clean inside for having been in a dusty Jeep for decades, I could see evidence of a few problems:  I saw at least one "blowed-up" capacitor near the audio amplifier.

Fortunately, the synthesizer itself seemed to be OK:  The tuning controls did their jobs properly, the tone in the speaker indicating that the radio was landing on the same frequency as the display.  The only "digital" problem seems to be that one of the segments of each digit on the display was constantly illuminated, weakly, possibly indicative of a problem with a segment driver.

Refurbishing

The first order of business was to replace the electrolytic capacitors.  As a few of them had clearly failed as evidenced by inspection, they all had to go - particularly since the radio had spent many summers in a closed vehicle during hot, Utah summers - plus, this radio is nearly a half-century old (which seems amazing when you consider that it's "digitally synthesized") so time would saved to simply "shotgun" them all.  Furthermore, many of the boards are "tethered" with soldered cables:  There is just enough slack to pull them out and work on the boards unsoldering only a wire or two, but doing so many, many times would not only be tedious, but risk fatiguing and breaking them - another reason to replace the capacitors in just one session.

Figure 5:
VCO/Synthesizer board.  There are two synthesizers - one
for them provides the 100Hz tuning steps.  Again, LS-TTL
logic is used, along with a few op-amps.
Click on the image for a larger version.

Capacitors, and more capacitors!

I took inventory, inspecting the entire radio and come up with the following list of capacitors - including those found in the PA module and places other than on the PC boards:

• (5) 470uf, 10 volt
  
• (2) 330uF, 16 volt (axial) 
  
• (2) 220uF, 16 volt
  
• (14) 100uF, 16 volt
• (13) 33uF, 16 volt
• (2) 10uF, 25 volt
• (4) 10uF, 16 volt
  
• (8) 4.7uF, 25 volt
• 1) 4.7uF, 16 volt
  
• (9) 1uF, 25 volt
• (4) 1uF, 50 volt
• (3) 1uF, 50 volt (axial)
• There are several dipped tantalum capacitors in low-level voltage and signal filtering lines that seem to be OK for now.   As none of these are on power rails there's no chance of a catastrophic failure (e.g. flames) should one short out.  These capacitors will be replaced in the future.
  

I suspect that the differing voltage ratings of some of the same-value capacitors was likely to save space (lower-voltage capacitors are generally smaller) and allow the use of less-expensive capacitors, but these days, capacitors are much smaller (and cheaper, in equivalent money) than their decades-old counterparts.  When ordering replacement capacitors I simply got same value rated for at least the voltage of the highest in the list above, but the new capacitors also had a temperature rating of 105C rather than the 85C of the original.  Since the electrolytic capacitors were pretty inexpensive - typically less than US$0.10/each for the smaller values - I ordered more than just the number above (in some cases, many more) in the event that I missed something.

Figure 6:
The pile of electrolytics removed from the radio.
Replacing every capacitor was the right choice!
Click on the image for a larger version.

Removing capacitors en masse is best done with the appropriate tools - particularly on an older circuit board.  Fortunately, I have a Hakko FR-300 desoldering iron/pump which made removal much easier and I was able to avoid damaging any traces on the board.

When replacing a bunch of capacitors, I prefer to do so methodically, moving from section to section on the circuit board - noting the polarity orientation of the capacitor before removing it and if there was any doubt as to which way it went, referring to the board layout diagram in the service manual - particularly since the circuit boards have neither solder mask or silkscreen as a visual reference.  Once a capacitor is replaced, I typically mark the top of the can with a colored marker to help make sure that I don't miss any.

One possible "gotcha" was that unlike modern electrolytic capacitors which are typically marked only on the negative lead, many (but not all) of the original capacitors in this radio had only their positive side marked - which was the custom of some manufacturers of the day - so I had to be particularly careful to identify the polarity correctly as I replaced each capacitor.

When I was done, the receiver seemed to be more "alive" than before, but it was still a bit deaf - and the synthesizer seemed to be a bit "wobbly", being very sensitive to slightly changes in power supply voltage.  The biggest change was the WWV receiver which was profoundly deaf prior to the capacitor change-out, but "normal" afterwards.

Capacitor brand implies longevity

After replacing the capacitors I went through the pile and found that most of them were "OK" - or at good enough that their respective circuits would have worked.  The brand seemed to be a pretty good indicator of which was likely bad:  The Japanese blue-label Nichicon and gray "Sun" and "Elna" brands were generally OK, the silver and gray Taiwanese "T.I." brand were all over the map, the lone "Sam Hwa" and "Towa" capacitors were marginal, but  all of the "Temple" branded capacitors (which seemed to have 1970 date codes, apparently already a few years old when the radio was made) were extremely bad.

After doing this I still believe that replacing all of the electrolytics was, in fact, the correct choice as I would have probably spent more time finding and diagnosing capacitors individually - and possibly suffered near-term failures - than simply swapping them all out.

A wobbly power supply

With all of the electrolytic capacitors replaced, I systematically went through the adjustment steps found in the user and service manual (which can be found online) - more or less.  Knowing that before you make ANY adjustments that you must make sure that the power supply is correct, I probed about with a volt meter noticing that the 11 volt supply was actually just below eight volts, likely accounting for its seeming deafness.  Locating the 11 volt regulator on the synthesizer board, I noted that the act of slightly adjusting the potentiometer resulted the voltage jumping, indicating that it was somewhat "stratchy", with the wiper likely not making good contact.  A bit of cleaning spray and exercising of this control resolved the issue and I reset the voltage to precisely 11.0 volts.

Figure 7:
Original S-meter coil.  It would seem that the coil winding
was broken in several places - hence, unsalvagable.
Click on the image for a larger version.

With the correct voltages now applied to the circuits in the radio, its sensitivity seemed to be much better and the synthesizer was no longer sensitive to fluctuations in the power supply, being able to tolerate a drop to about 11.25 volts at the radio's DC input before the synthesizer "wobbled".  

No S-meter!

Going through the alignment steps, I applied a signal from my generator and noted that while the sensitivity seemed to be about right - and the AGC was now working as it should - the S-meter did not move.  It's worth noting that the S-meter on this radio works ONLY when the meter switch is set to the "ALC" position - but I was getting no reading on any setting.  Using a voltmeter, I could see that the voltage across the S-meter's movement was increasing with the signal strength indicating that the AGC was working (which was also obvious by listening to off-air signals) but a quick check with an ohmmeter - after disconnecting one of the meter's leads - indicated that it was open circuit.

This was bad news, particularly since it was likely that I would never find a meter of the same, exact physical size - and even if I did find a replacement, I'd probably have to re-create the scale in the meter.  This wasn't impossible to do, but I took another path.

Figure 8:
The meter with its rewound meter coil using #30 wire.
As many turns were wound as would fit - the coil shaped to
prevent mechanical interference and then covered with
varnish to hold it in place.
Click on the image for a larger version.

Carefully disassembling the meter and inspecting it I noted that it was of the inexpensive "moving vane" type, the coil wound with very fine wire - probably around 46 AWG.  In probing very carefully I noted that one of these hair-thin wires was disconnected at the base of the coil.  Further probing showed that the wire itself was frayed where it was wound onto the phenolic paper stator - probably a victim of both temperature cycling and (possibly) some corrosion.  A bit of later inspection of the wire showed that it seemed "brittle" - something that I've seen on older gear:  I don't know if it's the copper hardening in some way or some sort of reaction between the wire, enamel and its environment that causes this.

Since the meter's coil was a total loss I decided to do something a bit drastic:  Rewind it.  Rather than trying to use #46 wire, I chose, instead, to use less-fragile wire - #30, which is about 10 times larger diameter:  I'd have used a smaller - but not overly fragile - wire (likely #36) if I'd had it on hand to get more turns and better sensitivity.  Of course, I was not going to get nearly as many turns on the stator as the original - which meant that it wasn't going to be as sensitive as it had been originally and would be unlikely to work properly in the circuit - but I had a plan for this.

Carefully winding the #30 wire into the phenolic stator until it was "full", I scrunched the coil down to reduce its height and then pushed it sideways to clear both the meter's axle and the moving magnets on the rotor before covering all of the windings with urethane varnish.  With the varnish dry, I reassembled the meter and found that it operated nonlinearly, particularly near the upper and lower ends of meter travel.

I quickly realized that the screwdriver that I'd used was slightly magnetic - and the two screws used to hold down the phenolic stator had become magnetized as well from using that screwdriver.  Using a TV degaussing coil (I could have used a soldering gun's magnetic field instead) I demagnetized the two screws and the screwdriver, solving this nonlinearity problem.

Re-zeroing the meter and using a series 470 ohm resistor and a variable bench power supply I found that the meter's full-scale sensitivity was about 23 milliamps - very much higher than the 500-ish microamp sensitivity that I'd calculated it to be originally.  In looking at the circuitry I noted that the negative side of the meter was grounded in all three of the front panel meter switch settings which meant that all I needed was to come up with a circuit to multiply the current linearly - and with one end of the meter being connected to circuit ground simplified that task:  Here's the circuit:

Figure 9:
Schematic of the circuit used to drive the re-wound meter on the CIR Astro 200.

The circuit shown in Figure 9 is the classic "precision current source" using an op  amp to drive a transistor and then the meter.  The input voltage is scaled with the trimmer potentiometer (R3) and applied to the non-inverting (+) input of the op amp with R4 in parallel to set an input resistance of about 250 ohms - which is my guess of the resistance of the original meter movement.  By its nature, the op amp will attempt to adjust its output to make the voltage on the inverting (-) input the same as the non-inverting (+) input and to do this, it turns on the transistor, causing current to flow through the meter and the current sense resistor, R2.  Resistor R1 is there to limit the maximum current to a "sane" value to prevent the meter from being slammed too hard in the case of an "oops".

Figure 10:
The as-built circuit from Figure 9 constructed on some
prototyping board.  This circuit is adhered to the top of the
meter itself.
Click on the image for a larger version.

The result of this is that this circuit will happily convert the voltage through R2 into a proportional current, the magnitude set by the adjustment of R3, allowing our now-rebuilt meter movement of arbitrary sensitivity to be used.

As the schematic shows, this circuit was built using the venerable LM324.  This device was chosen mainly because I have plenty of them, and it's one of the most common op amps that has an input and output voltage range that includes "ground" (V-):  Many "standard" op amps don't work near one or the other power supply rail and will work incorrectly if the input voltage is the same as the "V-" lead (ground, in our case) and about as many cannot output voltage down to the negative rail, either.

Since I needed only one of the four LM324's op amps, the other three were simply strapped to the power supply to keep them from floating and possibly causing noise issues:  It's possible that I could have used one or more of the op amp sections to directly drive the meter, but the single transistor was cheap and easy.  The circuit was built onto a small piece of glass-epoxy perfboard and attached to the top of the meter movement - the power supply from this circuit stolen from a trace containing the +11 volt supply found on the front-panel circuit board - but even the 13 volt, unregulated supply would have been fine.

Setting up the "new" meter

While the actual sensitivity of the original meter - which is believed to be around 500 microamps - is not known for certain, there is one step in the manual that is revealing in that it has no actual circuit adjustment, relying on the sensitivity of the meter itself for accuracy.  Because of this, we must do this step first and calibrate the sensitivity of our new meter circuit.

In the section of the manual about "Power Meter, Reflected Power Meter Adjustment" it describes connecting a 2:1 VSWR load (25 ohms using two 50 ohm dummy loads in parallel) and using an external power meter connected between the radio and the load:  The radio should be set for 40 meters for this step.  Switching to "CWW" (CW Wide - using the SSB filter) mode, set the Mic Gain to maximum (fully clockwise), key the radio and then increase the power (turning the Mic gain counter-clockwise to increase power) and adjusting R312 to limit the maximum power to 90 watts even when the Mic gain control is fully counter-clockwise (maximum power): These adjustments should be done quickly to avoid overheating the power amplifier.  The manual notes that with the meter set to the "REF" position, the meter should read "2" (for 2:1 VSWR) - and we quickly adjust R3 in Figure 9 for a reading of "2" on the meter.  Again, the key point here is that the REF meter gets its output from the reverse power detector amplifier - but since its threshold is fixed, when the power is being reduced by this circuit, it will always output the correct voltage/current to make the meter read "2".  In other words, this is fixed reference and we can use it to calibrate the meter for all other modes.

After this, the procedures for adjusting the S-meter, ALC and forward power readings outlined in the manual should be applied without further adjustment of R3, the 10 turn trimmer potentiometer.

It's worth reiterating the point that as the AGC, ALC, FWD and REF signals feeding the meter are ground-referenced, the circuit design was simple.  If the meter was driven by a "floating" circuit - one in which the negative side of the meter was at some potential other than ground - I would likely have used several sections of the LM324 configured as an "instrumentation amplifier" - one that measured the voltage drop across a fixed resistor (in lieu of current through the meter coil) regardless of the actual voltages.  This circuit would be somewhat more complex, but not overly so.

Radio alignment

With the capacitors replaced and the meter working, I went through the alignment steps outlined in the manual.  Fortunately, I had reviewed the manual in its entirety and noted a few "inconsistencies", notably:

• The listing of the carrier oscillator frequencies in the alignment steps shows the same frequency for LSB and USB.  The correct frequencies are shown on the previous page.
  

• When adjusting the ALC using potentiometer R296, the manual says to do so at mid-rotation in one place and and fully CW (clockwise) in another:  I presume that they meant fully CW.

Additionally, I would suggest the following additions to the procedure at the beginning of the procedure.

• Verify/adjust the setting of the 11.0 volt regulator on the synthesizer board (R92).
  

• Verify/adjust the 5.0 MHz oscillator on the synthesizer board using C52.
  

• If you had to re-wind the meter and add the circuit described above, I would do the reverse power meter calibration (described above) before the other meter calibration steps:  This is noted in the procedure at the end of this article.
  

After this, proceed with the alignment/calibration as described in the manual.  There is a revised/annotated alignment procedure at the end of this article.

Power cable

As I was unable to find the original power cable (it may still be in the Jeep) I needed to find the mating power connector.  Recognizing it as a "Jones" connector, I did a bit of research and found that I needed to get a Cinch-Jones S-306-CCT, which is a 6 pin female connector.  Unfortunately, this line of connectors was discontinued by the manufacturer several years ago, but EvilBay came to the rescue and I found a "new" one with the inline cable shroud and strain relief.

Using 12 AWG wire and an inline holder with a 30 amp blade fuse I put together a power cable with an Anderson power pole connector on the far end.  This allowed me to connect it to a high-current power supply so that I could get on with testing the radio's final power amplifier.

"Final" problems

With the radio otherwise aligned, I noted that I was unable to get anywhere near full power out of the power amplifier - about 35 watts on 80 meters, nearly 50 watts on 40 meters and 10-15 watts on 10 meters.  Checking the output on the main RF/AF board, I noted that the voltages were equal to or higher than noted in the manual so I removed the PA module from the back via its ten screws.

I immediately noticed something that further indicated that this was an "early" unit:  The PA driver transistors were Motorola, but marked as "604/438 Sample" and rather than using MRF454 outputs, they were CTC CD3435.  In poking around with an oscilloscope with about 10 watts of output on 40 meters I noticed that the waveforms on the collectors of the driver transistors were not equal - and neither were the corresponding waveforms on the output transistors:  This indicated that in each stage, at least one of the transistors had failed - or was badly degraded.

While annoying (the transistors aren't cheap!) it didn't surprise me.  It is (apparently) common for RF transistors from the 70s and, perhaps, into the early 80s to fail - even when not being used - due to internal defects that seem to "grow" over time.

Figure 11:
The repaired PA board with the new driver and output
transistors.
Click on the image for a larger version.

For the driver transistors, the originals were 2N6367, but the equivalent is the MRF433 or the 2SC2395 - but the MRF455 may work OK.  Rummaging around my bin of RF transistors I found a pair of pulled 2SC2395s (I don't recall where I got them) and put them in, saving me from spending about $100 for them.  Greeted with onlyabout 80 watts on 40 meters - and much lower power than that on 10 meters - I could still see from the waveform on the 'scope - probing the collector leads - that one of the output transistors was still an issue.

While I could get a pair of MRF454 transistors from RF Parts, I noted that they were available from Mouser Electronics for a lower price (about $55 each at the time of writing) and when they arrived, I saw that they sported a recent date code.  Plopping them in I saw that the PA was now capable of well over 125 watts on 80 and 40 meters - working as it should - allowing me to complete the adjustment procedures related to the ALC and power metering.

In testing the two original PA transistors out of circuit, I noted that both their beta and "diode drop" voltage were radically different.  I suspect that at least one of these devices had lost some "emitter sites" or tiny bond wires on the die, making it "less of a transistor" than it once had been.

With the final board now repaired, the radio met the specifications outlined in the manual:  100+ watts on all bands except 10 meters where the output was a bit over 85 watts.

A few loose ends...

The "stuck" LED segment

I also noted that the "stuck" segment on the LED display seemed to have fixed itself during a toggle of the "bright/dim" switch:  In looking at a YouTube video reviewing this radio I noted that it, too, had this exact problem - but I have no idea if it's common (e.g. happened on at least two different radios) or why it fixed itself - nor is there an obvious clue from the schematic diagram why that one particular segment would be affected on my radio and the one in the video.

Adding the clipper/limiter

As for the receiver, the sensitivity is good - but I decided to make a modification that apparently became standard in production just after this unit was produced.  I noted that when changing modes and bands, the S-meter would "pin" with the very loud "pop" that occurred, the AGC taking 5-10 seconds to recover

Figure 12:
The clipper circuit in tubing, installed in the radio.  One end
is connected to a leg of R290 - the other end to ground.
Click on the image for a larger version.

Noting that the manual included the description of a "Limiter" board - and that the radio in the YouTube channel - which had a serial number of about a dozen units higher - also had this board, I figured that this might be one of the reasons why it was added.

The circuit itself is simple:  Two pairs of diodes - one silicon and one germanium in series (for a clipping voltage of about 0.9 volts) - were placed in anti-parallel configuration and coupled with a 10uF capacitor.  This circuit was placed between ground and input of the AGC detector.  Rather than make a small circuit board as was done in the production units I simply wired the components in free space and covered them with PTFE and heat-shrink tubing, connecting the assembly between the AGC circuit and a handy ground pin as can be seen in Figure 12.

My suspicion about its later addition was confirmed:  While there is still a loud "click" when changing modes, the AGC now recovers much more quickly and the radio's AGC is also very much less prone to being badly deflected with a long recovery time when there is a loud static crash.

The T/R relay and filter module

Mentioned briefly, there was the problem with the intermittent T/R relay.  This is contained within a module that sits along the right edge, inside the radio that extends from the front panel to the back of the radio along with the band switch.

This module - in addition to the T/R relay - contains the receiver pre-selector filters, the transmit mixer filters and the transmit low-pass filters on a compact, shielded assembly.  To pull this assembly out of the radio would be quite a job, requiring the partial removal of the front panel, disconnecting (mostly unsoldering!) a number of wires, connectors and signal cables and pulling it out of the radio - something that I have not attempted to do.

Fortunately, the designers provided an access hole near the back panel of the radio (on the bottom side) that is covered with tape where one can burnish the relay's contacts and apply contact cleaner.  After both burnishing and the application of cleaner, the T/R relay is now working perfectly.

Using the radio

Tuning with switches

With the use of toggle switches instead of a round, "spinny" tuning knob, operating the Astro 200 is decidedly different than using a conventional radio.  As mentioned before, the previous owner told me that he thought using toggle switches was a bit better for tuning while bouncing along bumpy roads than a large knob - and in the days of analog radios, this was likely the case.

In perusing online references to this same radio, the users also noted that one quickly becomes accustomed to this method of tuning - but everyone had the same comment:  It's slow to tune across the band.  When powered up, this radio always starts at the bottom of the selected amateur band - and on 10 meters, this particular radio starts at 27.0000 MHz (transmit is inhibited below 28 MHz) which means that it takes about a minute to even get into the 10 meter band!

The AGC

The radio's AGC is not adjustable and the time constant is fine for CW, but a bit fast for SSB in my opinion.  As is common with many analog radios, the apparent AGC time constant gets shorter with more AGC action (e.g. higher S-meter reading).  This is a result of the "dB per Volts" curve getting steeper with many gain reduction schemes (e.g. more dB gain reduction per volt of change) effectively shortening the time constants.

Since this radio has a front panel RF attenuator control, switching this in to reduce the signal level helps with this effect somewhat.

Noise blanker

The noise blanker (enabled by pulling the "Squelch" control knob) seems to work pretty well, operating in the wideband IF prior to the crystal filters.  As is typical with noise blankers in analog receivers - and some modern digital radios - its efficacy is somewhat affected by very strong, adjacent signals which "desense" the noise detector - a difficult problem to overcome.

CW usage

As is common for radios of that era, the sidetone frequency in the CW mode has little to do with the frequency offset.  This radio uses USB and a positive transmit frequency shift when in CW which means that neither the display or the tuned frequency changes when going from USB to CW mode.  This was pretty common in the era (many makers - including Drake - did it this way) which meant that if the operator wanted to know the actual frequency of their transmitted signal that they would have to do some mental math.

One "quirk" that I need to investigate is that if this radio's heterodyne oscillator is set precisely according to the manual, the receive (and transmit) frequencies do not match the display, being offset by a bit more than 100 Hz.  This is easily corrected by setting the display to a known frequency, inputting a signal 1 kHz above and below (for USB and LSB, respectively) and adjusting for an audio tone of 1 kHz, but doing so shifts the passband of the crystal filters audibly - and in CW mode, it puts the center of the passband at about 1200 Hz.  This slight shift does not result in either "tinny" or "muffled" audio when using SSB on either sideband, and the radio sounds quite good on air!

As this offset - which is mentioned in the manual as being around 1000 Hz - appears to be programmed into PROMs, it does not seem possible to shift the local oscillator to overcome this issue - and while there's a difference between the USB and LSB passband, it is not a "show stopper" but a 1200 Hz-centered passband for CW is too high in my opinion.  I suspect that this being a very early production radio may have something to do with this issue and I'll have to think about possible ways to address it.

The Mic Gain:  When a "Mic Gain" control isn't really "Mic Gain"

Another unusual design feature of this radio is the transmit audio path.  From the microphone input, the signal path goes directly to the amplifier (there's no level adjustment preceding it) and into the clipper/compressor stage.  Interestingly, the clipper/compressor takes the form of a logarithmic amplifier which has less of a sharp "knee" than a typical clipper, making it quite effective in functioning very much like a compressor-type speech processor.

The designers made an interesting design choice here:  The control marked "Mic Gain" is placed in the signal path after the clipper/compressor - but this has some important implications.  In testing, I used an old Sure 440SL high impedance dynamic microphone which has a fairly high output level, but this caused the clipper/compressor to be "hit" very hard:  On-air reports indicated that that I was readable, but that my speech processing was very "heavy" and off-air recordings from a remote WebSDR verified this.  Since the "Mic Gain" control is between the clipper/compressor and the radio's balanced modulator, it affects only the RF output power and how hard one is "hitting" the ALC and doesn't affect the amount of audio compression at all.

What should really be done was to include a means of adjusting the microphone level into the clipper/compressor stage and this could take the form of having a level control on the microphone itself or in a box between the microphone and the radio, or, if the same microphone will always be used with the radio, put such a control inside the radio.

To accommodate this need, I rummaged around my parts box and found a 500k vertical chassis-mount trimmer potentiometer. This potentiometer was wired such that the "CCW" (counter-clockwise) end was grounded and the opposite end connected to the microphone jack with the audio to the radio on the wiper.

Figure 13:
A 500k potentiometer - reachable using a long, thin blade
screwdriver is accessible through the 1/4" TRS MIC/KEY
connector on this radio.  See text for more details.
Click on the image.

As depicted in Figure 13, behind the 1/4" MIC/KEY jack is a 5 volt regulator in a TO-3 case - but this doesn't line up with the connector, so I glued the potentiometer to a small piece of circuit board to allow it to be offset.  When I glued the pot to this board, I took care to avoid fouling the adjustment knob and after curing.

I then glued the small piece of circuit board to the top of the 5 volt regulator, taking care to offset it so that the potentiometer was aligned such that a long, thin blade screwdriver through the MIC connector could be used to adjust the level from the microphone being applied to the MIC amplifier.  The adhesive that I used was "Shoe Goo" which remains flexible:  I would not recommend epoxy, cyanoacrylate ("super") glue or hot-melt glue as none of these are a good choice in this application (e.g. the bonds will fail with temperature cycling and/or mechanical stress.)

Adjusting this new "MIC Level" control is an iterative process:  Plug in the mic - check the AGC deflection and output power, unplugging, and then making the necessary adjustments.  The goal here is to have enough audio to activate the compressor, but not so much that it sounds very "heavy" on-air.

As noted earlier, THIS radio uses a 1/4" TRS connector rather than the round, multi-pin connector used on later production models:  If this radio had this latter connector, blocking access to an adjustment behind it, I would have mounted the potentiometer facing down and drilled an access hole in the bottom of the chassis, probably making a right-angle bracket on which it was mounted.

Carrier balance

One interesting omission by the designers is the lack of a "carrier balance" control.  When SSB is generated, the "balanced modulator" - which literally mixes the audio with RF - this carrier is nulled on most radios via one or two adjustments to minimize the amplitude of the original carrier - but not on this radio.  This radio uses a diode-ring type of doubly-balanced mixer and by themselves these typically have a "bleedthrough" of between 25 and 35 dB- much less than the 40-50dB of a typical balanced modulator in an analog SSB transmitter after it has been carefully nulled.

What this means is that on 40 meters there is a carrier bleedthrough of about 200 milliwatts (which varies with band and operating temperature) when keyed down with no transmit audio.  Compared with a 100 watt output level, this represents a level that is 25-30 dB below peak power that cannot be adjusted.  This is nowhere near enough to impair efficiency of the transmitter by "wasting" power in the carrier but it is enough to be easily visible to the "waterfall police" using a modern digital radio if the conditions are good.

Frequency (in)stability

When it came out, this radio was remarkable compared to its contemporaries in that it didn't really drift:  You set the frequency and it just stayed there, within a few Hz.  Unlike most radios of the day, it moved only a few Hertz from the instant that it was turned on while most others at the time would change by hundreds of Hz in the first half hour or so - particularly if operated in a cold environment.

Compared to today's radios, the synthesizer is a bit crude - it has large (100 Hz) tuning steps and a bit slow to lock.  As the radio uses rather low reference frequencies (100 and 163 Hz) for its two synthesizers, their oscillators are rather slow to respond - but this also means that they are easily disturbed by slight changes in power supply voltage, mechanical vibration and just the physics of electronic circuits.

What this means is that the frequency can easily "wobble" a few Hz - or even 10s of Hz - around the nominal frequency in the short term.  This is generally unnoticeable for SSB usage or even RTTY - and most people will likely not even notice this when running CW - but it does make this radio unsuitable for some of the very narrow digital modes that are seen today, like FT-4, FT-8, WSPR, PSK31 or similar - a trait that it shares with its non-synthesized (VFO-only) predecessors.  These modern digital modes require that the radio be stable within 1-2 Hz at any given instant over the duration of the transmission/reception window - and this radio simply may not be able to do that.

Mechanical work

If you look very closely at Figure 1, you'll see aluminum brackets on either side of the front panel that were used to screw it to the underside of the dash on the CJ-7 in which it was mounted.  During my refurbishment, I drilled out the pop rivets on these brackets and filled the holes - and a few scratches - with metal-filled epoxy and sanded them down.

Even though the exterior of the case was in reasonable shape, it did show a bit of the wear of having been in a vehicle or two for over 40 years, so I decided to repaint it.  Having been in the vehicle for so long, the original light blue color was varied, depending on how much sun had faded it, but inside the top cover - out of sight - was a "virgin" section of paint to which I was able to find a very close match at the store:  Rustoleum satin "French Blue".  Just in case I - or someone else - wanted to match the original color, exactly, I masked off and left a patch of the original paint inside the lid.

Aside from a bit of wear on the knobs and slight yellowing of the panel meter's clear plastic - most of which was removed with the application of a bit of elbow grease and Novus plastic polish - the radio looks almost brand new.

On the air

I've made several contacts on the air with this radio and and have gotten good reports.  Even with the prevalence of waterfall displays these days, few people mention the slight carrier leakage - but I also wonder how many people actually look at their waterfall not to mention how many others would immediately recognize carrier leakage, anyway?

The addition of the "MIC Level" potentiometer was a good one.  When properly adjusted, the radio now sounds "normal" rather than very heavily "compressed" as before.

I haven't used the radio enough to become very adept at quickly tuning across the band using the UP/DOWN toggle switches, constantly overshooting signals - but I'd guess that this would be a skill that could be readily acquired.  At the risk of sacrilege, I'm considering the addition of a small, PIC-based microcontroller board that will track the button presses and the current band selection to "pre-set" the frequency when the unit is powered up and band is changed, making it a bit more convenient to use:  Such a modification would be completely reversible

Final comments

One should treat this radio in a way similar to "vintage" radios of decades gone by.  It's remarkable in its capability and design considering that it's nearly a half-century old and it needed relatively little in repair - and even more remarkable in that most of the parts that it uses are still available from electronics suppliers at the end of the first quarter of the 21^st century.

As a general-purpose radio for SSB, CW and even RTTY operation, it's still very usable:  It's small size belies its capabilities, particularly in context with its vintage.  Being made prior to 1980, it obviously lacks the WARC bands (30, 17 and 12 meters) - but so do other radios of that time period.  Once the radio was restored - mostly a matter of replacing electrolytic capacitors - it operates pretty much as it did when it was new and it would not seem out of place on the air among modern radios on the air.

Given its quirks (no tuning knob being the most obvious) it is a bit of curiosity, reminding the user of a time just before completely analog radios gave way to synthesized radios becoming the norm - a revolution not too dissimilar to the more recent trend of "analog" radios giving way to those that are almost entirely digital from the antenna port to the speaker.

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Alignment notes

Here are notes related to aligning the Astro 200 (Non "A" version) - although they should be generally correct for the "A" version as well.  These should be used to augment the instructions noted in the operation/maintenance manual.

Power supply check - IMPORTANT!

• Verify 11.0 volt power supply - adjust R92 on synthesizer board as appropriate.
• Verify that 11.0 volt supply will remain stable down to a supply voltage of at least 11.5 volts as measured on the radio's voltage input.
  
• Verify 8.0 and 5.0 volt supplies (each being +/- 0.25 volts of nominal).  Note that there are two separate 5.0 volt regulated supplies.
  

Reference (Master) oscillator:

• Frequency counter to set to 5.000000 MHz or use WWV setting (which listens to 10 MHz via a direct-conversion receiver) and listen for zero beat
  
• Set C52 for 5 MHz, exactly.  This is accessible via a small hole in the bottom cover.
  

Carrier oscillator:

• USB/LSB
• MIC Gain CCW
• RIT and FINE at 12:00 position
• MODE to USB
  
• Key radio and adjust C180 for 5.601650 MHz
• MODE to LSB
• Key radio and adjust C174 for 5.598350 MHz
• MODE to CWN
• MIC Gain fully CW (for minimum CW TX power) and connect radio to dummy load.
  
• Key radio and adjust C204 for 5.60060 MHz

RX Delay adjustment - used to delay time between release of PTT/VOX and RX activation

• Adjust R239 for desired delay time preference in switching from TX back to RX when PTT is released.
  

VOX Trip and Anti-Trip

• Turn on VOX and set volume to desired level using your typical ham shack speaker/audio environment.
  
• Adjust R181 for VOX activation level with normal speaking voice.
  
• Adjust R158 for anti-VOX level with signals/static present to prevent unwanted triggering.
  

Meter adjustments.  Be sure to view meter "straight on" and consistently to minimize parallax for the readings below.

• VSWR shutdown/reflected power:  R312 calibrates the VSWR shutdown of power.  DO THIS STEP AS QUICKLY AS POSSIBLE.  Be sure to view the meter "straight on" to avoid parallax in the following steps.
• NOTE:  As mentioned earlier in this article, I had to "repair" the meter by re-winding its coil and using an external driver circuit.  If you restore the meter in this manner, do THIS step before the other "Meter adjustment" steps.
  
• Connect two 50 ohm loads in parallel for 2:1 VSWR (25 ohms) - use the shortest length coaxial cable possible.
• Set to a mid-band frequency on 20 meters.
• Set meter switch to REF
  
• In CWW mode, turn MIC gain fully CW, key transmitter.
• Increase power.  Quickly adjust R312 so that the forward power can not be increased to more than 90 watts on the forward meter and unkey.
  
• In VSWR mode, the meter should read about 2.

• Forward power:  R306 calibrates forward power reading.
  
• Connect 50 ohm dummy load and power meter.
• Set the radio to a mid-band 40 meter frequency and pre-set the MIC gain control fully CW to set minimum power.
  
• In CWW mode, turn MIC gain CCW, key transmitter and adjust for 100 watts on the power meter.
  
• Adjust R306 for full-scale indication indication (to the "Set" marking) on meter.
  

• ALC Setting.  Be sure to view the meter "straight on" to avoid parallax in the following steps.
• Connect 50 ohm dummy load and power meter.
• Set MIC gain to 12:00 position, meter mode to FWD. (CONFLICT:  Manual says says fully CW in earlier section about adjustment)
• Note:  Since the transceiver has no actual "Microphone Gain" adjustment prior to the clipper, the fully-CW adjustment setting would make sense as it will maximally drive the ALC (worst-case).
  
• Key transmitter and whistle or produce tone into the microphone.
  
• Adjust R296 for a reading of an average of 40 watts on the power meter.  This should correspond roughly with a reading of "30 over" on the meter.
  

• ALC Meter setting.  Be sure to view the meter "straight on" to avoid parallax in the following steps.
• Connect to 50 ohm dummy load.
  
• Set mode to CWW, meter to ALC and set MIC Gain fully CLOCKWISE
• Key transmitter:  There should be low/no power.
• Adjust R291 for FULL SCALE ALC meter deflection.

AGC set-up.  Be sure to view the meter "straight on" to avoid parallax in the following steps.

• Connect signal generator to antenna input and mode to CWW.
• Set front attenuator switch to OFF (down)
  
• Set for 20 meters and tune to a frequency mid-band and adjust the signal generator so that there is a tone of about 1 kHz
• Set the signal generator for an output of 1.5 microvolts (-103.4dBm)
• Adjust R280 for an S-meter reading of S3.
  
• Increase the signal to 50 microvolts (-73dBm)
• Adjust R272 for an S-9 meter reading
  
• Re-check the steps above for 1.5 and 50 microvolts and adjust as necessary.

Sidetone Level set

• Connect to 50 ohm dummy load, set to CWN and adjust MIC gain control fully CLOCKWISE (minimum power)
• Key transmitter and adjust R257 for desired sidetone level in speaker.

In-depth alignment:

Carrier oscillator peaking

• Using and oscilloscope or high-impedance RF voltmeter, measure the amplitude at the base of Q60
• Adjust L11 for maximum amplitude.  Use only a plastic adjustment tool to avoid breaking the core.
  
• Check carrier oscillator frequencies as noted above - adjust as appropriate.

TX mixer and ALC attenuator

• Connect 50 ohm dummy load.
• Set to CWN and adjust fully CCW (max power)
• Key down and adjust L6 for maximum signal on collector of Q20 using an oscilloscope or RF voltmeter.  Use only a plastic adjustment tool to avoid breaking the core.

WWV receiver adjustment

• Set MODE switch to WWV and turn AF gain all of the way down.
  
• Apply signal generator at 50uV (e.g. -73dBm - equivalent to S9) to antenna, offset from 10 MHz by about 1 kHz so that a tone will be heard.
  
• Connect AC voltmeter to speaker and adjust level to indicate on meter, but keep it well below clipping.
  
• Tune L15 for maximum speaker output.  Use only a plastic adjustment tool to avoid breaking the core.
• Remove input signal.
  
• Using a high-impedance RF voltmeter or oscilloscope, adjust L16 for maximum 10 MHz at collector of Q77.  Use only a plastic adjustment tool to avoid breaking the core.
• A signal of 5uV (-93dBm) should be audible.

Noise blanker adjustment

• Connect a signal generator to the antenna input.
• Adjust receiver and signal generator for a mid-band 20 meter frequency and adjust for a level of 100uV (-67dBm) and an approx. 1 kHz tone in the speaker.
• Adjust L9 and L10 for maximum voltage on D30.  Use only a plastic adjustment tool to avoid breaking the core.

SWR Bridge adjustment

• Connect two 50 ohm loads in parallel for 2:1 VSWR (25 ohms) - use the shortest length coaxial cable possible.
  
• Set MODE switch to CWW and set MIC Gain control fully CW (minimum power) and set to mid-band on 20 meters.
• NOTE:  Do the following measurements as quickly as possible to minimize stress on power amplifier.
• Key down.  Increase power (MIC gain turned CCW) and note that SWR protection limits to 90 watts as adjusted in SWR protection steps noted above.
• Note power reading on front panel meter and external wattmeter (if used) and then un-key.
  
• In the same manner, check the maximum power into the same 2:1 VSWR on 80, 40 and 15 meters.
• Adjust C3 as necessary for flattest (most consistent) power reduction on all bands:  Power should be between 80 and 105 watts.
• On 10 meters, power into a 2:1 VSWR may be in the 70-80 watt range.

RF Tuning assembly

This is the unit inline with the BAND switch.  The coils noted below correspond with the frequency range and should be adjust for best response across that noted below.

NOTE:  As the receive and transmit filter inductors are not normally accessible, it is necessary to remove the band switch module to perform these adjustments - a laborious task which requires unsoldering a lot of different cables and removal of the front panel.  It should be done ONLY if problems are suspected.  These adjustments should only be done with a spectrum analyzer and tracking generator OR a VNA/SNA.  If the sensitivity of the receiver is adequate and the transmit drive is within specifications, there is probably little need to even touch these adjustments.  As my radio was "up to spec" in terms of sensitivity and TX drive, I did not pull the module and make any adjustments.

Use only plastic adjustment tools to avoid breaking the cores!

Receive filters

• 80 Meters:  L101, L102 - 3.5-4.5 MHz
• 40 Meters:  L103, L104 - 7.0-7.5 MHz
• 20 Meters:  L105, L106 - 14.0-14.5 MHz
• 15 Meters:  L107, L108 - 21.0-21.5 MHz
• 10 Meters:  L109, L110 - 28.0-30 MHz
• WWV:  L111, peaked at 10.0 MHz.

Transmit mixer band-pass filters

• 80 Meters:  L201, L202 - 3.5-4.5 MHz
• 40 Meters:  L203, L204 - 7.0-7.5 MHz
• 20 Meters:  L205, L206 - 14.0-14.5 MHz
• 15 Meters:  L207, L208 - 21.0-21.5 MHz
• 10 Meters:  L209, L210 - 28.0-30.0 MHz

Synthesizer adjustments

Unless the synthesizer has difficulty locking - particularly at the upper or lower edge of one or more bands - there's probably no need to make these adjustments.

Major Loop VCO

• Adjustments should be made at low edge of the respective band.
• Coil should be set for a voltage of 2.5 +/- 0.25 volts on R18
• Exception:  For units that can tune to 27.0 MHz, the voltage should be 3.0 +/- 0.25 volts when tuned to 28.0 MHz.
• Start with the highest band first.  For the progressively-lower bands, the following inductors are in series meaning that a higher-band coil's adjustment will affect all lower bands.
  
• 10M:  L9
• 15M:  L8
• 20M:  L7
• 40M:  L6
• 80M:  L12
  
• Notch filter:  Adjust R11 and R15 for minimum amplitude of 100 Hz signal on the output (pin 6) of IC21 (0.035Vpp or lower)

Minor Loop VCO

• Adjustments should be made on low edge of the respective band.  (Manual isn't clear about this)
  
• If adjustment is needed, it will be necessary to remove the brass shield by unsoldering its three corners.
• Coils should be set for a voltage of 1.6 +/- 0.2 volts as measured on R21.
• Start with the highest band first.  For the progressively-lower bands, the following inductors are in series meaning that a higher-band coil's adjustment will affect all lower bands.
• 10M:  L5
• 15M:  L4
• 20M:  L3
• 40M:  L2
• 80M:  L1
  
• Notch filter:  Adjust R25 and R27 for minimum 165 Hz signal (0.025Vpp or lower) on the output (pin 6) of IC20.
  

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This page stolen from ka7oei.blogspot.com

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